05420005

1502 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 3, MARCH 2012

An Improved Direct Torque Control for Three-LevelInverter-Fed Induction Motor Sensorless Drive

Yongchang Zhang, Member, IEEE, Jianguo Zhu, Senior Member, IEEE, Zhengming Zhao, Senior Member, IEEE,Wei Xu, Member, IEEE, and David G. Dorrell, Senior Member, IEEE

Abstract—A sensorless three-level neutral-point-clampedinverter-fed induction motor drive is proposed in this paper. Theconventional direct torque control (DTC) switching table fails toconsider the circuit limitations, such as neutral-point-balance andsmooth vector switching, caused by the topology of a three-levelinverter. Two kinds of modified schemes for three-level DTC areproposed to solve these problems. They also provide performanceenhancement while maintaining robustness and simplicity. Fuzzylogic control and the speed-adaptive flux observer (with novelgain and load toque observation) are introduced to enhance theperformance of the system. The issue of large starting current isinvestigated and solved by introducing the technique of preexci-tation. A 32-bit fixed-point DSP-based motor drive is developedto achieve high-performance sensorless control over a wide speedrange. The effectiveness of the proposed schemes is confirmed bysimulation implementation and experimental validation.

Index Terms—AC motor drives, adaptive observer, fuzzy logic,induction motor (IM) drives, speed sensorless, three-level inverter,torque control.

I. INTRODUCTION

D IRECT torque control (DTC) has emerged as an alterna-tive to field-oriented control (FOC) for high-performance

ac drives, since it was first proposed in the mid-1980s [1], [2].The merits of DTC can be summarized as fast torque response,simple structure (no need of complicated coordinate transforma-tion, current regulation, or modulation block), and robustnessagainst motor parameter variation [3]–[5].

On the other hand, multilevel inverters have attracted con-siderable attention, especially in high-power application ar-eas [6]–[9]. The three-level neutral-point-clamped (NPC) in-verter is one of the most commonly used multilevel inverter

Manuscript received November 27, 2009; revised January 10, 2010; acceptedFebruary 5, 2010. Date of current version February 7, 2012. This work wassupported in part by grant from the Natural Science Foundation of China underProject 50737002. The part of this paper was presented at the Annual Confer-ence of IEEE Industrial Electronics Society (IECON), 3–5 November, 2009,Porto Portugal. Recommended for publication by Associate Editor J. O. Ojo.

Y. Zhang was with the Faculty of Engineering and Information Technology,University of Technology, Sydney, NSW 2007, Australia. He is now with thePower Electronics and Motor Drive Engineering Center, North China Universityof Technology, Beijing, 100144, China. (e-mail: [email protected]).

J. Zhu, W. Xu, and D. G. Dorrell are with the Faculty of Engineering and In-formation Technology, University of Technology, Sydney, NSW 2007, Australia(e-mail: [email protected]; [email protected]; [email protected]).

Z. Zhao is with the Department of Electrical Engineering, State Key Labo-ratory of Power Systems, Tsinghua University, Beijing 100084, China (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2010.2043543

topologies in high-power ac drives. Compared to the standardtwo-level inverter, the three-level inverter presents its superi-ority in terms of lower voltage distortion, lower stress acrossthe semiconductors, less harmonic content, and lower switch-ing frequency [10]. Because of the merits mentioned earlier, thethree-level inverter-fed DTC motor drive has become an im-portant research topic in industry and the academic communityover the past decades [11]–[19].

Compared to two-level DTC, three-level DTC motor driveshave two particular aspects. The first is concerned with elec-tromagnetic performance enhancement, including torque ripplereduction and low-speed performance improvement; in a sim-ilar manner to that of two-level DTC. The second is inheritedfrom the topology of three-level inverter, i.e., the neutral pointpotential balance and smooth vector switching.

Neutral point unbalance will cause higher voltage in the powersemiconductors. This increases the demand of capacity so thatthe cost increases. The balance of neutral point potential can beachieved by exploiting the opposite effect of redundant vectorson the neutral point potential [8]–[10].

Smooth vector switching requires that there is no excessivevoltage “jumps” in both phase voltage and line voltage. Phasevoltage “jumps” fail to utilize the advantages of three-level in-verter and may endanger the safe operation of inverter by caus-ing full dc voltage across one switch during transient commuta-tion, especially when the inverter uses only one snubber circuitfor three phases. Line voltage “jumps” produce more harmoniccontent in output voltage waveform, which is unfriendly to themotor and increase the burden on the filter.

Most of the literature on three-level inverter DTC motor drivesconcentrates on performance improvement. These employ com-plicated algorithms based on analytical methods [12]–[14], co-ordinate transformation [15], and continuous space vector mod-ulation [19], among others. Although good performance, suchas torque ripple reduction or fixed switching frequency, wasachieved (at the expense of sacrificing the simplicity of DTC),they failed to consider the limitations caused by the circuit topol-ogy, i.e., neutral point potential balance was sometimes not ad-dressed and smooth vector smoothing was almost not mentionedat all. Some paper (such as [18]) considered the allowable tran-sitions between different vectors. However, there still exists thepossibility of undesired excessive line voltage jumps, e.g., from++− to +00 [18, Fig. 7].

The aim of this paper is to achieve high-performance sen-sorless DTC for a three-level inverter-fed induction motor (IM)drive, as well as considering the neutral point potential balanceand smooth vector switching. Two kinds of DTC schemes are

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ZHANG et al.: IMPROVED DIRECT TORQUE CONTROL FOR THREE-LEVEL INVERTER-FED INDUCTION MOTOR SENSORLESS DRIVE 1503

Fig. 1. Main circuit of three-level inverter.

proposed to obtain high-performance control of torque and statorflux and the switching principle is illustrated to meet the limita-tions required by the topology of three-level inverter. As DTCis often referred for its drawbacks of large starting-up currentand flux droop at low speed [12], [20], the technology of preex-citation is introduced to limit the starting current [16], [21]. Toenhance the low-speed performance, especially under sensorlessoperation, a fuzzy logic controller (FLC) [22] is incorporatedin the outer speed loop to improve the speed response and aspeed-adaptive flux observer with novel gain and load torqueobservation [19] is employed to estimate stator flux, torque, androtor speed over a wide speed range. Simulations, as well asexperimental results, are presented to validate the effectivenessof the schemes proposed in this paper.

II. PRINCIPLE OF DTC AND THREE-LEVEL INVERTER

A. Three-Level Inverter

The main circuit of a three-level inverter is illustrated in Fig. 1and there are three states for one-phase output: +Udc/2, 0,and−Udc/2, with the neutral point as reference. To be moreuniversal, the negative dc bus voltage will be selected as ref-erence ground, and the three states are indicated by “2,” “1,”and “0,” for “+Udc/2,” “0,” and “−Udc/2,” respectively. Moreoutput levels provide more freedom in vector selection and itis possible to synthesize waveforms that are more sinusoidal inshape. However, the complexity of vector selection rises withthe number of vectors. In addition, there are further problems,including neutral point balance and smooth vector switching,which need to be carefully solved.

B. Basic Principle of DTC

A mathematical model of an IM described by space vectorsin a stationary frame can be expressed as follows:

us = Rsis + pψs (1)

0 = Rr ir + pψr − jωrψr (2)

ψs = Lsis + Lm ir (3)

ψr = Lm is + Lr ir (4)

where us , is , ir , ψs , and ψr are the stator voltage vector, statorcurrent vector, rotor current vector, stator flux linkage vector,and rotor flux linkage vector, respectively; Rs , Rr , Ls , Lr , andLm are the stator resistance, rotor resistance, stator inductance,rotor inductance, and mutual inductance, respectively; and ωr

is the rotor speed and p = d/dt is the differential operator.From the stator voltage equation (1), it can be seen that, by

omitting the stator resistance voltage drop, the stator flux canbe controlled directly from the stator voltage. This is a crudeanalysis and may cause error at low speed.

The electromagnetic torque can be obtained from

Te =32Np

Lm

σLsLrψr ⊗ ψs =

32Np

Lm

σLsLr‖ψr‖ ‖ψs‖ sin δsr

(5)where δsr is the spatial angle between the stator and rotor fluxes,Np is the motor pole-pair number, and Te is the electromagnetictorque. In DTC, the amplitude of the stator flux is kept constantand a fast torque response is obtained by changing angle δsr

quickly. From (1)–(4), the relationship between the stator androtor fluxes can be obtained as

pψr +(

1σTr

− jωr

)ψr =

Lm

σLsTrψs (6)

where σ = 1 − L2m /(LsLr ) and Tr = Lr/Rr . Equation (6) in-

dicates that the dynamic response of the rotor flux is a first-orderlag with respect to the stator flux, so the torque can be changedquickly by changing the angle of stator flux.

III. TORQUE AND FLUX RIPPLE REDUCTION IN DTC

The conventional switching table for two-level DTC cannotbe directly extended to three-level DTC. This is because it is notonly the performance that is of concern, the limitation caused bythe topology of three-level inverter also should be considered.Two kinds of scheme for three-level DTC are proposed in thispaper to solve these problems.

A. DTC Method I

The first DTC scheme utilizes the vectors of the three-levelinverter directly and inserts appropriate intermediate vectors tomeet the demand of neutral point balance and smoothed vectorswitching. The switching principle is described in detail in thefollowing.

First, the vector is selected according to the demand of theflux and torque; vector switching and neutral point balance willbe considered later. Fig. 2 shows the space vector diagram fora three-level DTC control strategy and its sector division. The27 vectors are marked by V1 ,V2 . . .V27 . There are 12 sectorsand the shadowed area is the first sector, which is different fromthat of the conventional two-level DTC. The basic principlesof the vector selection are shown in Table I and these meetthe demands of the flux and torque; k represents the stator fluxlocated in kth sector. In addition, “↑” means increase, “↓” meansdecrease, and “=” means no change is needed. It should be noted


Fig. 2. Space vector diagram for the three-level DTC.

TABLE IVECTOR SELECTION TABLE FOR THREE-LEVEL DTC

that there may exist more than two vectors to meet the demandsof the flux, and the one which meets the torque response betteris preferred.

However, in many cases, the selected vector usually cannotmeet the requirements of the vector switching and neutral pointbalance, which means that the selected vector cannot be appliedto the three-level inverter directly. For example, suppose thestator flux is located in the first sector, and the working voltagevector at the moment is V1(200). To increase the stator fluxand torque, according to Table I, V3(220) would be selected.But there is a high-voltage jump in phase B from 0 to 2, whichshould be avoided. In this case, V2(210) will be inserted as anintermediate vector to smooth the high-voltage jump. There arethree aspects with respect to voltage jumps: 1) phase voltagejump, 2) line voltage jump, and 3) three-phase jump at the sametime. High-voltage jump increases harmonic content and thestress across power semiconductors, which negates the advan-tages of the three-level inverter. To overcome this problem, anappropriate intermediate vector should be inserted to meet therequirement of the voltage jump.

Another issue is the problem of neutral point balance, which isinherited from the topology of three-level inverter. Many papershave investigated this problem. Neutral point balance is mainlycontrolled by selecting appropriate small vectors [9], [10]; thisis because of the opposite effects of redundant vectors. In this

paper, we also adopt the redundant states of small vectors tokeep the neutral point balance.

The final vector selection rules are obtained by consideringthe aspects introduced earlier, and the principles are summarizedas follows.

Step I: Select vector according to the demands for flux andtorque, which are listed in Table I.

Step II: If the selected vector cannot meet the requirementof the voltage jump and neutral point balance, an appropriateintermediate vector will be inserted. The principles for selectingthe intermediate vectors are as follows.

1) Large vectors or middle vectors should be selected prefer-ably to increase the utilization ratio of the dc bus.

2) Middle vectors can switch to adjacent small vectors andlarge vectors freely.

3) Large vectors can switch to the small vectors in the samespatial orientation.

4) Small vectors can switch to zero vectors freely.5) When small vectors are available, select the one, which

can meet the requirement of neutral point balance. Usingthe steps earlier, an appropriate vector can be selected tomeet the demand of the flux and torque, as well as therequirement of voltage jump and neutral point balance,which ensures the safe operation of the three-level inverter.

B. DTC Method II

In DTC method I, by inserting the appropriate intermedi-ate vector, the problems of neutral point balance and smoothvector switching were solved. However, it may degrade theperformance of torque and increase the complexity of vectorselection, so another scheme is proposed here. Method II makesuse of synthesizing vectors, which is termed discrete space vec-tor modulation (DSVM). This was first proposed in two-levelDTC [4], [5]. The two-level DSVM-DTC incorporates a morecomplicated and accurate switching table by dividing one sam-pling period into two or three intervals, and thus, more vectorsare obtained. In [4], speed is also taken into account and morelevels of hysteresis are adopted to make the switching table moreaccurate. The benefits of DSVM-DTC are reduced torque andflux ripple at a little extra expense of computational time [4].This paper extends DSVM to three-level DTC by using syn-thesizing vectors and the main aim of introducing DSVM is tosolve the problems of neutral point balance and smooth vectorswitching. To reduce the complexity of the algorithm, the samestructure as Table I is adopted and the speed was not taken intoaccount in the switching table, as in [4].

First, we should synthesize some vectors, which are expectedto solve the problems of neutral point balance and smoothswitching between any two vectors simultaneously. This meansthat the vector selection, according to the need of the torqueand flux, is decoupled from the circuit limitation introduced bythe three-level topology. A series of novel synthesizing vectorsare produced, which are illustrated in Fig. 3 and marked byV s1 , V s2 , . . . ,V s12 . Take V s1 , for example, it is synthesizedby the nearest three vectors, namely, V1(200), V2 (210), andV13/V14(100/211). The duration time of each vector can be


Fig. 3. Synthesis vector diagram.

TABLE IINOVEL VECTOR SYNTHESIS

calculated easily by utilizing the principle of volt-second bal-ance. To smooth the vector switching, zero vector V26(111) isincorporated at the beginning and ending of each synthesizesequence, taking up 10% or less (depending on the minimumpulsewidth of switches) duty of the whole period. The 12 syn-thesizing vectors are distributed uniformly in the fixed-anglespace (15◦ for V s1) with constant or variable amplitude. In thispaper, constant amplitude for the synthesis vector is selected forsimplicity, so the duration of each vector in V s1 ,V s2 , . . . ,V s12can be obtained offline and stored in a look-up table for real-time implementation. The final synthesizing vectors are listed inTable II and the sector division for three-level DTC is presentedin Fig. 3, which has a 15◦ shift compared to that in Fig. 2. FromTable II, it is seen that the switching between any arbitrary twovectors or adjacent vectors in a synthesis sequence are smooth.The neutral point balance can be solved by adjusting the “last-ing time” of the small vectors in one sampling period. TakingV s1 as an example, 211 and 100 are a pair of small vectors andtheir total “lasting time” is fixed during one sampling period,but their individual working time can be arranged according tothe requirement of neutral point balance.

DTC method II employs the same switching table, as shownin Table I, except that the selected vector is replaced by the novelsynthesis vector listed in Table II. For example, if the selectedvector number is k according to Table I, the synthesized vector

Fig. 4. Example of switching pattern. (a) DTC method I. (b) DTC method II.

Fig. 5. Diagram of the FLC.

V sk will be selected as the output vector, and number 26 meansthe zero vector 111. For DTC method I, a further step II should betaken before the final vector is selected; however, this processis not needed in method II, which simplifies the selection ofvector. An example of switching pattern for the two kinds ofDTC method is illustrated in Fig. 4. It is seen that for DTCmethod I, there is only one vector in one sampling period, whilethere is a sequence of vectors for DTC method II, with 111 asthe beginning and ending.

IV. PERFORMANCE ENHANCEMENT OF DTC

To enhance the dynamic performance and steady-state accu-racy, as well as robustness to external disturbance and motorparameters variations, FLC is used in the outer speed loop [22].Furthermore, a full-order speed-adaptive flux observer withnovel gains is employed in this paper, and a load torque ob-servation is introduced to improve the dynamic response of thespeed estimation [19]. Also, the well-known problem of largestart-up current in DTC is addressed by introducing the tech-nique of preexcitation. These three aspects are described in thissection in detail.

A. Fuzzy Logic Speed Control

FLC can handle complicated nonlinear systems, which have adegree of uncertainty. It does not require exact system modelingand parameters; this makes FLC very suitable for motor drivecontrol [22], [23]. A classical FLC is composed of three parts:fuzzification of input variables, fuzzy reasoning, and defuzzifi-cation. A diagram for the FLC used in this paper is illustratedin Fig. 5. The inputs are the error between commanding valueand real value, and its derivative. The output is the control incre-ment du, whose integral is the real output. The input and outputvariables are scaled to the range of (−1.4, 1.4). For the inputvariables, there are seven variables defined in the fuzzy sets: PB,


Fig. 6. Input membership function.

Fig. 7. Output membership function.

PM, PS, Z, NS, NM, and NB in descending order. To improvethe dynamic performance and obtain more refined output, twoother variables are added in the fuzzy sets for the output vari-ables, i.e., PVB and NVB. Figs. 6 and 7 show the membershipfunctions of the inputs and the output, respectively. To obtain afast response for dynamic performance, and high accuracy forsteady state, the asymmetric triangle is selected as the member-ship function, which is different from the conventional design.Table III shows the inference rule used in this paper, which isthe key part of the FLC. By careful design of the inference rule,excellent performance can be achieved. The mapping relation-ship between the input and the output variable is illustrated inFig. 8.

B. Speed-Adaptive Flux Observer

Sensorless motor drives can work in a hostile environment,which increases their reliability and decreases their complexity

TABLE IIIRULE MATRIX OF FLC

Fig. 8. Control surface of FLC.

and the cost of the system. Hence, sensorless operation is veryattractive proposition in many industrial applications. Amongvarious sensorless approaches, observer-based techniques arevery popular and versatile [24]. Compared with open-loop speedestimation techniques, the observer-based methods are morerobust to motor parameter variations because they introduce anerror feedback of the stator current estimation.

In a classical observer, the gain is such designed that the polesof observer are proportional to those of the IM (usually a fac-tor k > 1). This strategy of gain selection results in poles withlarge imaginary parts, which may cause instability in high-speedoperation [25]. In addition, the observer gain usually containsspeed-dependent terms, which may be affected by the accuracyof the observed speed. This paper adopts a speed-adaptive fluxobserver with novel gains to improve the stability of the sys-tem. Load torque observation is also incorporated to enhancethe dynamic response of the speed estimation [19], [25]. Themathematical model of the observer can be expressed as follows:

pis = −[(

1σTs

+1

σTr

)− jωr

]is +

1σLs

(1Tr

− jωr

)ψs

+1

σLsus + G1(is − is) (7)

pψs = −Rs is + us + G2(is − is) (8)

pωr =p

J(T e − T L ) + Kω [Δis ⊗ (λLr ψs − is)] (9)

pT L = −KT [Δis ⊗ (λLr ψs − is)] (10)


where is and ψs are the estimated stator current and the es-timated stator flux, respectively. J is the motor inertia andTs = Ls/Rs . G1 = −(g1r + jg1i) and G2 = −(g2r + jg2i) arethe observer gains. kω and kT are positive constant gains. Thispaper employs a novel observer gain [19] to improve the stabilityof observer, where

g1r = 2b (11)

g1i = 0 (12)

g2r = σLsb (13)

g2i = 0 (14)

and b is a negative constant gain.

C. Decreasing Starting Current

To decrease the starting current and maintain sufficient start-ing torque, preexcitation of the stator flux is proposed in thispaper. For an open-loop V /f motor drive, the starting currentis restricted by switching between nonzero and zero vectorsaccording to the error between the reference and measured cur-rents [21]. It cannot control the stator flux accurately becauseno observer or estimator is employed to obtain the stator flux asa feedback.

In the system proposed here, as an adaptive flux observeris incorporated in the system, the stator flux can be controlledaccurately and achieve the preexcitation in the true sense. Itshould be noted that current limitation is needed and it worksonly during the preexcitation process, because the magnetizationprocess without current limitation still produces undesired largecurrent. However, the current limit in the preexcitation doesnot have to be large to produce the needed stator flux. Theamplitude of current limitation only affects the lasting time ofpreexcitation. In this paper, the current limit it set to be 80% ofrated current. During the preexcitation process, when the currentexceeds the limitation setting, a zero vector will be selected toreduce the current; otherwise a fixed vector will be selected toproduce stator flux, which acts in a bang-bang fashion. Whenstator flux tilts the lower limit of flux hysteresis, the process ofpreexcitation is terminated and the motor can start with sufficienttorque by DTC, because there is sufficient flux to produce torque.

In all, the three-level DTC drive can restrict the start-up cur-rent effectively and establish accurate stator flux at the sametime, which benefits from the flux observation employed in thesystem.

V. SIMULATION AND EXPERIMENTAL RESULTS

To validate the effectiveness of the two DTC methods, athree-level DTC motor drive was developed and experimentalresults are presented here. The sensorless three-level DTC driveis illustrated in Fig. 9. FLC is employed in the outer speed loopfor speed control, and a speed-adaptive flux observer with loadtorque observation is used to estimate the rotor speed, statorflux, and torque. The estimated states are fed back to the outerloop of speed, flux, and torque. The system parameters are listedin Table IV. The motor inertia is 0.05 kg · m2 , and the sampling

Fig. 9. Sensorless three-level DTC drive.

TABLE IVSIMULATION AND EXPERIMENT PARAMETERS

Fig. 10. Starting response of DTC method I with preexcitation.

frequency of system for DTC method I and II are 30 and 10 kHz,respectively.

A. Simulation Results

Figs. 10 and 11 show the starting response for DTC methodI from 0 to 1200 r/min, and with and without preexcitation,respectively. In Fig. 10, the stator flux is first established by usingthe preexcitation technique, which can be seen from the upperhalf in Fig. 10. The motor then accelerates up to 1200 r/min withthe permitted maximum torque. The maximum starting currentis restricted to 7.5 A; this value reaches almost 27 A without


Fig. 11. Starting response of DTC method I without preexcitation.

Fig. 12. Starting response of DTC method II with preexcitation.

preexcitation, as shown in Fig. 11. For DTC method II, similarresults (presented in Figs. 12 and 13) can be obtained, whichproves the effectiveness of preexcitation technique. The statorflux is established with current limitation before starting themotor, so sufficient torque can be produced, which may resultbetter dynamic performance. However, the dynamic responsedifference between the one with and without preexcitation isnot significant, if the preexcitation time is excluded; this isbecause the stator flux can be established in several milliseconds,unfortunately, at the expense of large current, which can be seenin Figs. 11 and 13.

B. Hardware Implementation

An experimental prototype has been developed to implementthe sensorless three-level DTC drive, including four compo-nents: the three-level inverter, a 2.2 kW motor generator, aTMS320F2812 DSP-based control and sampling board, and a6RA70 controller, which are illustrated in Fig. 14(a)–(d), respec-tively. The TMS320F2812 is a fixed-point 32-bit DSP charac-terized by powerful computation ability, and it is very suitable

Fig. 13. Starting response of DTC method II without preexcitation.

Fig. 14. Experimental setup of three-level DTC drive. (a) Three-level inverter.(b) DSP control board. (c) 2.2 kW motor generator. (d) 6RA70 controller.

for digital control of a motor drive. The IQmath library and Clanguage were adopted for high precision and fast computation.There are many peripheral resources in the DSP, including ahigh-speed AD converter, two event managers (EVs), rich com-munication interfaces, etc. These facilitate the development ofthe motor control. A CPLD EPM7256AE was also incorporatedin the control board to implement dead time, pulse manage-ment, and to generate the final 12 triggering pulses. The DSPcontroller provides a sufficient ability for intensive computa-tions, and it only needs 3.2 μs to fulfill the computation of thespeed-adaptive observer. The 6RA70 controller is used to applythe external load and implement four-quadrant operation.

C. Experimental Results

The experimental results were obtained from a three-levelinverter-fed sensorless DTC motor drive. Various operationalcharacteristics were investigated, such as the starting response,


Fig. 15. Starting response without preexcitation for DTC method I.

Fig. 16. Starting response with preexcitation for DTC method I.

response to external disturbance, steady-state response, and low-speed operation with and without load. All internal variables,such as stator flux, torque, estimated speed, and real speed, wereviewed and recorded through a four-channel DA converter in theDSP control board. The line current and voltage was recordedusing a voltage probe and current sensor. It should be noted thatthe real speed was obtained from an encoder mounted on themotor; this was used only for data logging and comparison.

The starting response was first investigated. Figs. 15 and 16show the starting response using DTC method I over a speedrange from 0 to 1500 r/min, and with and without preexcitation.From top to bottom, the curves in Fig. 15 are the real andestimated speeds, the estimated stator flux and torque, the linevoltage, and the line current. It can be seen that if the motor startsdirectly, there is large starting current, and the peak reaches 15A. By first establishing the stator flux (preexcitation), the motorcan still start with sufficient torque, but the peak current is nowonly 9 A; hence validating the preexcitation technique. Similarresults can be found in Figs. 17 and 18 for DTC method II.

The performance under external disturbance was investigatedin the following. Full load was applied and then removed sud-denly when the motor was operating at 1500 r/min. For DTCmethod I, the results are presented in Fig. 19. It can be seen

Fig. 17. Starting response without preexcitation for DTC method II.

Fig. 18. Starting response without preexcitation for DTC method II.

Fig. 19. Response to external load disturbance for DTC method I.

that the torque responded quickly and the stator flux was keptconstant. There was only a slight drop in speed response. Thisillustrates that the system exhibits strong robustness to externaldisturbance. For DTC method II, there is greater speed drop,as shown in Fig 20. Analysis reveals the reason for this is thatthe synthesis vector amplitude is not high enough to providesufficient output torque. This can be overcome by increasing


Fig. 20. Response to external load disturbance for DTC method II.

Fig. 21. Zoomed line voltage at 1500 r/min for DTC method I.

Fig. 22. Zoomed line voltage at 1500 r/min for DTC method II.

the synthesis vector amplitudes, or the value of dc bus. A moreflexible scheme is achieved by adjusting the synthesis vectoramplitudes according to the range of speed and load condition.However, it can be seen that during the dynamic period, theestimated speed follows the real speed exactly, hence validatingthe effectiveness of the proposed observer.

Both methods can achieve smooth vector switching, as illus-trated in Fig. 21 for DTC method I and Fig. 22 for DTC method

Fig. 23. Upper and lower capacitor voltage during the deceleration and accel-eration for DTC method I.

Fig. 24. Upper and lower capacitor voltage during the deceleration and accel-eration for DTC method II.

II. It can be observed that there is no high-voltage jump in theline voltage, which proves that the proposed optimal switchingtable and the principles of the vector selection are correct. Thevoltages across the upper and lower capacitor during the motordeceleration and acceleration is presented in Figs. 23 and 24, forDTC method I and DTC method II, respectively. The capacitorvoltage variations are caused by the energy flow between themotor and the capacitor in the dynamic process. It can be seentheir values are almost the same during the dynamic and steadyprocess, which validate the effectiveness of the neutral-point-balance algorithm.

The final investigation addressed the performance under low-speed operating conditions. Fig. 25 shows the steady operationat 60 r/min for DTC method I. It can be seen that the estimatedand real speeds exactly match. The sensorless drive can workdown to 30 r/min with full load, as illustrated in Fig. 26. Thisshows excellent performance in the low-speed range. For DTCmethod II, the sensorless drive can work at even lower speed.Figs. 27 and 28 illustrate that the system can operate down to 6r/min without load and 30 r/min with 80% rated load. It shouldbe noted that the ripple in the real speed shown in Fig. 27 wasdue to the speed measurement method, where there was a limited


Fig. 25. Steady state of 60 r/min without load for DTC method I.

Fig. 26. Steady state of 30 r/min with full load for DTC method I.

Fig. 27. Steady state of 6 r/min without load for DTC method II.

pulse number from the shaft encoder. This was especially trueat low speed, and this gave an error in the recorded actual speed.The motor operated at a steady speed in the experimental test.This was confirmed by the waveform of current and estimatedspeed. It can be seen that there are only slight oscillations in theestimated speed, and the stator flux is smooth and steady.

Fig. 28. Steady state of 30 r/min with 80% rated load for DTC method II.

D. Performance Analysis

The two DTC methods have different switching character-istics, so it is not easy to compare the switching frequencythoroughly and fairly. The first DTC method has variable andrelatively low-switching frequency under the same sampling fre-quency, compared to the second one, because only one vector isselected during one sampling period. For DTC method II, thereare 16 leg state changes for the whole 12 switches during onesampling period, so the mean switching frequency is constant,i.e., 2/3 of the sampling frequency. For example, at the steadystate of 1500 r/min without load, the mean switching frequencyfor DTC method I and II are 2.1 and 6.67 kHz, respectively, soDTC method I has less switching loss. However, lots of logi-cal judgments are needed in DTC method I to obtain the finaloptimized vector, because it has to consider the effects of neu-tral point balance and smooth vector switching. This problem issimplified in the second DTC method with the help of DSVM,so the control of torque and flux is decoupled from the controlof neutral point balance and vector switching, resulting betterlow-speed performance (as low as 6 r/min shown in Fig. 27).

VI. CONCLUSION

Two kinds of modified DTC schemes have been proposed inthis paper and both achieve high-performance sensorless con-trol of a three-level inverter-fed motor drive. They both workover a wide speed range and overcome the limitations causedby the topology of the three-level inverter. By using appropri-ate intermediate vectors, the problems of neutral point balanceand smooth vector switching are solved. Furthermore, a novelvector synthesis sequence was proposed and this decoupled theperformance control from the circuit limitation. A FLC and aspeed-adaptive flux observer were incorporated in the sensor-less drive to enhance the performance of system. In addition,the issue of large starting current are investigated and solvedby introducing the technique of preexcitation. Very low speedsensorless operation of 6 r/min was demonstrated. Simulationsas well as experimental results were presented to verify theeffectiveness of the proposed schemes.


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Yongchang Zhang (M’10) received the B.S. degreefrom Chongqing University, Chongqing, China, in2004, and the Ph.D. degree from Tsinghua University,Beijing, China, in 2009, respectively, both in electri-cal engineering.

He was a Postdoctoral Fellow with the Uni-versity of Technology Sydney, Sydney, Australia.Since August 2011, he has been a faculty memberof North China University of Technology, Beijing,China. His research interests include sensorless andhigh-performance control of ac motor drives, control

of multilevel converters, optimal pulsewidth modulation, pulsewidth modula-tion rectifier, and advanced digital control with realtime implementation.

Jianguo Zhu (S’93–M’96–SM’03) received the B.E.degree from Jiangsu Institute of Technology, Jiangsu,China, in 1982, the M.E. degree from Shanghai Uni-versity of Technology, Shanghai, China, in 1987,and the Ph.D degree from University of Technology,Sydney (UTS), Sydney, Australia, in 1995.

He is currently a Professor of electrical engineer-ing and Head for School of Electrical, Mechanical,and Mechatronic Systems, UTS. His research inter-ests include electromagnetics, magnetic properties ofmaterials, electrical machines and drives, power elec-

tronics, and renewable energy systems.

Zhengming Zhao (M’02–SM’03) received the B.Sc.and M.Sc. degrees, both in electrical engineering,from Hunan University, Hunan, China, in 1982and 1985, respectively, and the Ph.D. degree fromTsinghua University, Beijing, China, in 1991.

In 1993, he was in the Department of Electri-cal Engineering, Tsinghua University as an Asso-ciate Professor in 1993 and a Full Professor in 1999.From 1994 to 1996, he was a Postdoctoral Fellow atthe Ohio State University, Columbus, OH, and then,joined in the University of California at Irvine as a

Visiting Scholar for one year. In 1998, he joined as a Visiting Scholar for threemonths at the University of British Columbia, BC, Canada and in 1999, hewas invited to Hong Kong University as a Research Assistant Professor foranother three months. He is currently the Deputy Chairman of the ElectricalEngineering Department, Tsinghua University and a Vice Director of the Na-tional Key Laboratory on power system. His research interests include powerelectronics, motor design and drive, integration of drive system, and solar energyapplications.


Wei Xu (M’09) received the B.E. and M.E. degreesfrom Tianjin University, Tianjin, China, in 2002 and2005, and the Ph.D. degree from the Institute of Elec-trical Engineering, Chinese Academy of Sciences,Beijing, China, in 2008, respectively, all in electricalengineering.

He is currently a Postdoctoral Fellow in the Centerfor Electric Machine and Power Electronics, Univer-sity of Technology Sydney (UTS), Sydney, Australia.His research interests include electromagnetic designand performance analysis of linear machine, perma-

nent magnet machine, and switched reluctance machine.

David G. Dorrell (M’95–SM’08) received theB.Eng. (Hons.) degree in electrical and electronic en-gineering from the University of Leeds, Leeds, U.K.,in 1988, the M.Sc. degree from the University ofBradford, Bradford, U.K., in power electronics en-gineering, in 1989, and the Ph.D. degree from theUniversity of Cambridge, Cambridge, U.K., in engi-neering, in 1993.

He was a Lecturer with the Robert Gordon Univer-sity and the University of Reading. He was a SeniorLecturer with the University of Glasgow, U.K. for

several years. He is currently an Associate Professor with the University ofTechnology Sydney, Sydney, Australia. He is also an Adjunct Associate Pro-fessor with the National Cheng Kung University, Tainan City, Taiwan. He hasauthored or coauthored more than 100 technical publications. His research in-terests include the design and analysis of various electrical machines and alsorenewable energy systems.

Dr. Dorrell is a Charter Engineer in U.K. and a Fellow of the Institution ofEngineering and Technology.

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