2252 ieee transactions on communications, vol. 58, โ€ฆhxm025000/pilotiqtcom2010aug.pdfminn and...

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2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010 Pilot Designs for Channel Estimation of MIMO OFDM Systems with Frequency-Dependent I/Q Imbalances Hlaing Minn, Senior Member, IEEE, and Daniel Munoz, Student Member, IEEE Abstractโ€”Multiple input multiple output (MIMO) orthogonal frequency division multiplexing (OFDM) systems facilitate high data rate wireless communications, and require reliable channel estimates to fully materialize their advantages. The semicon- ductor downscaling trend has exacerbated device impairments such as inphase and quadrature (I/Q) imbalances which cause inter-carrier interferences in OFDM systems which cannot be remedied by increasing signal power. Different RF chains of MIMO branches can cause different I/Q imbalances which further complicates MIMO OFDM channel estimation. This paper proposes several pilot designs for the estimation of the combined responses of MIMO frequency-selective channels and frequency-dependent I/Q imbalances. The proposed designs re- quire much smaller pilot overhead than the existing designs, and also provide estimation mean-squares error optimality (under white noise) and general applicability to preamble as well as pilot-data-multiplexed symbols in MIMO systems with or without null guard tones. Performance analyses and simulation results corroborate advantages of the proposed designs. Index Termsโ€”Channel estimation, I/Q imbalance, MIMO, OFDM, pilot design. I. I NTRODUCTION T HE advances in semiconductor down-scaling have en- abled a substantial growth of signal processing power but exacerbated the variations of the device characteristics due to more dif๏ฌcult control in the fabrication process [1]. The inphase (I) and quadrature (Q) imbalance, which repre- sents mismatch between the I and Q branches, is a major impairment especially for direct-conversion transceivers with a high modulation order [2]. The higher data rate of next- generation wireless systems in a typical bandwidth limited regime requires -ary quadrature amplitude modulation ( - QAM) with a large . Both the semiconductor down-scaling trend and the need of a high modulation order underline the importance of I/Q imbalance issue for next generation wireless systems. A viable transmission technology for current and next gen- eration wireless systems is OFDM, while MIMO systems have Paper approved by G. Bauch, the Editor for MIMO, Coding and Relaying of the IEEE Communications Society. Manuscript received March 12, 2009; revised July 23, 2009 and November 9, 2009. The result for SISO systems was presented at IEEE WCNC 2009, the [TDM; Null] design at IEEE VTC 2009 (Fall), and [CDM-F; Null], [FDM; Null], [CDM-T; C-T] without Self-Mirror Tones, and [CDM-F or FDM; C-T] designs at IEEE Globecom 2009. The authors are with the Dept. of Electrical Engineering, University of Texas at Dallas, Richardson, TX 75083 USA (e-mail: {hlaing.minn, djm072000}@utdallas.edu). Digital Object Identi๏ฌer 10.1109/TCOMM.2010.08.090150 become instrumental in providing higher data rate and better reliability in diverse wireless channels. The I/Q impairment in OFDM causes inter-carrier interferences. In MIMO OFDM, due to different RF chains for different antennas, I/Q imbal- ances may differ across antennas as well as across frequency (especially for ultra-wideband systems) for each antenna. For data detection, the multitudes of I/Q imbalance and channel parameters or their combined responses need to be estimated and compensated (see [3]โ€“[9] for representative I/Q imbalance compensation methods). An ef๏ฌcient and practical way for the estimation is to transmit pilot tones. The main issue is how to design these pilots while considering the interferences among mirror tones and different transmit antennas as well as the estimation performance and overhead. The pilot designs may depend on whether they are developed for separate estimation of I/Q imbalances and channel responses or estimation of the combined responses. We consider the latter. There are several pilot or training signal designs for MIMO OFDM channel estimation without I/Q imbalance (e.g., [10]โ€“ [14]). There also exist several pilot designs for I/Q imbalance estimation. The work in [15] applies two OFDM training symbols to perform per-subcarrier estimation of frequency- dependent (FD) I/Q imbalance in single input single output (SISO) systems. The ๏ฌrst training symbol has null tones at all negative subcarrier indexes and the second symbol contains null tones at all positive subcarrier indexes. This method neither optimizes the pilot overhead (large overhead) nor considers pilot-data-multiplexed symbols, MIMO OFDM, and exploitation of frequency-domain correlation. Similar draw- backs apply to [16] except it considers (mainly frequency- independent (FI)) transmitter I/Q imbalance only, and uses a different pilot design. It uses an even number of OFDM training symbols with non-zero pilots where the pilots at the negative (positive) subcarrier indexes of the even symbols are the same as (negatives of) the corresponding pilots at the odd training symbols. The same pilot design using two OFDM training symbols is applied in [17] for the FI receiver I/Q imbalance, but [17] requires an additional training symbol for channel estimation, resulting in a larger overhead. It considers pilot-data-multiplexed symbols, but the other drawbacks still hold. [18] applies a pilot design (a combination of the designs in [15] and [16]) which uses an even number of OFDM training symbols with null pilots at the negative subcarrier indexes of the ๏ฌrst half of OFDM training symbols and at the positive subcarrier indexes of the second half of OFDM 0090-6778/10$25.00 c โƒ 2010 IEEE

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Page 1: 2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, โ€ฆhxm025000/PilotIQTCOM2010Aug.pdfMINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT

2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

Pilot Designs for Channel Estimation ofMIMO OFDM Systems with

Frequency-Dependent I/Q ImbalancesHlaing Minn, Senior Member, IEEE, and Daniel Munoz, Student Member, IEEE

Abstractโ€”Multiple input multiple output (MIMO) orthogonalfrequency division multiplexing (OFDM) systems facilitate highdata rate wireless communications, and require reliable channelestimates to fully materialize their advantages. The semicon-ductor downscaling trend has exacerbated device impairmentssuch as inphase and quadrature (I/Q) imbalances which causeinter-carrier interferences in OFDM systems which cannot beremedied by increasing signal power. Different RF chains ofMIMO branches can cause different I/Q imbalances whichfurther complicates MIMO OFDM channel estimation. Thispaper proposes several pilot designs for the estimation of thecombined responses of MIMO frequency-selective channels andfrequency-dependent I/Q imbalances. The proposed designs re-quire much smaller pilot overhead than the existing designs, andalso provide estimation mean-squares error optimality (underwhite noise) and general applicability to preamble as well aspilot-data-multiplexed symbols in MIMO systems with or withoutnull guard tones. Performance analyses and simulation resultscorroborate advantages of the proposed designs.

Index Termsโ€”Channel estimation, I/Q imbalance, MIMO,OFDM, pilot design.

I. INTRODUCTION

THE advances in semiconductor down-scaling have en-abled a substantial growth of signal processing power

but exacerbated the variations of the device characteristicsdue to more difficult control in the fabrication process [1].The inphase (I) and quadrature (Q) imbalance, which repre-sents mismatch between the I and Q branches, is a majorimpairment especially for direct-conversion transceivers witha high modulation order [2]. The higher data rate of next-generation wireless systems in a typical bandwidth limitedregime requires ๐‘€ -ary quadrature amplitude modulation (๐‘€ -QAM) with a large ๐‘€ . Both the semiconductor down-scalingtrend and the need of a high modulation order underline theimportance of I/Q imbalance issue for next generation wirelesssystems.

A viable transmission technology for current and next gen-eration wireless systems is OFDM, while MIMO systems have

Paper approved by G. Bauch, the Editor for MIMO, Coding and Relayingof the IEEE Communications Society. Manuscript received March 12, 2009;revised July 23, 2009 and November 9, 2009.

The result for SISO systems was presented at IEEE WCNC 2009, the[TDM; Null] design at IEEE VTC 2009 (Fall), and [CDM-F; Null], [FDM;Null], [CDM-T; C-T] without Self-Mirror Tones, and [CDM-F or FDM; C-T]designs at IEEE Globecom 2009.

The authors are with the Dept. of Electrical Engineering, Universityof Texas at Dallas, Richardson, TX 75083 USA (e-mail: {hlaing.minn,djm072000}@utdallas.edu).

Digital Object Identifier 10.1109/TCOMM.2010.08.090150

become instrumental in providing higher data rate and betterreliability in diverse wireless channels. The I/Q impairment inOFDM causes inter-carrier interferences. In MIMO OFDM,due to different RF chains for different antennas, I/Q imbal-ances may differ across antennas as well as across frequency(especially for ultra-wideband systems) for each antenna. Fordata detection, the multitudes of I/Q imbalance and channelparameters or their combined responses need to be estimatedand compensated (see [3]โ€“[9] for representative I/Q imbalancecompensation methods). An efficient and practical way for theestimation is to transmit pilot tones. The main issue is how todesign these pilots while considering the interferences amongmirror tones and different transmit antennas as well as theestimation performance and overhead. The pilot designs maydepend on whether they are developed for separate estimationof I/Q imbalances and channel responses or estimation of thecombined responses. We consider the latter.

There are several pilot or training signal designs for MIMOOFDM channel estimation without I/Q imbalance (e.g., [10]โ€“[14]). There also exist several pilot designs for I/Q imbalanceestimation. The work in [15] applies two OFDM trainingsymbols to perform per-subcarrier estimation of frequency-dependent (FD) I/Q imbalance in single input single output(SISO) systems. The first training symbol has null tones at allnegative subcarrier indexes and the second symbol containsnull tones at all positive subcarrier indexes. This methodneither optimizes the pilot overhead (large overhead) norconsiders pilot-data-multiplexed symbols, MIMO OFDM, andexploitation of frequency-domain correlation. Similar draw-backs apply to [16] except it considers (mainly frequency-independent (FI)) transmitter I/Q imbalance only, and usesa different pilot design. It uses an even number of OFDMtraining symbols with non-zero pilots where the pilots at thenegative (positive) subcarrier indexes of the even symbols arethe same as (negatives of) the corresponding pilots at the oddtraining symbols. The same pilot design using two OFDMtraining symbols is applied in [17] for the FI receiver I/Qimbalance, but [17] requires an additional training symbol forchannel estimation, resulting in a larger overhead. It considerspilot-data-multiplexed symbols, but the other drawbacks stillhold. [18] applies a pilot design (a combination of the designsin [15] and [16]) which uses an even number of OFDMtraining symbols with null pilots at the negative subcarrierindexes of the first half of OFDM training symbols and atthe positive subcarrier indexes of the second half of OFDM

0090-6778/10$25.00 cโƒ 2010 IEEE

Page 2: 2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, โ€ฆhxm025000/PilotIQTCOM2010Aug.pdfMINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT

MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2253

training symbols. It considers FI receiver I/Q imbalance only,and is also associated with the above drawbacks.

For estimating the frequency-domain combined responsesof the channel and FD I/Q imbalances, [19] presents two pilotdesigns using two OFDM training symbols. In the first design,the second OFDM training symbol is the 90 degree phaserotated version of the first symbol. The second design is thesame as the one in [16]. It also provides a design conditionfor multiple OFDM symbols. But it uses a large overhead,addresses neither pilot-data-multiplexed symbols nor MIMOOFDM, and does not exploit frequency-domain correlation.

In [20], the authors consider a time-domain estimationwhich exploits frequency-domain correlation. It transmitsequal-amplitude pilots on all subcarriers of one OFDM sym-bol. The mirror-tone interference is cancelled by designing thesum of the phases of the ๐‘žth pilot and its mirror tone to be ๐‘ž๐œ‹.In this design the phases of the pilots at tone indexes 0 and๐‘/2 cannot be arbitrary as opposed to the statement in [20].Moreover, when null guard tones are added at the band edgesas in [21], the estimation optimality is no longer maintained.It does not address for pilot-data-multiplexed symbols andMIMO OFDM systems, and uses a large overhead. The pilotdesign in [22] uses all subcarriers of an OFDM symbol forpilots (large overhead), and is not applicable to systems witha pilot-data multiplexed format and/or with null guard tones.It only considers FI I/Q imbalance and requires a two stepapproach of estimation of I/Q parameters and compensation.

All of the above pilot designs for I/Q imbalance addressfor SISO OFDM systems only. Recently, [23] proposes a pilotdesign for the transmitter and receiver FI I/Q imbalances inMIMO-OFDM systems. Its extension to the FD I/Q imbalanceis considered in [24]. Their pilot design uses the design in [16]with two OFDM training symbols as a basic block which isrepeated according to Hadamard sequences for multiple trans-mit antennas. For ๐‘Tx transmit antennas, they require 2๐‘Tx

OFDM training symbols, and hence a large pilot overhead.Moreover, they are not applicable to pilot-data-multiplexedsymbols, and do not provide the estimation optimality sincethe frequency-domain correlation is not exploited. A pilotdesign for space-time block coded systems in single-carrierfrequency-flat fading channels is recently proposed in [25].But it uses a large pilot overhead (at least two space-timecode blocks), lacks the estimation optimality, and cannot beapplied to systems with more than two transmit antennas.

In brief, the existing pilot designs for MIMO OFDM chan-nel estimation do not consider the I/Q imbalance effects, whilethe existing pilot designs for the I/Q imbalance estimationdo not address the channel estimation optimality. All of thelatter designs require large pilot overheads, and most ofthem cannot be applied to MIMO systems and pilot-data-multiplexed symbols. In contrast, in this paper, we proposeseveral pilot designs for MIMO OFDM equivalent channelestimation in the presence of transmitter and receiver FDI/Q imbalances, which avoid the drawbacks of the existingmethods. Our designs require much less pilot overhead, andare more general and white-noise optimal in estimation.

The paper is organized as follows. Section II describes thesignal model and Section III presents pilot design criteria. InSection IV, we study the relationships of the design criteria to

)(tw

Equivalent Lowpass SISO System

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Bandpass SISO System

Channel)(th

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)(tgDT

)(tgMT( )*

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Fig. 1. Bandpass and equivalent lowpass SISO system models.

pilot symbols and structures, and propose several pilot designs.Performance evaluation and simulation results are discussed inSection V, and conclusions are provided in Section VI.

Notation: A bold capital (small) letter represents a matrix(a column vector). The superscripts โˆ—, ๐‘‡ , and ๐ป representthe conjugate, the transpose, and the conjugate transposeoperations, respectively. [๐‘ฟ ]๐‘˜,๐‘š denotes the ๐‘˜-th row, ๐‘š-thcolumn element of ๐‘ฟ . All indices start from 0. The all-one (all-zero) square matrix of size-๐‘˜, the ๐‘˜ ร— ๐‘š all-zeromatrix, and the ๐‘˜ร— ๐‘˜ identity matrix are denoted by 1๐‘˜ (0๐‘˜),0๐‘˜ร—๐‘š, and ๐‘ฐ๐‘˜, respectively. Tr[๐‘ฟ] denotes the trace of ๐‘ฟ .diag{๐’™} represents a diagonal matrix whose diagonal elementsare defined by ๐’™. The ๐‘™-cyclic-forward-shifted version of ๐’„is denoted by ๐’„(๐‘™). * and โŠ— denote the convolution and theKronecker product, respectively. (๐‘™)๐‘ represents ๐‘™ modulo ๐‘ .โŒŠ๐‘‹โŒ‹ denotes the integer part of ๐‘‹ while โŒˆ๐‘‹โŒ‰ represents thesmallest integer greater than or equal to ๐‘‹ . ๐‘ญ denotes the๐‘ -point unitary discrete Fourier transform matrix and ๐’‡๐‘˜ isthe ๐‘˜-th column of ๐‘ญ . The subcarrier indexes are modulo ๐‘where ๐‘ is the total number of OFDM subcarriers. We willoften use โˆ’๐‘˜ to denote the subcarrier index (โˆ’๐‘˜)๐‘ , and & torepresent the logical โ€œandโ€ operation.

II. SIGNAL MODEL

First, consider a single antenna system with FI and FD I/Qimbalances at both transmitter and receiver sides as shown inFig. 1 where {๐‘Ž๐ผ๐‘ก , ๐‘Ž๐‘„๐‘ก } and {๐œƒ๐ผ๐‘ก , ๐œƒ๐‘„๐‘ก } represent the FI gainsand phase offsets of the I and Q branches at the transmitter(see [1]). The equivalent pulse shaping filters (i.e., the overallimpulse responses including DAC, amplifier, pulse shaping,and FD I/Q imbalances) for the I and Q branches of thetransmitter are denoted by ๐‘”๐ผ๐‘ก (๐‘ก) and ๐‘”๐‘„๐‘ก (๐‘ก). These filtersinclude effects of the FD I/Q imbalances [4], [21]. The receiverside parameters are defined similarly with the subscript ๐‘กreplaced by ๐‘Ÿ. An equivalent low-pass system is also presentedin Fig. 1. The channel impulse response โ„Ž(๐‘ก) is the low-pass-equivalent of the bandpass channel โ„Žbp(๐‘ก). The transmitsystem with I/Q imbalance can be viewed as the summationof two systems namely the direct system whose input is thesame as the transmitter input signal ๐‘ (๐‘ก) = ๐‘ ๐ผ(๐‘ก)+ ๐‘—๐‘ ๐‘„(๐‘ก) andthe mirror system whose input is ๐‘ โˆ—(๐‘ก). The impulse responsesof the direct and mirror systems at the transmitter are denotedby ๐‘”๐ท๐‘‡ (๐‘ก) and ๐‘”๐‘€๐‘‡ (๐‘ก), and those at the receiver are represented

Page 3: 2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, โ€ฆhxm025000/PilotIQTCOM2010Aug.pdfMINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT

2254 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

][krj

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)(, tgD jR)(tw

Fig. 2. A simplified equivalent lowpass MIMO system model.

by ๐‘”๐ท๐‘… (๐‘ก) and ๐‘”๐‘€๐‘… (๐‘ก). Then, they can be given as

๐‘”๐ท๐‘‡ (๐‘ก) =1

2[๐‘Ž๐ผ๐‘ก ๐‘’

๐‘—๐œƒ๐ผ๐‘ก ๐‘”๐ผ๐‘ก (๐‘ก) + ๐‘Ž๐‘„๐‘ก ๐‘’

๐‘—๐œƒ๐‘„๐‘ก ๐‘”๐‘„๐‘ก (๐‘ก)] (1)

๐‘”๐‘€๐‘‡ (๐‘ก) =1

2[๐‘Ž๐ผ๐‘ก ๐‘’

๐‘—๐œƒ๐ผ๐‘ก ๐‘”๐ผ๐‘ก (๐‘ก)โˆ’ ๐‘Ž๐‘„๐‘ก ๐‘’

๐‘—๐œƒ๐‘„๐‘ก ๐‘”๐‘„๐‘ก (๐‘ก)] (2)

๐‘”๐ท๐‘… (๐‘ก) =1

2[๐‘Ž๐ผ๐‘Ÿ๐‘’

โˆ’๐‘—๐œƒ๐ผ๐‘Ÿ๐‘”๐ผ๐‘Ÿ (๐‘ก) + ๐‘Ž๐‘„๐‘Ÿ ๐‘’

โˆ’๐‘—๐œƒ๐‘„๐‘Ÿ ๐‘”๐‘„๐‘Ÿ (๐‘ก)] (3)

๐‘”๐‘€๐‘… (๐‘ก) =1

2[๐‘Ž๐ผ๐‘Ÿ๐‘’

๐‘—๐œƒ๐ผ๐‘Ÿ๐‘”๐ผ๐‘Ÿ (๐‘ก)โˆ’ ๐‘Ž๐‘„๐‘Ÿ ๐‘’

๐‘—๐œƒ๐‘„๐‘Ÿ ๐‘”๐‘„๐‘Ÿ (๐‘ก)]. (4)

In MIMO systems, different RF chains for different antennasmay give rise to different I/Q imbalances. The above responsescorresponding to the ๐‘–th transmit antenna and the ๐‘—th receiveantenna are denoted by ๐‘”๐ท๐‘‡,๐‘–(๐‘ก), ๐‘”

๐‘€๐‘‡,๐‘–(๐‘ก), ๐‘”

๐ท๐‘…,๐‘—(๐‘ก), and ๐‘”๐‘€๐‘…,๐‘—(๐‘ก),

respectively. Then, a simplified low-pass-equivalent signalmodel for MIMO can be obtained as in Fig. 2 for the ๐‘—threceive antenna where the impulse responses of the overalldirect channel ๐‘๐‘–๐‘—(๐‘ก) and the overall mirror channel ๐‘ž๐‘–๐‘—(๐‘ก) forthe ๐‘–th transmit antenna and the ๐‘—th receive antenna read as

๐‘๐‘–๐‘—(๐‘ก) = ๐‘”๐ท๐‘‡,๐‘–(๐‘ก) โˆ— โ„Ž๐‘–๐‘—(๐‘ก) โˆ— ๐‘”๐ท๐‘…,๐‘—(๐‘ก)

+ (๐‘”๐‘€๐‘‡,๐‘–(๐‘ก))โˆ— โˆ— โ„Žโˆ—๐‘–๐‘—(๐‘ก) โˆ— ๐‘”๐‘€๐‘…,๐‘—(๐‘ก) (5)

๐‘ž๐‘–๐‘—(๐‘ก) = ๐‘”๐‘€๐‘‡,๐‘–(๐‘ก) โˆ— โ„Ž๐‘–๐‘—(๐‘ก) โˆ— ๐‘”๐ท๐‘…,๐‘—(๐‘ก)

+ (๐‘”๐ท๐‘‡,๐‘–(๐‘ก))โˆ— โˆ— โ„Žโˆ—๐‘–๐‘—(๐‘ก) โˆ— ๐‘”๐‘€๐‘…,๐‘—(๐‘ก). (6)

The receive filter output signal corresponding to the receiveantenna ๐‘— and transmit signals {๐‘ ๐‘–(๐‘ก)} is

๐‘Ÿ๐‘—(๐‘ก) =

๐‘Txโˆ’1โˆ‘๐‘–=0

(๐‘ ๐‘–(๐‘ก) โˆ— ๐‘๐‘–(๐‘ก) + ๐‘ โˆ—๐‘– (๐‘ก) โˆ— ๐‘ž๐‘–(๐‘ก)) + ๐‘›๐‘—(๐‘ก) (7)

where the complex Gaussian noise ๐‘›๐‘—(๐‘ก) is given by

๐‘›๐‘—(๐‘ก) = ๐‘ค(๐‘ก) โˆ— ๐‘”๐ท๐‘…,๐‘—(๐‘ก) + ๐‘คโˆ—(๐‘ก) โˆ— ๐‘”๐‘€๐‘…,๐‘—(๐‘ก). (8)

When MIMO channels are independent or their joint statis-tics are unknown, the logical approach to the estimation ofthe equivalent direct and mirror channels is to estimate themat each receive antenna independently. Hence, we just needto describe them for one receive antenna. In the following,we drop the receive antenna index ๐‘—. We consider an OFDMsystem with a cyclic prefix (CP) interval (๐‘CP samples) longerthan the maximum span (๐ฟ samples) of {๐‘๐‘–(๐‘ก)} and {๐‘ž๐‘–(๐‘ก)}.

Now, consider a low-pass-equivalent discrete-time OFDMsystem with ๐‘ subcarriers. The channels are assumed to be

constant over ๐พ symbols. The discrete-time transmit trainingsignal from the ๐‘–th transmit antenna during the ๐‘™th symbol isdenoted by ๐‘ (๐‘™)๐‘– [๐‘˜] with the integer index ๐‘˜ โˆˆ [โˆ’๐‘CP, ๐‘ โˆ’ 1],and the CP samples ๐‘ 

(๐‘™)๐‘– [๐‘š] = ๐‘ 

(๐‘™)๐‘– [๐‘ โˆ’ ๐‘š] for ๐‘š โˆˆ

[โˆ’๐‘CP,โˆ’1]. Similar notations apply to the data signal ๐‘ฅ(๐‘™)๐‘– [๐‘˜].The discrete-time versions of ๐‘๐‘–(๐‘ก) and ๐‘ž๐‘–(๐‘ก) are denoted by๐ฟร—1 vectors ๐’‘๐‘– and ๐’’๐‘–, respectively. The time-domain ๐‘ ร—1received signal vector after the CP removal at each receiveantenna for the ๐‘™th OFDM symbol, denoted by ๐’“๐‘™, can beexpressed in a general pilot-data multiplexed setup (whichincludes pilot only or data only symbols as special cases) as

๐’“๐‘™ =

๐‘Txโˆ’1โˆ‘๐‘–=0

{(๐‘บ๐‘–[๐‘™] +๐‘ฟ๐‘–[๐‘™])๐’‘๐‘– + (๐‘บโˆ—๐‘– [๐‘™] +๐‘ฟโˆ—

๐‘– [๐‘™])๐’’๐‘–}+ ๐’๐‘™

(9)

where (๐‘š, ๐‘˜)th element of the ๐‘ ร— ๐ฟ signal matrix ๐‘บ๐‘–[๐‘™] (or๐‘ฟ๐‘–[๐‘™]) is ๐‘ (๐‘™)๐‘– [๐‘š โˆ’ ๐‘˜] (or ๐‘ฅ(๐‘™)๐‘– [๐‘š โˆ’ ๐‘˜]) with ๐‘š โˆˆ [0, ๐‘ โˆ’ 1]and ๐‘˜ โˆˆ [0, ๐ฟ โˆ’ 1]. The received signal vector for ๐พ OFDMsymbols is given by

๐’“ = ๐‘บ๐’‘+ ๐‘บโˆ—๐’’ +๐‘ฟ๐’‘+๐‘ฟโˆ—๐’’ + ๐’ (10)

where ๐’“ = [๐’“๐‘‡0 ๐’“๐‘‡1 . . . ๐’“

๐‘‡๐พโˆ’1]

๐‘‡ , ๐’‘ = [๐’‘๐‘‡0 ๐’‘

๐‘‡1 . . .๐’‘

๐‘‡๐‘Txโˆ’1

]๐‘‡ ,๐’’ = [๐’’๐‘‡0 ๐’’

๐‘‡1 . . .๐’’

๐‘‡๐‘Txโˆ’1

]๐‘‡ , ๐’ = [๐’๐‘‡0 ๐’

๐‘‡1 . . .๐’

๐‘‡๐พโˆ’1]

๐‘‡ , the(๐‘˜, ๐‘–)th submatrice of ๐‘บ and ๐‘ฟ are respectively given by ๐‘บ๐‘–[๐‘˜]and ๐‘ฟ๐‘–[๐‘˜], with ๐‘˜ โˆˆ [0,๐พ โˆ’ 1] and ๐‘– โˆˆ [0, ๐‘Tx โˆ’ 1]. Thecomplex Gaussian noise vectors {๐’๐‘™} are given by

๐’๐‘™ = ๐‘ฎ๐‘…,๐ท๐’˜๐‘™ +๐‘ฎ๐‘…,๐‘€๐’˜โˆ—๐‘™ (11)

where {๐’˜๐‘™} are independent and identically distributed (i.i.d.)random vectors, each consisting of i.i.d. circularly-symmetriccomplex Gaussian random variables with the variance ๐œŽ2๐‘ค. Let๐œ† denote the maximum of the numbers of taps of ๐‘”๐ท๐‘… [๐‘˜] and๐‘”๐‘„๐‘… [๐‘˜]. Then, ๐‘ฎ๐‘…,๐ท and ๐‘ฎ๐‘…,๐‘€ are ๐‘ร—(๐‘+๐œ†) matrices withtheir first rows given by [๐‘”๐ท๐‘… [0], ๐‘”

๐ท๐‘… [1], . . . , ๐‘”

๐ท๐‘… [๐œ†], 01ร—๐‘โˆ’1]

and [๐‘”๐‘€๐‘… [0], ๐‘”๐‘€๐‘… [1], . . . , ๐‘”

๐‘€๐‘… [๐œ†], 01ร—๐‘โˆ’1], respectively, and

their next ๐‘˜th rows are cyclically ๐‘˜-right-shift versions of theircorresponding first rows.

Let ๐‘๐‘–,๐‘™[๐‘˜] and ๐‘๐‘–,๐‘™[๐‘˜], respectively, denote the pilot sym-bol and the data symbol on the ๐‘˜th subcarrier of the ๐‘™thOFDM symbol from the ๐‘–th transmit antenna. Define ฮ›๐‘–,๐‘™ โ‰œdiag{๐‘๐‘–,๐‘™[0], ๐‘๐‘–,๐‘™[1], . . . , ๐‘๐‘–,๐‘™[๐‘ โˆ’ 1]} and ๐‘ซ๐‘–,๐‘™ โ‰œ diag{๐‘๐‘–,๐‘™[0],๐‘๐‘–,๐‘™[1], . . . , ๐‘๐‘–,๐‘™[๐‘ โˆ’ 1]} . Let ๐‘ญ๐ฟ denote the first ๐ฟ columnsof the ๐‘ร—๐‘ unitary discrete Fourier transform (DFT) matrix๐‘ญ . Then the time domain signal matrices can be given as

๐‘บ๐‘–[๐‘™] =โˆš๐‘๐‘ญ๐ปฮ›๐‘–,๐‘™๐‘ญ ๐ฟ, ๐‘ฟ๐‘–[๐‘™] =

โˆš๐‘๐‘ญ๐ป๐‘ซ๐‘–,๐‘™๐‘ญ ๐ฟ. (12)

Define ๐‘ท โ‰œ ๐‘ญ๐‘ญ ๐‘‡ . Then, we have ๐‘ท = ๐‘ท ๐‘‡ , ๐‘ท๐‘ท ๐‘‡ = ๐‘ฐ๐‘ ,๐‘ท๐‘ญ โˆ—

๐ฟ = ๐‘ญ ๐ฟ, and

ฮ›ฬƒ๐‘–,๐‘™ โ‰œ ๐‘ทฮ›๐ป๐‘–,๐‘™๐‘ท (13)

= diag{๐‘โˆ—๐‘–,๐‘™[0], ๐‘โˆ—๐‘–,๐‘™[๐‘ โˆ’ 1], ๐‘โˆ—๐‘–,๐‘™[๐‘ โˆ’ 2], . . . , ๐‘โˆ—๐‘–,๐‘™[2], ๐‘โˆ—๐‘–,๐‘™[1]}๏ฟฝฬƒ๏ฟฝ๐‘–,๐‘™ โ‰œ ๐‘ท๐‘ซ๐ป

๐‘–,๐‘™๐‘ท (14)

= diag{๐‘โˆ—๐‘–,๐‘™[0], ๐‘โˆ—๐‘–,๐‘™[๐‘ โˆ’ 1], ๐‘โˆ—๐‘–,๐‘™[๐‘ โˆ’ 2], . . . , ๐‘โˆ—๐‘–,๐‘™[2], ๐‘โˆ—๐‘–,๐‘™[1]}.

From the above and the DFT of (9), the received symbol

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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2255

๐‘…๐‘™[๐‘˜] on the ๐‘˜th subcarrier of the ๐‘™th OFDM symbol reads as

๐‘…๐‘™[๐‘˜] =

๐‘Txโˆ’1โˆ‘๐‘–=0

{(๐‘๐‘–,๐‘™[๐‘˜] + ๐‘๐‘–,๐‘™[๐‘˜])๐‘ƒ๐‘–[๐‘˜]

+(๐‘โˆ—๐‘–,๐‘™[โˆ’๐‘˜] + ๐‘โˆ—๐‘–,๐‘™[โˆ’๐‘˜]

)๐‘„๐‘–[๐‘˜]

}(15)

where ๐‘ƒ๐‘–[๐‘˜] and ๐‘„๐‘–[๐‘˜] are the ๐‘˜th elements ofโˆš๐‘๐‘ญ ๐ฟ๐’‘๐‘– andโˆš

๐‘๐‘ญ๐ฟ๐’’๐‘–, respectively. Clearly, the I/Q imbalance introducesto the received ๐‘˜th subcarrier symbol an interference from themirror tone (i.e., index โˆ’๐‘˜) with a FD coefficient of ๐‘„๐‘–[๐‘˜].This interference, if not properly taken into account, can causea significant performance degradation.

III. MIMO OFDM PILOT DESIGN CRITERIA

For a coherent detection, the direct channel ๐’‘ and the mirrorchannel ๐’’ need to be estimated at the receiver. In practicalsystems, the statistics of the channel and the transceiverimperfections are unknown and they can be non-stationary aswell. This leads to the practical choice of least-squares typechannel estimators as considered in this paper. The estimatesof the direct and mirror CIR vectors are given by

๏ฟฝฬ‚๏ฟฝ =(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐’“ (16)

๏ฟฝฬ‚๏ฟฝ =(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡ ๐’“. (17)

Our pilot designs (and the optimality criterion) will be basedon minimizing the channel estimation mean-square error(MSE). Substituting (10) into (16) and (17) gives

๏ฟฝฬ‚๏ฟฝ =(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘บ๐’‘+(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘บโˆ—๐’’

+(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘ฟ๐’‘+(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘ฟโˆ—๐’’

+(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐’ (18)

๏ฟฝฬ‚๏ฟฝ =(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘บโˆ—๐’’ +(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘บ๐’‘

+(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘ฟ๐’‘+(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘ฟโˆ—๐’’

+(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐’. (19)

All five terms in each of the above equations can affect theMSE. The second to the fourth terms are interferences andtheir contributions to the MSE is minimized if they becomezeros (no interferences). The first term needs to yield theparameter under estimation so that its contribution to the MSEis minimized (zero). The last term is due to the noise andwhen its covariance matrix is unknown (for the most generalcase), minimization of its contribution to MSE is impractical.However, in practice, the noise term is almost white as willbe explained below. Under this condition, the first and thelast terms are exactly the same as the system in [11] andhence, the MSE-minimizing condition (due to the first andthe last terms) is the same as that in [11]. All of the abovementioned (practically feasible) MSE-minimizing conditionscan be elaborated as follows:

1) Estimation Identifiability Condition: The identifiabilityof ๐’‘ and ๐’’ estimation requires that ๐‘บ๐ป๐‘บ is of full-rank.

2) Zero Cross Channel Interference Condition: The MSEdue to the second term (cross channel interference) in(18) and (19) is removed when ๐‘บ๐ป๐‘บโˆ— = 0๐ฟ๐‘Tx .

3) Zero Data Interference Condition: The random datainterference is completely suppressed when ๐‘บ๐ป๐‘ฟ =0๐ฟ๐‘Tx and ๐‘บ๐ป๐‘ฟโˆ— = 0๐ฟ๐‘Tx .

4) White Noise Optimality Condition: When the equivalentreceive-filter is a square-root Nyquist filter, the MSEdue to the noise is minimized when ๐‘บ๐ป๐‘บ = ๐ธ๐พ๐‘ฐ๐ฟ๐‘Tx

where ๐ธ๐พ is the training signal energy from a trans-mit antenna over ๐พ symbols (excluding CPs). The FIreceiver I/Q imbalance with a square-root raised cosinereceive filter represents this scenario.

5) For the scenario with FD receiver I/Q imbalance, thenoise covariance matrix is unknown a priori, and hence itis infeasible to develop optimal pilot designs. However,the frequency selectivity of the receiver I/Q imbalanceis typically very small (and the noise would be almostwhite) since the amplifiers/filters are designed to havea frequency-flat response within the transmission band(see also [4], [21], [27], [28]). A practical approachin this case is to assume frequency-flat receiver I/Qimbalance1 in the pilot designs which leads to therequirement of ๐‘บ๐ป๐‘บ = ๐ธ๐พ๐‘ฐ๐ฟ๐‘Tx .

When the identifiability and zero data interference condi-tions are met, the MSEs of ๐’‘ and ๐’’ estimation become

MSE๐’‘ = Tr

[(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘บโˆ—๐ธ[๐’’๐’’๐ป ]๐‘บ๐‘‡๐‘บ(๐‘บ๐ป๐‘บ

)โˆ’1

+(๐‘บ๐ป๐‘บ

)โˆ’1

๐‘บ๐ป๐‘ช๐’๐‘บ(๐‘บ๐ป๐‘บ

)โˆ’1]

(20)

MSE๐’’ = Tr

[(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘บ๐ธ[๐’‘๐’‘๐ป ]๐‘บ๐ป๐‘บโˆ—(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

+(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1

๐‘บ๐‘‡๐‘ช๐’๐‘บโˆ—(๐‘บ๐‘‡๐‘บโˆ—

)โˆ’1]

(21)

where the noise covariance matrix ๐‘ช๐’ is given by

๐‘ช๐’ = ๐œŽ2๐‘ค(๐‘ฎ๐‘…,๐ท๐‘ฎ๐ป๐‘…,๐ท +๐‘ฎ๐‘…,๐‘€๐‘ฎ๐ป

๐‘…,๐‘€ )โŠ— ๐‘ฐ๐พ . (22)

If the zero cross channel interference condition is also met,the above MSE expressions become

MSE๐’‘ = MSE๐’’ = Tr[(๐‘บ๐ป๐‘บ)โˆ’1๐‘บ๐ป๐‘ช๐’๐‘บ(๐‘บ

๐ป๐‘บ)โˆ’1]

(23)

Additionally, if the demodulator output noise samples arewhite (i.e., ๐‘ช๐’ = ๐œŽ2๐‘›๐‘ฐ), the MSE expressions simplify to

MSE๐’‘ = MSE๐’’ = ๐œŽ2๐‘›Tr[(๐‘บ๐ป๐‘บ)โˆ’1

].

The MIMO OFDM pilot design criterion satisfying theestimation identifiability, zero data interference condition, zerocross channel interference, and white noise optimality readsas (

๐‘บ๐ป๐‘บ)= ๐ธ๐พ๐‘ฐ๐‘Tx๐ฟ &

(๐‘บ๐ป๐‘บโˆ—

)= 0๐‘Tx๐ฟ

๐‘บ๐ป๐‘ฟ = 0๐‘Tx๐ฟ & ๐‘บ๐ป๐‘ฟโˆ— = 0๐‘Tx๐ฟ

}(24)

1The transmitter I/Q imbalances need not be frequency-flat since they donot affect the receiver noise statistics.

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2256 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

The above criterion can be detailed as

Condition 1 :

๐พโˆ’1โˆ‘๐‘™=0

๐‘บ๐ป๐‘– [๐‘™]๐‘ฟ๐‘˜[๐‘™] = 0๐ฟ, โˆ€๐‘–, ๐‘˜ (25)

Condition 2 :

๐พโˆ’1โˆ‘๐‘™=0

๐‘บ๐ป๐‘– [๐‘™]๐‘ฟ

โˆ—๐‘˜[๐‘™] = 0๐ฟ, โˆ€๐‘–, ๐‘˜ (26)

Condition 3 :

๐พโˆ’1โˆ‘๐‘™=0

๐‘บ๐ป๐‘– [๐‘™]๐‘บ๐‘–[๐‘™] = ๐ธ๐พ๐‘ฐ๐ฟ, โˆ€๐‘– (27)

Condition 4 :

๐พโˆ’1โˆ‘๐‘™=0

๐‘บ๐ป๐‘– [๐‘™]๐‘บ๐‘˜[๐‘™] = 0๐ฟ, โˆ€๐‘– โˆ•= ๐‘˜ (28)

Condition 5 :๐พโˆ’1โˆ‘๐‘™=0

๐‘บ๐ป๐‘– [๐‘™]๐‘บ

โˆ—๐‘˜[๐‘™] = 0๐ฟ, โˆ€๐‘–, ๐‘˜. (29)

The corresponding MSE becomes

MSE๐’‘ = MSE๐’’ =๐œŽ2๐‘ค๐ธ2๐พ

Tr[(

๐‘บ๐ป((๐‘ฎ๐‘…,๐ท๐‘ฎ๐ป

๐‘…,๐ท

+๐‘ฎ๐‘…,๐‘€๐‘ฎ๐ป๐‘…,๐‘€ )โŠ— ๐‘ฐ๐พ

)๐‘บ)โˆ’1

]. (30)

Following [26], we can also have the Cramer-Rao lowerbound (CRB) for the estimation of [๐’‘๐‘‡ , ๐’’๐‘‡ ]๐‘‡ (i.e., a theoret-ical lower bound for MSE๐’‘ + MSE๐’’) as

CRB = Tr

[{([๐‘บ, ๐‘บโˆ—]๐ป ๐‘ช๐’

โˆ’1 [๐‘บ, ๐‘บโˆ—])โˆ’1

}], (31)

which will be used as a benchmark for the performanceevaluation of the proposed pilot designs.

IV. MIMO OFDM PILOT DESIGNS

We will first investigate what characteristics of pilots willsatisfy each of the above conditions, separately. Using (12),(13) and (14), we obtain

๐‘บ๐ป๐‘– [๐‘™]๐‘ฟ๐‘˜[๐‘™] = ๐‘๐‘ญ๐ป

๐ฟฮ›๐ป๐‘–,๐‘™๐‘ซ๐‘š,๐‘™๐‘ญ ๐ฟ (32)

๐‘บ๐ป๐‘– [๐‘™]๐‘ฟ

โˆ—๐‘˜[๐‘™] = ๐‘๐‘ญ๐ป

๐ฟฮ›๐ป๐‘–,๐‘™๏ฟฝฬƒ๏ฟฝ

๐ป

๐‘š,๐‘™๐‘ญ ๐ฟ (33)

๐‘บ๐ป๐‘– [๐‘™]๐‘บ๐‘˜[๐‘™] = ๐‘๐‘ญ๐ป

๐ฟฮ›๐ป๐‘–,๐‘™ฮ›๐‘š,๐‘™๐‘ญ ๐ฟ (34)

๐‘บ๐ป๐‘– [๐‘™]๐‘บ

โˆ—๐‘˜[๐‘™] = ๐‘๐‘ญ๐ป

๐ฟฮ›๐ป๐‘–,๐‘™ฮ›ฬƒ

๐ป

๐‘š,๐‘™๐‘ญ ๐ฟ. (35)

As data are random, using (32) and (33), we can concludethat Condition-1 and 2 require

ฮ›๐ป๐‘–,๐‘™๐‘ซ๐‘š,๐‘™ = 0๐‘ & ฮ›๐ป

๐‘–,๐‘™๏ฟฝฬƒ๏ฟฝ๐ป

๐‘š,๐‘™ = 0๐‘ โˆ€๐‘–,๐‘š, ๐‘™. (36)

In other words, at each OFDM symbol, the pilot tone indexset for all antennas and the data tone index set for all antennasneed to be disjoint and each set is composed of pairs of mirrortones only. The indexes of a mirror pair is given by (๐‘˜,โˆ’๐‘˜) for๐‘˜ = 0, 1, . . . , ๐‘/2. Note that ๐‘˜ = 0 is a self-mirror tone andso is ๐‘˜ = ๐‘/2. Define ๐’ฅ๐‘™ โ‰œ โˆช๐‘Txโˆ’1

๐‘–=0 ๐’ฅ๐‘™,๐‘– and โ„๐‘™ โ‰œ โˆช๐‘Txโˆ’1๐‘–=0 โ„๐‘™,๐‘–

where ๐’ฅ๐‘™,๐‘– and โ„๐‘™,๐‘– denote the pilot (including null pilot) indexset and the data index set of ๐‘–th antenna at ๐‘™th OFDM symbol,respectively. Note that ๐’ฅ๐‘™,๐‘– can be separated into non-zero pilottone index set ๐’ฅ pilot

๐‘™,๐‘– and null pilot tone index set ๐’ฅ null๐‘™,๐‘– . Then,

Condition-1 and 2 require that if ๐‘˜ โˆˆ ๐’ฅ๐‘™ and ๐‘š โˆˆ โ„๐‘™, then(โˆ’๐‘˜)๐‘ โˆˆ ๐’ฅ๐‘™, (โˆ’๐‘š)๐‘ โˆˆ โ„๐‘™, and ๐‘˜ โˆ•= ๐‘š.

Next, by using (12), Condition-3 becomes

๐พโˆ’1โˆ‘๐‘™=0

๐‘ญ๐ป๐ฟ ฮ›

๐ป๐‘–,๐‘™ฮ›๐‘–,๐‘™๐‘ญ๐ฟ = ๐ธ๐พ๐‘ฐ๐ฟ, โˆ€๐‘– (37)

or equivalently, for ๐‘‘ โˆˆ {โˆ’๐ฟ+ 1,โˆ’๐ฟ+ 2, . . . , ๐ฟโˆ’ 1},๐‘โˆ’1โˆ‘๐‘˜=0

(๐พโˆ’1โˆ‘๐‘™=0

โˆฃ๐‘๐‘–,๐‘™[๐‘˜]โˆฃ2)๐‘’๐‘—2๐œ‹๐‘‘๐‘˜/๐‘ = ๐ธ๐พ๐›ฟ[๐‘‘]. (38)

For a typical FFT size ๐‘ which is a power of 2, define ๐ฟ0 =2โŒˆlog2(L)โŒ‰, ๐ฟ๐‘– = 2

๐‘–๐ฟ0, and ๐‘€๐‘– = ๐‘/๐ฟ๐‘–. Then, Condition-3 issatisfied when

๐พโˆ’1โˆ‘

๐‘™=0

โˆฃ๐‘๐‘–,๐‘™[๐‘˜]โˆฃ2 =

log2(๐‘€0)โˆ’1โˆ‘

๐‘š=0

๐‘€๐‘šโˆ’1โˆ‘

๐‘™=0

๐ฟ๐‘šโˆ’1โˆ‘

๐‘›=0

๐‘Ž๐‘š,๐‘™๐›ฟ[๐‘˜ โˆ’ ๐‘›๐‘€๐‘š โˆ’ ๐œ๐‘š,๐‘™],

(39)

where {๐‘Ž๐‘š,๐‘™} take on real non-negative values and satisfy

log2(๐‘€0)โˆ’1โˆ‘๐‘š=0

๐‘€๐‘šโˆ’1โˆ‘๐‘™=0

๐‘Ž๐‘š,๐‘™ = ๐ธ๐พ , ๐‘Ž๐‘š,๐‘™ โ‰ฅ 0, ๐œ๐‘š,๐‘™ โˆˆ [0,๐‘€๐‘š โˆ’ 1].(40)

Similarly, Condition-4 becomes

๐พโˆ’1โˆ‘๐‘™=0

๐‘ญ๐ป๐ฟฮ›๐ป

๐‘–,๐‘™ฮ›๐‘š,๐‘™๐‘ญ๐ฟ = 0๐ฟ, โˆ€๐‘– โˆ•= ๐‘š (41)

or equivalently, for ๐‘‘ โˆˆ {โˆ’๐ฟ+ 1,โˆ’๐ฟ+ 2, . . . , ๐ฟโˆ’ 1},๐‘โˆ’1โˆ‘๐‘˜=0

๐พโˆ’1โˆ‘๐‘™=0

๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘๐‘š,๐‘™[๐‘˜]๐‘’๐‘—2๐œ‹๐‘‘๐‘˜/๐‘ = 0, โˆ€๐‘– โˆ•= ๐‘š. (42)

With the definitions of ๐บ๐‘–,๐‘š[๐‘™, ๐‘˜] โ‰œ ๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘๐‘š,๐‘™[๐‘˜] and{๐‘”๐‘–,๐‘š[๐‘™, ๐‘›]} being the inverse DFT of {๐บ๐‘–,๐‘š[๐‘™, ๐‘˜]}, a sufficientcondition for satisfying Condition-4 is either of the twofollowing conditions:

๐พโˆ’1โˆ‘๐‘™=0

๐บ๐‘–,๐‘š[๐‘™, ๐‘˜] = 0, โˆ€๐‘– โˆ•= ๐‘š, โˆ€๐‘˜ (43)

๐‘”๐‘–,๐‘š[๐‘™, ๐‘˜] = 0,

๐‘˜ = 0, 1, . . . , ๐ฟโˆ’ 1, ๐‘ โˆ’ ๐ฟ+ 1, . . . , ๐‘ โˆ’ 1, โˆ€๐‘– โˆ•= ๐‘š. (44)

Condition-5 can be expressed as

๐พโˆ’1โˆ‘๐‘™=0

๐‘ญ๐ป๐ฟฮ›๐ป

๐‘–,๐‘™๐‘ญ๐‘ญ ๐‘‡ฮ›๐ป๐‘š,๐‘™๐‘ญ

โˆ—๐ฟ = 0๐ฟ, โˆ€๐‘–,๐‘š. (45)

Then, Condition-5 becomes, for all ๐‘– and ๐‘š,

๐พโˆ’1โˆ‘๐‘™=0

๐‘ญ๐ป๐ฟฮ›๐ป

๐‘–,๐‘™ฮ›ฬƒ๐ป

๐‘š,๐‘™๐‘ญ ๐ฟ = ๐‘ญ๐ป๐ฟ

(๐พโˆ’1โˆ‘๐‘™=0

ฮ›๐ป๐‘–,๐‘™ฮ›ฬƒ

๐ป

๐‘š,๐‘™

)๐‘ญ ๐ฟ = 0๐ฟ,

(46)

or equivalently, for ๐‘‘ โˆˆ {โˆ’๐ฟ+ 1,โˆ’๐ฟ+ 2, . . . , ๐ฟโˆ’ 1},๐‘โˆ’1โˆ‘๐‘˜=0

(๐พโˆ’1โˆ‘๐‘™=0

๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘โˆ—๐‘š,๐‘™[๐‘ โˆ’ ๐‘˜]

)๐‘’๐‘—2๐œ‹๐‘‘๐‘˜/๐‘ = 0, โˆ€๐‘–,๐‘š. (47)

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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2257

This mandates that either the vector{(โˆ‘๐พโˆ’1๐‘™=0 ๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘

โˆ—๐‘š,๐‘™[๐‘ โˆ’ ๐‘˜]

): ๐‘˜ = 0, . . . , ๐‘ โˆ’ 1

}be in

the null space of the columns of ๐‘ญ๐ฟ and ๐‘ญ๐ป๐ฟ , โˆ€๐‘–,๐‘š, ๐‘˜, as

๐พโˆ’1โˆ‘๐‘™=0

๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘โˆ—๐‘š,๐‘™[๐‘ โˆ’ ๐‘˜] = 0, (48)

๐พโˆ’1โˆ‘๐‘™=0

๐‘โˆ—๐‘–,๐‘™[๐‘˜]๐‘โˆ—๐‘š,๐‘™[๐‘ โˆ’ ๐‘˜] = ๐‘’๐‘—2๐œ‹๐‘‘๐‘˜/๐‘ , ๐‘‘ โˆˆ {๐ฟ, . . . , ๐‘ โˆ’ ๐ฟ},

(49)

or, for ๐‘‘ = โˆ’(๐ฟโˆ’ 1), . . . , (๐ฟโˆ’ 1),๐‘โˆ’1โˆ‘๐‘˜=0

๐‘๐‘–,๐‘™[๐‘˜]๐‘๐‘š,๐‘™[๐‘ โˆ’ ๐‘˜]๐‘’๐‘—2๐œ‹๐‘‘๐‘˜/๐‘ = 0, โˆ€๐‘–,๐‘š, ๐‘™. (50)

By considering ๐‘– = ๐‘š in (50), one can easily find that 2๐ฟ0

pilot tones (may include null pilots) are necessary for onetransmit antenna.

Combining all of the above pilot characteristics, we presentseveral pilot designs which satisfy the five design criteria.These designs will be denoted by two terms โ€“ how pilotsof different antennas are decoupled and how mirror toneinterferences are suppressed. The same pilot signal energyover ๐พ symbols for each antenna as required in Condition-3 will not be explicitly mentioned in the following designs.Without loss of generality, we will present unit-amplitudepilot symbols. The existing MIMO OFDM pilot designsfrom [11], namely time division multiplexing (TDM), codedivision multiplexing across time domain (CDM-T), CDMacross frequency domain (CDM-F), frequency division multi-plexing (FDM), and time and frequency division multiplexing(TFDM), will be extensively used in our designs. Due to spacelimitation, details of these existing designs are referred to[11]. How our pilot designs satisfy the five design criteriawill be briefly mentioned for the first design, but it shouldbe obvious and hence will be skipped for the other designs.Examples of our pilot designs are provided in Tables I and IIfor illustration of the concepts where parameters are chosen forthe convenience of the presentation. In some designs, severaloptions are presented by separating them with dashed lines.

A. [TDM; TD / C-F] Design

In this design, pilots of different antennas are decoupled byTDM design while mirror tone interferences are suppressed bymeans of time disjointness (TD) or code design in frequencydomain (C-F). The pilot and data index sets for the ๐‘–th transmitantenna at the ๐‘™th OFDM symbol are given by

๐’ฅ๐‘™,๐‘– =

{ {0,๐‘€0/2,๐‘€0, . . . , ๐‘ โˆ’๐‘€0/2}, if ๐‘– = ๐‘™โˆ…, else

(51)

โ„๐‘™ = {0, 1, 2, . . . , ๐‘ โˆ’ 1} โˆ– ๐’ฅ๐‘™ (52)

where TDM pilot assignment for different antennas can beeasily noticed. Due to TDM, there is no mirror tone interfer-ence across antennas (i.e., mirror tone interference suppression

through TD). For each antenna, the pilots are given by

๐‘๐‘–,๐‘–[0] = ยฑ๐‘๐‘–,๐‘–[๐‘/2] = ยฑ1 or ยฑ ๐‘— (53)

๐‘๐‘–,๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜ , arbitrary ๐œ™๐‘–,๐‘˜,

๐‘˜ โˆˆ {๐‘€0/2,๐‘€0, . . . , (๐‘ โˆ’๐‘€0)/2} (54)

๐‘๐‘–,๐‘–[๐‘˜] = (โˆ’1)(2๐‘˜โˆ’๐‘)/๐‘€0๐‘2๐‘–,๐‘–[0]๐‘โˆ—๐‘–,๐‘–[โˆ’๐‘˜],

๐‘˜ โˆˆ {(๐‘ +๐‘€0)/2, ๐‘/2 +๐‘€0, . . . , ๐‘ โˆ’๐‘€0/2} (55)

๐‘๐‘–,๐‘–[๐‘˜] = 0, ๐‘˜ /โˆˆ ๐’ฅ๐‘–,๐‘–. (56)

For example, {๐‘๐‘–,๐‘–[๐‘˜] : ๐‘˜ โˆˆ ๐’ฅ๐‘–,๐‘–} can be either {ยฑ1, ๐‘Ž1, ๐‘Ž2,. . . , ๐‘Ž๐ฟ0โˆ’1, ยฑ1,โˆ’๐‘Žโˆ—๐ฟ0โˆ’1, ๐‘Žโˆ—๐ฟ0โˆ’2, โˆ’๐‘Žโˆ—๐ฟ0โˆ’3, . . . , ๐‘Žโˆ—2,โˆ’๐‘Žโˆ—1}or {ยฑ๐‘—, ๐‘Ž1, ๐‘Ž2, . . . , ๐‘Ž๐ฟ0โˆ’1, ยฑ๐‘—, ๐‘Žโˆ—๐ฟ0โˆ’1, โˆ’๐‘Žโˆ—๐ฟ0โˆ’2, ๐‘Ž

โˆ—๐ฟ0โˆ’3,

. . . ,โˆ’๐‘Žโˆ—2, ๐‘Žโˆ—1} where {๐‘Ž๐‘˜} are arbitrary unit amplitude sym-bols. The above pilot codes across the frequency domain aredesigned such that the mirror interference becomes zero, i.e.,to satisfy (50) for ๐‘– = ๐‘š. Note that (51) and (52) guaranteeConditions 1 and 2, while TDM fulfills Conditions 3 and 4.TDM and C-F guarantee Condition 5. This design requires2๐ฟ0 pilot tones in each symbol and a total of ๐พ = ๐‘Tx

symbols. ๐ฟ๐‘š and ๐‘€๐‘š with ๐‘š > 0 can be used instead of ๐ฟ0

and ๐‘€0, but they cost more overhead.

B. [CDM-T; C-T] Design Without Self-Mirror Tones

Pilots of different antennas are decoupled by CDM-T de-sign. Pilot mirror tone interferences are suppressed by codedesign across time domain (C-T). The pilot and data indexsets are given by

๐’ฅ๐‘™,๐‘– = {๐‘€0

2,3๐‘€0

2,5๐‘€0

2, . . . , ๐‘ โˆ’ ๐‘€0

2}, โˆ€๐‘–, ๐‘™ (57)

โ„๐‘™ = {0, 1, 2, . . . , ๐‘ โˆ’ 1} โˆ– ๐’ฅ๐‘™ (58)

where self-mirror tones (i.e., index 0 and ๐‘/2) are excludedfrom the pilot index set. Define

๐’„๐‘–[๐‘˜] โ‰œ [๐‘๐‘–,0[๐‘˜], ๐‘๐‘–,1[๐‘˜], . . . , ๐‘๐‘–,๐พโˆ’1[๐‘˜]]๐‘‡ (59)

[๐’—๐‘š]๐‘˜ โ‰œ ๐‘’๐‘—2๐œ‹๐‘š๐‘˜/๐พ ,๐‘š, ๐‘˜ โˆˆ {0, 1, . . . ,๐พ โˆ’ 1},๐พ โ‰ฅ 2(๐‘Tx + ๐‘›), ๐‘› โˆˆ {0, 1, 2, . . .}. (60)

For ๐‘˜ โˆˆ {๐‘€0/2, 3๐‘€0/2, . . . , ๐‘/2 โˆ’ ๐‘€0/2}, the pilots aregiven as

๐’„0[๐‘˜] = ๐‘’๐‘—๐œ™0,๐‘˜ , ๐‘˜ โˆˆ ๐’ฅ pilot0 (61)

๐’„0[โˆ’๐‘˜] = ๐‘’๐‘—๐œ™0,โˆ’๐‘˜diag{๐’—1}๐’„โˆ—0[๐‘˜] (62)

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜diag{๐’—2๐‘–}๐’„0[๐‘˜], ๐‘– = 1, . . . , ๐‘Tx โˆ’ 1 (63)

๐’„๐‘–[โˆ’๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,โˆ’๐‘˜diag{๐’—2๐‘–+1}๐’„โˆ—0[๐‘˜], ๐‘– = 1, . . . , ๐‘Tx โˆ’ 1(64)

where {๐œ™๐‘–,ยฑ๐‘˜} are arbitrary. In the above design, for ๐‘˜ โˆˆ{๐‘€0/2, 3๐‘€0/2, . . . , ๐‘/2 โˆ’๐‘€0/2}, {๐’„๐‘–[๐‘˜]} for different an-tenna ๐‘– โˆˆ {0, 1, . . . , ๐‘Txโˆ’1} are constructed using diag{๐’—2๐‘–}while {๐’„๐‘–[โˆ’๐‘˜]} are based on diag{๐’—2๐‘–+1}. Note that anypermutation of {๐‘ฃ๐‘– : ๐‘– = 1, . . . , 2๐‘Tx โˆ’ 1} can be associatedto {๐’„๐‘–[๐‘˜]} in the above equations. Overhead consideration willset ๐พ = 2๐‘Tx. Although ๐‘€๐‘– can be used in place of ๐‘€0, itis not desirable due to larger overhead.

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2258 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

TABLE IEXAMPLES OF PROPOSED PILOT DESIGNS

Design antenna symbol Pilotsindex ๐‘– index ๐‘™

[TDM; TD / C-F] ๐’ฅ๐‘™,๐‘– = {0, 2, 4, 6, 8, 10, 12, 14}๐‘ = 16, ๐ฟ0 = 4, ๐‘€0 = 4, 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘—, ๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘—, ๐‘Ž

โˆ—3 ,โˆ’๐‘Žโˆ—

2, ๐‘Žโˆ—1}

๐พ = 2, ๐‘Tx = 2 1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘—, ๐‘Ž4, ๐‘Ž5, ๐‘Ž6,โˆ’๐‘—, ๐‘Žโˆ—6 ,โˆ’๐‘Žโˆ—

5, ๐‘Žโˆ—4}

[TDM; TD / C-F] ๐’ฅ๐‘™,๐‘– = {0, 2, 4, 6, 8, 10, 12, 14}๐‘ = 16, ๐ฟ0 = 4, ๐‘€0 = 4, 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {1, ๐‘Ž1, ๐‘Ž2, ๐‘Ž3, 1,โˆ’๐‘Žโˆ—

3, ๐‘Žโˆ—2 ,โˆ’๐‘Žโˆ—

1}๐พ = 2, ๐‘Tx = 2 1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {โˆ’1, ๐‘Ž4, ๐‘Ž5, ๐‘Ž6, 1,โˆ’๐‘Žโˆ—

6, ๐‘Žโˆ—5 ,โˆ’๐‘Žโˆ—

4}[CDM-T; C-T] ๐’ฅ๐‘™,๐‘– = {2, 6, 10, 14}

without self-mirror 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž1, ๐‘Ž5, ๐‘Žโˆ—5, ๐‘Ž

โˆ—1}

๐‘ = 16, ๐ฟ0 = 4, ๐‘€0 = 4, 0 1 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž2, ๐‘Ž6, ๐‘Žโˆ—6๐‘’

๐‘—๐œ‹/2, ๐‘Žโˆ—2๐‘’

๐‘—๐œ‹/2}๐พ = 4, ๐‘Tx = 2 0 2 ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž3, ๐‘Ž7,โˆ’๐‘Žโˆ—

7,โˆ’๐‘Žโˆ—3}

0 3 ๐‘3,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž4, ๐‘Ž8, ๐‘Žโˆ—8๐‘’

โˆ’๐‘—๐œ‹/2, ๐‘Žโˆ—4๐‘’

โˆ’๐‘—๐œ‹/2}1 0 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž1, ๐‘Ž5, ๐‘Ž

โˆ—5, ๐‘Ž

โˆ—1}

1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {โˆ’๐‘Ž2,โˆ’๐‘Ž6, ๐‘Žโˆ—6๐‘’

๐‘—๐œ‹/2, ๐‘Žโˆ—2๐‘’

๐‘—๐œ‹/2}1 2 ๐‘2,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘Ž3, ๐‘Ž7,โˆ’๐‘Žโˆ—

7,โˆ’๐‘Žโˆ—3}

1 3 ๐‘3,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {โˆ’๐‘Ž4,โˆ’๐‘Ž8, ๐‘Žโˆ—8๐‘’

โˆ’๐‘—๐œ‹/2, ๐‘Žโˆ—4๐‘’

โˆ’๐‘—๐œ‹/2}[CDM-T; C-T] ๐’ฅ๐‘™,๐‘– = {0, 4, 8, 12}with self-mirror 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {1, ๐‘Ž1, 1, ๐‘Ž

โˆ—1}

๐‘ = 16, ๐ฟ0 = 4, ๐‘€0 = 4, 0 1 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’๐‘—๐œ‹/4, ๐‘Ž2, ๐‘’๐‘—๐œ‹/4, ๐‘Žโˆ—

2๐‘’๐‘—๐œ‹/2}

๐พ = 4, ๐‘Tx = 2 0 2 ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’๐‘—๐œ‹/2, ๐‘Ž3, ๐‘’๐‘—๐œ‹/2,โˆ’๐‘Žโˆ—

3}0 3 ๐‘3,0[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’โˆ’๐‘—๐œ‹/4, ๐‘Ž4, ๐‘’

โˆ’๐‘—๐œ‹/4, ๐‘Žโˆ—4๐‘’

โˆ’๐‘—๐œ‹/2}1 0 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {1, ๐‘Ž1, 1, ๐‘Ž

โˆ—1}

1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’๐‘—๐œ‹/4, ๐‘Ž2, ๐‘’๐‘—๐œ‹/4, ๐‘Žโˆ—

2๐‘’๐‘—๐œ‹/2}

1 2 ๐‘2,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’๐‘—๐œ‹/2, ๐‘Ž3, ๐‘’๐‘—๐œ‹/2,โˆ’๐‘Žโˆ—

3}1 3 ๐‘3,1[๐‘˜ โˆˆ ๐’ฅ๐‘™,๐‘–] = {๐‘’โˆ’๐‘—๐œ‹/4, ๐‘Ž4, ๐‘’

โˆ’๐‘—๐œ‹/4, ๐‘Žโˆ—4๐‘’

โˆ’๐‘—๐œ‹/2}[CDM-F; Null] ๐’ฅ pilot

๐‘– = {1, 2, 9, 10}๐‘ = 16, ๐ฟ0 = 2, ๐‘€0 = 8, ๐’ฅ null

๐‘– = {6, 7, 14, 15}๐พ = 1, ๐‘Tx = 2 0 0 ๐‘0[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘Ž4}1 0 ๐‘1[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1,โˆ’๐‘Ž2, ๐‘Ž3,โˆ’๐‘Ž4}[CDM-F; Null] ๐’ฅ pilot

๐‘– = {1, 5, 9, 13, 17, 21, 25, 29, }with equi-spaced pilots ๐’ฅ null

๐‘– = {3, 7, 11, 15, 19, 23, 27, 31}๐‘ = 32, ๐ฟ0 = 2, ๐‘€0 = 16, 0 0 ๐‘0[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘Ž4, ๐‘Ž5, ๐‘Ž6, ๐‘Ž7, ๐‘Ž8}๐พ = 1, ๐‘Tx = 4 1 0 ๐‘1[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1๐‘’๐‘—๐œ‹/16, ๐‘Ž2๐‘’

๐‘—5๐œ‹/16, ๐‘Ž3๐‘’๐‘—9๐œ‹/16, ๐‘Ž4๐‘’

๐‘—13๐œ‹/16 ,

๐‘Ž4๐‘’๐‘—โˆ’15๐œ‹/16 , ๐‘Ž3๐‘’

๐‘—โˆ’11๐œ‹/16, ๐‘Ž2๐‘’๐‘—โˆ’7๐œ‹/16, ๐‘Ž1๐‘’

๐‘—โˆ’3๐œ‹/16}2 0 ๐‘2[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1๐‘’๐‘—๐œ‹/8, ๐‘Ž2๐‘’

๐‘—5๐œ‹/8, ๐‘Ž3๐‘’๐‘—โˆ’7๐œ‹/8, ๐‘Ž4๐‘’

๐‘—โˆ’3๐œ‹/8,

๐‘Ž4๐‘’๐‘—๐œ‹/8, ๐‘Ž3๐‘’

๐‘—5๐œ‹/8, ๐‘Ž2๐‘’๐‘—โˆ’7๐œ‹/8, ๐‘Ž1๐‘’

๐‘—โˆ’3๐œ‹/8}3 0 ๐‘3[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1๐‘’๐‘—3๐œ‹/8, ๐‘Ž2๐‘’

๐‘—โˆ’๐œ‹/8, ๐‘Ž3๐‘’๐‘—โˆ’3๐œ‹/8, ๐‘Ž4๐‘’

๐‘—7๐œ‹/8,

๐‘Ž4๐‘’๐‘—3๐œ‹/8, ๐‘Ž3๐‘’

๐‘—โˆ’๐œ‹/8, ๐‘Ž2๐‘’๐‘—โˆ’3๐œ‹/8, ๐‘Ž1๐‘’

๐‘—7๐œ‹/8}[FDM; Null] ๐’ฅ pilot

๐‘– = {1, 3, 9, 11}๐‘ = 16, ๐ฟ0 = 2, ๐‘€0 = 8, ๐’ฅ null

๐‘– = {5, 7, 13, 15}๐พ = 1, ๐‘Tx = 2 0 0 ๐‘0[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1, 0, ๐‘Ž2, 0}1 0 ๐‘1[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {0, ๐‘Ž3, 0, ๐‘Ž4}[TFDM; Null / C-F] ๐’ฅ null

๐‘™,๐‘– =โˆช๐’ฅ๐‘š,๐‘˜ where ๐‘š โˆ•= ๐‘™ and ๐‘˜ โˆ•= ๐‘–

๐‘ = 64, ๐ฟ0 = 2, ๐‘€1 = 16, 0 0 ๐’ฅ0,0 = ๐’ฏ2,๐œ0,๐‘’ = {1, 33, } ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ0,0] = {๐‘Ž1, ๐‘Ž2}๐พ = 2, ๐‘Tx = 3 0 1 ๐’ฅ0,1 = ๐’ฏ2,๐œ0,๐‘œ = {17, 49} ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ0,1] = {๐‘Ž3, ๐‘Ž4}

1 0 ๐’ฅ1,0 = ๐’ฏ2,๐œ1,๐‘’ = {15, 47, } ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ1,0] = {๐‘Ž5, ๐‘Ž6}1 1 ๐’ฅ1,1 = ๐’ฏ2,๐œ1,๐‘œ = {31, 63} ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ1,1] = {๐‘Ž7, ๐‘Ž8}2 0 ๐’ฅ2,0 = ๐’ฏ2,๐‘€1 = {0, 16, 32, 48} ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ2,0 ] = {ยฑโˆฃ๐‘1โˆฃ, ๐‘2,ยฑโˆฃ๐‘1โˆฃ,โˆ’๐‘โˆ—2}2 0 ๐’ฅ2,0 = ๐’ฏ2,๐‘€1,๐‘œ = {0, 32} ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ2,0] = {1, 1}

(alternative) 1 ๐’ฅ2,1 = ๐’ฏ2,๐‘€1,๐‘’ = {16, 48} ๐‘2,1[๐‘˜ โˆˆ ๐’ฅ2,1] = {๐‘1, ๐‘2}2 0 ๐’ฅ2,0 = ๐’ฏ2,๐‘€1/2,๐‘œ = {8, 40} ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ2,0] = {๐‘Ž9, ๐‘Ž10}

(alternative) 1 ๐’ฅ2,1 = ๐’ฏ2,๐‘€1/2,๐‘’ = {24, 56} ๐‘2,1[๐‘˜ โˆˆ ๐’ฅ2,1] = {๐‘Ž11, ๐‘Ž12}[TFDM / CDM-T; Null / C-T] ๐’ฅ null

๐‘™,๐‘– =โˆช๐’ฅ๐‘š,๐‘˜ where ๐‘š โˆ•= ๐‘™ and ๐‘˜ โˆ•= ๐‘–

๐‘ = 64, ๐ฟ0 = 4, ๐‘€0 = 16, 0 0 ๐’ฅ0,0 = ๐’ฏ0,๐œ0,0 = {1, 17} ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ0,0] = {๐‘Ž1, ๐‘Ž2}๐พ = 4, ๐‘Tx = 5 0 1 ๐’ฅ0,1 = ๐’ฏ0,๐œ0,0 = {1, 17} ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ0,1] = {๐‘Ž3, ๐‘Ž4}

0 2 ๐’ฅ0,2 = ๐’ฏ0,๐œ0,1 = {33, 49} ๐‘0,2[๐‘˜ โˆˆ ๐’ฅ0,2] = {๐‘Ž5, ๐‘Ž6}0 3 ๐’ฅ0,3 = ๐’ฏ0,๐œ0,1 = {33, 49} ๐‘0,3[๐‘˜ โˆˆ ๐’ฅ0,2] = {๐‘Ž7, ๐‘Ž8}1 0 ๐’ฅ1,0 = ๐’ฏ0,๐œ0,0 = {1, 17} ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ1,0 ] = {๐‘Ž1๐‘’

๐‘—๐œ™1 ,โˆ’๐‘Ž2๐‘’๐‘—๐œ™2}

1 1 ๐’ฅ1,1 = ๐’ฏ0,๐œ0,0 = {1, 17} ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ1,1 ] = {โˆ’๐‘Ž3๐‘’๐‘—๐œ™1 , ๐‘Ž4๐‘’

๐‘—๐œ™2}1 2 ๐’ฅ1,2 = ๐’ฏ0,๐œ0,1 = {33, 49} ๐‘1,2[๐‘˜ โˆˆ ๐’ฅ1,2] = {๐‘Ž5๐‘’

๐‘—๐œ™3 , ๐‘Ž6๐‘’๐‘—๐œ™4}

1 3 ๐’ฅ1,3 = ๐’ฏ0,๐œ0,1 = {33, 49} ๐‘1,3[๐‘˜ โˆˆ ๐’ฅ1,2] = {โˆ’๐‘Ž7๐‘’๐‘—๐œ™3 ,โˆ’๐‘Ž8๐‘’

๐‘—๐œ™4}2 0 ๐’ฅ2,0 = ๐’ฏ0,๐œ15,0 = {15, 31} ๐‘2,0[๐‘˜ โˆˆ ๐’ฅ2,0] = {๐‘1, ๐‘2}2 1 ๐’ฅ2,1 = ๐’ฏ0,๐œ15,0 = {15, 31} ๐‘2,1[๐‘˜ โˆˆ ๐’ฅ2,1] = {๐‘3, ๐‘4}2 2 ๐’ฅ2,2 = ๐’ฏ0,๐œ15,1 = {47, 63} ๐‘2,2[๐‘˜ โˆˆ ๐’ฅ2,2] = {๐‘5, ๐‘6}2 3 ๐’ฅ2,3 = ๐’ฏ0,๐œ15,1 = {47, 63} ๐‘2,3[๐‘˜ โˆˆ ๐’ฅ2,2] = {๐‘7, ๐‘8}3 0 ๐’ฅ3,0 = ๐’ฏ0,๐œ15,0 = {15, 31} ๐‘3,0[๐‘˜ โˆˆ ๐’ฅ3,0] = {๐‘1๐‘’๐‘—๐œ™5 , ๐‘2๐‘’

๐‘—๐œ™6}3 1 ๐’ฅ3,1 = ๐’ฏ0,๐œ15,0 = {15, 31} ๐‘3,1[๐‘˜ โˆˆ ๐’ฅ3,1] = {โˆ’๐‘3๐‘’

๐‘—๐œ™5 ,โˆ’๐‘4๐‘’๐‘—๐œ™6}

3 2 ๐’ฅ3,2 = ๐’ฏ0,๐œ15,1 = {47, 63} ๐‘3,2[๐‘˜ โˆˆ ๐’ฅ3,2] = {๐‘5๐‘’๐‘—๐œ™7 , ๐‘6๐‘’๐‘—๐œ™8}

3 3 ๐’ฅ3,3 = ๐’ฏ0,๐œ15,1 = {47, 63} ๐‘3,3[๐‘˜ โˆˆ ๐’ฅ3,2] = {๐‘7๐‘’๐‘—๐œ™7 , ๐‘8๐‘’๐‘—๐œ™8}

4 0 ๐’ฅ4,0 = ๐’ฏ0,๐œ2,0 = {4, 20} ๐‘4,0[๐‘˜ โˆˆ ๐’ฅ4,0] = {๐‘1, ๐‘2}4 1 ๐’ฅ4,1 = ๐’ฏ0,๐œ2,1 = {44, 60} ๐‘4,1[๐‘˜ โˆˆ ๐’ฅ4,1] = {๐‘3, ๐‘4}4 ๐’ฅ4,0 = ๐’ฏ0,๐‘€0/2 = {0, 8, 16, 24, 32, 40, 48, 56}

(alternative) 0 ๐‘4,0[๐‘˜ โˆˆ ๐’ฅ4,0] = {ยฑโˆฃ๐‘1โˆฃ, ๐‘1, ๐‘2,ยฑโˆฃ๐‘0โˆฃ,โˆ’๐‘โˆ—2, ๐‘โˆ—1,โˆ’๐‘โˆ—0}

Page 8: 2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, โ€ฆhxm025000/PilotIQTCOM2010Aug.pdfMINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT

MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2259

TABLE IIEXAMPLES OF PROPOSED PILOT DESIGNS CONTโ€™D.

Design antenna symbol Pilotsindex ๐‘– index ๐‘™

[CDM-F; C-T] ๐’ฅ๐‘– = {4, 12, 20, 28, 36, 44, 52, 60}๐‘ = 64, ๐ฟ0 = 4, ๐‘€0 = 16 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘Ž4, ๐‘Ž5, ๐‘Ž6, ๐‘Ž7, ๐‘Ž8}

๐พ = 2, ๐‘Tx = 2 0 1 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘—๐‘Ž1, ๐‘—๐‘Ž2, ๐‘—๐‘Ž3, ๐‘—๐‘Ž4, ๐‘—๐‘Ž5, ๐‘—๐‘Ž6, ๐‘—๐‘Ž7, ๐‘—๐‘Ž8}1 0 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1,โˆ’๐‘Ž2, ๐‘Ž3,โˆ’๐‘Ž4, ๐‘Ž5,โˆ’๐‘Ž6, ๐‘Ž7,โˆ’๐‘Ž8}1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘—๐‘Ž1,โˆ’๐‘—๐‘Ž2, ๐‘—๐‘Ž3,โˆ’๐‘—๐‘Ž4, ๐‘—๐‘Ž5,โˆ’๐‘—๐‘Ž6, ๐‘—๐‘Ž7,โˆ’๐‘—๐‘Ž8}

[FDM; C-T] ๐’ฅ๐‘– = {4, 12, 20, 28, 36, 44, 52, 60}๐‘ = 64, ๐ฟ0 = 4, ๐‘€0 = 16 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, 0, ๐‘Ž2, 0, ๐‘Ž3, 0, ๐‘Ž4, 0}

๐พ = 2, ๐‘Tx = 2 0 1 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘—๐‘Ž1, 0, ๐‘—๐‘Ž2, 0, ๐‘—๐‘Ž3, 0, ๐‘—๐‘Ž4, 0}1 0 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0, ๐‘Ž5, 0, ๐‘Ž6, 0, ๐‘Ž7, 0, ๐‘Ž8}1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0, ๐‘—๐‘Ž5, 0, ๐‘—๐‘Ž6, 0, ๐‘—๐‘Ž7, 0, ๐‘—๐‘Ž8}

[FDM; C-T] ๐’ฅ๐‘– = {4, 12, 20, 28, 36, 44, 52, 60}(alternative) 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, 0, ๐‘Ž2, 0, ๐‘Ž3, 0, ๐‘Ž4, 0}

๐‘ = 64, ๐ฟ0 = 4, ๐‘€0 = 16 0 1 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, 0, ๐‘Ž2, 0,โˆ’๐‘Ž3, 0,โˆ’๐‘Ž4, 0}๐พ = 2, ๐‘Tx = 2 1 0 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0, ๐‘Ž5, 0, ๐‘Ž6, 0, ๐‘Ž7, 0, ๐‘Ž8}

1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0, ๐‘Ž5, 0, ๐‘Ž6, 0,โˆ’๐‘Ž7, 0,โˆ’๐‘Ž8}[FDM; C-T] ๐’ฅ๐‘– = {4, 12, 20, 28, 36, 44, 52, 60}(alternative) 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, 0, ๐‘Ž2, 0, ๐‘Ž3, 0, ๐‘Ž4, 0}

๐‘ = 64, ๐ฟ0 = 4, ๐‘€0 = 16 0 1 ๐‘0,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {๐‘Ž1, 0, ๐‘Ž2, 0, ๐‘Ž3, 0, ๐‘Ž4, 0}๐พ = 2, ๐‘Tx = 2 1 0 ๐‘1,0[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0, ๐‘Ž5, 0, ๐‘Ž6, 0, ๐‘Ž7, 0, ๐‘Ž8}

1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ๐‘–] = {0,โˆ’๐‘Ž5, 0,โˆ’๐‘Ž6, 0,โˆ’๐‘Ž7, 0,โˆ’๐‘Ž8}[TDM; Null] ๐’ฅ pilot

๐‘– = {1, 2, 9, 10}๐‘ = 16, ๐ฟ0 = 2, ๐‘€0 = 8, ๐’ฅ null

๐‘– = {6, 7, 14, 15}๐พ = 2, ๐‘Tx = 2 0 0 ๐‘0,0[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘Ž4}1 1 ๐‘1,1[๐‘˜ โˆˆ ๐’ฅ pilot

๐‘– ] = {๐‘Ž1, ๐‘Ž2, ๐‘Ž3, ๐‘Ž4}

C. [CDM-T; C-T] Design With Self-Mirror Tones

The C-T design in the above subsection uses ๐’—๐‘™ and ๐’—๐‘š with๐‘™ โˆ•= ๐‘š for tones ๐‘˜ and โˆ’๐‘˜. This approach becomes irrelevantfor self-mirror tones (i.e., for ๐‘˜ = (โˆ’๐‘˜)๐‘ ). When the pilotscontain self-mirror tones, the index sets are given by

๐’ฅ๐‘™,๐‘– = {0,๐‘€0, 2๐‘€0, . . . , ๐‘ โˆ’๐‘€0}, โˆ€๐‘–, ๐‘™ (65)

โ„๐‘™ = {0, 1, 2, . . . , ๐‘ โˆ’ 1} โˆ– ๐’ฅ๐‘™. (66)

For non-self-mirror tones, the C-T design from the previoussubsection is applied with ๐พ defined below. At self-mirrortones ๐‘˜ = 0 and ๐‘/2, we present two designs. The first designis given by

๐’„0[๐‘˜] = ๐‘’๐‘—๐œ™0,๐‘˜๐’—โ€ฒ๐‘›, ๐‘› โˆˆ {1, . . . ,๐พ โˆ’ 1} (67)

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜diag{๐’—๐‘š๐‘–}๐’„โˆ—0[๐‘˜], ๐‘– = 1, . . . , ๐‘Tx โˆ’ 1 (68)

where {๐œ™๐‘–,๐‘˜} are arbitrary, [๐’—โ€ฒ๐‘›]๐‘˜ โ‰œ

โˆš[๐’—๐‘›]๐‘˜ and the indexes

๐‘› and ๐‘š๐‘– satisfy i) (๐‘š๐‘– + ๐‘š๐‘— โˆ’ ๐‘›)๐พ โˆ•= 0, ๐‘š๐‘–,๐‘š๐‘— โˆˆ{0, 1, . . . , ๐‘Txโˆ’1},๐‘š๐‘– โˆ•= ๐‘š๐‘— if ๐‘– โˆ•= ๐‘—, ii) (๐‘›โˆ’๐‘š๐‘–)๐พ โˆ•= 0, iii)(๐‘š๐‘–โˆ’๐‘š๐‘—)๐พ โˆ•= 0, โˆ€๐‘š๐‘– โˆ•= ๐‘š๐‘— . For simplicity, we can set ๐‘› = 1in the above equations which yields 2 โ‰ค ๐‘š, ๐‘™ โ‰ค ๐พ/2 and๐พ = 2๐‘Tx. ๐พ can also be 2(๐‘Tx+๐œ) with ๐œ โˆˆ {0, 1, 2, . . .}.

The second design at self-mirror tones is defined by

๐’„0[๐‘˜] = ๐‘’๐‘—๐œ™0,๐‘˜๐’—๐‘›, ๐‘› โˆˆ {1, . . . ,๐พ โˆ’ 1} & ๐‘› โˆ•= ๐พ/2 (69)

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜diag{๐’—๐‘š๐‘–}๐’„โˆ—0[๐‘˜], ๐‘– = 1, . . . , ๐‘Tx โˆ’ 1 (70)

with the indexes ๐‘› and ๐‘š๐‘– satisfying i) ๐‘š๐‘– โˆ•= ๐‘›, (๐‘š๐‘– โˆ’๐‘›)๐พ/2 โˆ•= 0, ii) (๐‘š๐‘– + ๐‘š๐‘— โˆ’ 2๐‘›)๐พ โˆ•= 0, ๐‘š๐‘–,๐‘š๐‘— โˆˆ{0, 1, . . . , ๐‘Tx โˆ’ 1},๐‘š๐‘– โˆ•= ๐‘š๐‘— if ๐‘– โˆ•= ๐‘—, iii) (2๐‘›โˆ’๐‘š๐‘–)๐พ โˆ•= 0,and {๐œ™๐‘–,๐‘˜} are arbitrary. By simply setting ๐‘› = 1, we have3 โ‰ค ๐‘š โ‰ค ๐พ/2 and ๐พ = 2(๐‘Tx + ๐œ) with ๐œ โˆˆ {1, 2, . . .}.Using ๐‘€๐‘š in place of ๐‘€0 will require more overhead.

D. [CDM-F; Null] Design

This design uses ๐‘‰ ๐ฟ๐‘š constant amplitude pilot tones withthe index set ๐’ฅ pilot

0,๐‘– = ๐’ฅ pilot0 and ๐‘‰ ๐ฟ๐‘š null tones with the

index set ๐’ฅ null0,๐‘– = ๐’ฅ null

0 where ๐’ฅ pilot0 and ๐’ฅ null

0 are mirrorsof each other, and ๐‘‰ โ‰ฅ ๐‘Tx and ๐‘‰ ๐ฟ๐‘š โ‰ค ๐‘/2. Self-mirrortones cannot be used. Define ๐’ฏ๐‘›,๐‘˜ โ‰œ [๐‘˜, ๐‘˜ +๐‘€๐‘›, ๐‘˜ + 2๐‘€๐‘›,. . . , ๐‘˜+๐‘โˆ’๐‘€๐‘›] which consists of cyclically equi-spaced ๐ฟ๐‘›

indexes from [0, ๐‘ โˆ’ 1]. Then the index sets are given by

๐’ฅ pilot0,๐‘– = ๐’ฅ pilot

0 =

๐‘‰โˆ’1โˆช๐‘˜=0

๐’ฏ๐‘›,๐œ๐‘˜ (71)

๐’ฅ null0,๐‘– = ๐’ฅ null

0 = {๐‘ โˆ’ ๐’ฅ pilot0 } (72)

โ„๐‘™ = {0, 1, . . . , ๐‘ โˆ’ 1} โˆ– {๐’ฅ pilot0 โˆช ๐’ฅ null

0 } (73)

where ๐œ๐‘˜ โˆˆ {{1, 2, . . . ,๐‘€๐‘› โˆ’ 1} โˆ– {๐‘€๐‘›/2}}, ๐œ๐‘˜ โˆ•= ๐œ๐‘› if ๐‘˜ โˆ•=๐‘›, {๐œ๐‘˜} โˆฉ {๐‘€๐‘› โˆ’ ๐œ๐‘˜} = โˆ…, and ๐‘€๐‘› โ‰ฅ 2๐‘Tx + 2.

Due to the mirror null tones, the I/Q imbalance inducedinterferences are completely suppressed. The pilots of differentantennas are decoupled by CDM-F design as

๐’„0[๐‘˜] = ๐‘’๐‘—๐œ™๐‘˜ , arbitrary ๐œ™๐‘˜, ๐‘˜ โˆˆ ๐’ฅ pilot0 (74)

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—2๐œ‹๐‘š๐‘–

๐พ ๐’„0[๐‘˜], ๐‘˜ โˆˆ ๐’ฏ๐‘›,๐œ๐‘š , ๐‘– โˆˆ {1, . . . , ๐‘Tx โˆ’ 1} (75)

๐’„๐‘–[๐‘˜] = 0, ๐‘˜ /โˆˆ ๐’ฅ pilot0 , ๐‘– โˆˆ {0, . . . , ๐‘Tx โˆ’ 1}. (76)

If the elements of ๐’ฅ pilot0 are cyclically equi-spaced, the pilots

can also be given by

๐’„0[๐‘˜] = ๐‘’๐‘—๐œ™0,๐‘˜ , ๐‘˜ โˆˆ ๐’ฅ pilot0 (77)

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–๐‘’๐‘—2๐œ‹๐‘˜๐œ๐‘–/๐‘๐’„0[๐‘˜], ๐‘– โˆˆ {1, 2, . . . , ๐‘Tx โˆ’ 1},๐ฟ โ‰ค ๐œ1 โ‰ค ๐ฟ๐‘›, ๐ฟ โ‰ค ๐œ๐‘–+1 โˆ’ ๐œ๐‘– โ‰ค ๐ฟ๐‘› (78)

๐’„๐‘–[๐‘˜] = 0, ๐‘˜ /โˆˆ ๐’ฅ pilot0 , ๐‘– โˆˆ {0, 1, . . . , ๐‘Tx โˆ’ 1} (79)

where {๐œ™๐‘–, ๐œ™0,๐‘˜} are arbitrary phases. The choice of ๐‘‰ = ๐‘Tx

and ๐ฟ๐‘š = ๐ฟ0 requires minimum overhead. Note that for ๐พ =1, the maximum number of transmit antennas this design cansupport is ๐‘

2๐ฟ0โˆ’ 1 since the null design cannot be applied to

๐’ฏ1,0 which contains mirror pairs. However, the antenna ๐‘2๐ฟ0

can transmit pilots on ๐’ฏ1,0 using C-F design.

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2260 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

E. [FDM; Null] Design

This design decouples pilots of different antennas throughFDM and suppresses mirror tone interferences by means ofmirror null tones. It uses ๐พ = 1 symbol with ๐‘Tx๐ฟ๐‘š โ‰ค ๐‘/2,and consists of ๐‘Tx๐ฟ๐‘š constant amplitude pilot tones withthe index set ๐’ฅ pilot

0 = โˆช๐‘–๐’ฅ pilot0,๐‘– and ๐‘Tx๐ฟ๐‘š null tones with

the index set ๐’ฅ null0 . The differences from [CDM-F; Null] are

(i) the constant amplitude pilots can be arbitrary within andacross antennas in this design while they are dependent acrossantennas in [CDM-F; Null] and (ii) a different antenna trans-mits its constant amplitude pilots only on distinct cyclicallyequi-spaced ๐ฟ๐‘š tones in this design while each antenna usesthe same ๐‘Tx๐ฟ๐‘š tones in [CDM-F; Null]. The index sets of[FDM; Null] design are defined by

๐’ฅ pilot0,๐‘– = ๐’ฏ๐‘›,๐œ๐‘– , ๐œ๐‘˜ โˆ•= ๐œ๐‘› if ๐‘˜ โˆ•= ๐‘›, (80)

๐’ฅ null0,๐‘– = ๐’ฅ null

0 = {๐‘ โˆ’ ๐’ฅ pilot0 } (81)

โ„๐‘™ = {0, 1, . . . , ๐‘ โˆ’ 1} โˆ– {๐’ฅ pilot0 โˆช ๐’ฅ null

0 } (82)

where ๐œ๐‘˜ โˆˆ {{1, 2, . . . ,๐‘€๐‘› โˆ’ 1} โˆ– {๐‘€๐‘›/2}}, {๐œ๐‘˜} โˆฉ {๐‘€๐‘› โˆ’๐œ๐‘˜} = โˆ…. The pilot tones are given by

๐’„๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜ , ๐‘˜ โˆˆ ๐’ฅ pilot0,๐‘– , ๐‘– โˆˆ {0, . . . , ๐‘Tx โˆ’ 1} (83)

๐’„๐‘–[๐‘˜] = 0, ๐‘˜ /โˆˆ ๐’ฅ pilot0,๐‘– (84)

where {๐œ™๐‘–,๐‘˜} are arbitrary. For ๐พ = 1, the maximum numberof transmit antennas this design can support is ๐‘

2๐ฟ0โˆ’ 1 which

is the same scenario as discussed in [CDM-F; Null] design.

F. [TFDM; Null / C-F] Design

In this design, pilots of different antennas are decoupled byTFDM design, while intra-antenna mirror tone interferencesare addressed by mirror null tones, and inter-antenna mirrortone interferences are suppressed by code design across fre-quency domain (C-F) or mirror null tones across differentantennas. This design uses ๐พ = 2 symbols over whichnon-zero pilots (with the index set ๐’ฏ๐‘›,๐œ๐‘–) of each antenna๐‘– are spread out evenly. It requires ๐‘Tx๐ฟ๐‘› โ‰ค ๐‘ , and๐ฟ๐‘› โ‰ฅ 2๐ฟ0. The pilot index sets over two symbols for (2๐‘–)thand (2๐‘–+1)th antennas are chosen as ๐’ฏ๐‘›,๐œ2๐‘– and ๐’ฏ๐‘›,๐œ2๐‘–+1 where๐œ๐‘˜ โˆˆ {{1, 2, . . . ,๐‘€๐‘› โˆ’ 1} โˆ– {๐‘€๐‘›/2} and ๐œ2๐‘– + ๐œ2๐‘–+1 = ๐‘€๐‘›.If ๐‘Tx is an odd number, the pilot index set over the twosymbols for the last antenna (๐‘– = ๐‘Tx โˆ’ 1) is ๐’ฏ๐‘›,๐‘€๐‘›/2 whichconsists of mirror pairs excluding self-mirror tones. ๐’ฏ๐‘›,๐œ๐‘˜ isdivided into two decimated sets ๐’ฏ๐‘›,๐œ๐‘˜,๐‘’ and ๐’ฏ๐‘›,๐œ๐‘˜,๐‘œ, consistingof even elements and odd elements (their values can be evenor odd) of ๐’ฏ๐‘›,๐œ๐‘˜ , respectively. For each antenna pairs 2๐‘– and2๐‘– + 1, ๐’ฏ๐‘›,๐œ2๐‘–,๐‘’ and ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘œ form mirror pairs, and so do๐’ฏ๐‘›,๐œ2๐‘–,๐‘œ and ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘’. Antennas 2๐‘– and 2๐‘–+1 transmit pilotson ๐’ฏ๐‘›,๐œ2๐‘–,๐‘’ and ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘œ, respectively, in the first symbol,and ๐’ฏ๐‘›,๐œ2๐‘–,๐‘œ and ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘’, respectively, in the second symbol.Mathematically, the index sets are given by

๐’ฅ pilot0,2๐‘– = ๐’ฏ๐‘›,๐œ2๐‘–,๐‘’, ๐’ฅ pilot

0,2๐‘–+1 = ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘œ, (85)

๐’ฅ pilot1,2๐‘– = ๐’ฏ๐‘›,๐œ2๐‘–,๐‘œ, ๐’ฅ pilot

1,2๐‘–+1 = ๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘’ (86)

๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘œ = {๐‘ โˆ’ ๐’ฏ๐‘›,๐œ2๐‘–,๐‘’}, (87)

๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘’ = {๐‘ โˆ’ ๐’ฏ๐‘›,๐œ2๐‘–,๐‘œ} (88)

๐’ฅ pilot0,๐‘Txโˆ’1 = ๐’ฏ๐‘›,๐‘€๐‘›/2,๐‘’ & ๐’ฅ pilot

1,๐‘Txโˆ’1 = ๐’ฏ๐‘›,๐‘€๐‘›/2,๐‘œ, odd ๐‘Tx

(89)

๐’ฏ๐‘›,๐‘Txโˆ’1,๐‘’ = {๐‘ โˆ’ ๐’ฏ๐‘›,๐‘Txโˆ’1,๐‘œ}, odd ๐‘Tx (90)

โ„๐‘™ = {0, . . . , ๐‘ โˆ’ 1} โˆ– ๐’ฅ pilot๐‘™ , even ๐‘Tx (91)

โ„๐‘™ = {0, . . . , ๐‘ โˆ’ 1} โˆ– {๐’ฅ pilot๐‘™ โˆช ๐’ฏ๐‘›,๐‘Txโˆ’1}, odd ๐‘Tx.

(92)

For each antenna pair (2๐‘–, 2๐‘–+1), the C-F design is given by

๐‘2๐‘–,0[[๐’ฏ๐‘›,๐œ2๐‘–,๐‘’]๐‘š] = ๐‘’๐‘—๐œ™0,๐‘š ,๐‘š = 0, 1, . . . , ๐ฟ๐‘›โˆ’1 โˆ’ 1 (93)

๐‘2๐‘–+1,0[[๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘œ]๐ฟ๐‘›โˆ’1โˆ’๐‘š] = ๐‘’๐‘—2๐œ‹๐œ†2๐‘š/๐ฟ๐‘› ๐‘’โˆ’๐‘—๐œ™0,๐‘š ,

๐œ† โˆˆ {๐ฟ,๐ฟ+ 1, . . . , ๐ฟ๐‘›โˆ’1 โˆ’ ๐ฟ} (94)

๐‘2๐‘–,1[[๐’ฏ๐‘›,๐œ2๐‘–,๐‘œ]๐‘š] = ๐‘’๐‘—๐œ™1,๐‘š ,๐‘š = 0, 1, . . . , ๐ฟ๐‘›โˆ’1 โˆ’ 1 (95)

๐‘2๐‘–+1,1[[๐’ฏ๐‘›,๐œ2๐‘–+1,๐‘’]๐ฟ๐‘›โˆ’1โˆ’๐‘š] = ๐‘’๐‘—2๐œ‹๐›ผ(2๐‘š+1)/๐ฟ๐‘› ๐‘’โˆ’๐‘—๐œ™1,๐‘š ,

๐›ผ โˆˆ {๐ฟ,๐ฟ+ 1, . . . , ๐ฟ๐‘›โˆ’1 โˆ’ ๐ฟ} (96)

where {๐œ™0,๐‘š} and {๐œ™1,๐‘š} are arbitrary. For different antennapairs, {๐œ™0,๐‘š}, {๐œ™1,๐‘š}, ๐œ†, and ๐›ผ can be independently chosen.For an odd ๐‘Tx, ๐‘๐‘Txโˆ’1,0[๐‘˜ โˆˆ ๐’ฏ๐‘›,๐‘€๐‘›/2,๐‘œ] and ๐‘๐‘Txโˆ’1,1[๐‘˜ โˆˆ๐’ฏ๐‘›,๐‘€๐‘›/2,๐‘’] can be set to arbitrary unit amplitude symbols. Ateach of the two symbols, this design uses ๐‘Tx๐ฟ๐‘›โˆ’1 pilottones for an even ๐‘Tx and (๐‘Tx โˆ’ 1)๐ฟ๐‘›โˆ’1 + ๐ฟ๐‘› pilot tones(including null tones) for an odd ๐‘Tx.

G. [CDM-F or FDM; C-T] Design

This design uses two OFDM symbols. In CDM-F, each an-tenna transmits ๐‘Tx๐ฟ๐‘› constant-amplitude pilot tones in eachsymbol, while in FDM each antenna transmits ๐ฟ๐‘› constant-amplitude pilot tones and (๐‘Txโˆ’1)๐ฟ๐‘› null pilot tones in eachsymbol. These pilot indexes are all mirror-pairs (may includeself-mirror tones). The index sets are given by

๐’ฅ๐‘™ =

๐‘Txโˆ’1โˆช๐‘š=0

๐’ฏ๐‘›,๐œ๐‘š , ๐œ๐‘š โˆˆ {0, 1, . . . ,๐‘€๐‘› โˆ’ 1}, (97)

โ„๐‘™ = {0, 1, 2, . . . , ๐‘ โˆ’ 1} โˆ– ๐’ฅ๐‘™, (98)

๐’ฅ๐‘™,๐‘– =

{ ๐’ฅ๐‘™, CDM-F๐’ฅ๐‘™,๐‘– = ๐’ฏ๐‘›,๐œ๐‘– , FDM

(99)

where ๐œ๐‘š โˆ•= ๐œ๐‘˜ if ๐‘š โˆ•= ๐‘˜ and {๐œ๐‘š} = {(๐‘€๐‘› โˆ’ ๐œ๐‘š)๐‘€๐‘›}.The choice of ๐‘› = 0 in ๐ฟ๐‘› and ๐‘€๐‘› yields minimum pilot

overhead. The pilots from different antennas are decoupledby CDM-F or FDM, while the mirror tone interferences aresuppressed by C-T over two symbols. For CDM-F, eachantenna transmits constant amplitude pilots on ๐’ฅ๐‘™. For FDM,the ๐‘šth antenna transmits constant amplitude pilots on ๐’ฏ๐‘›,๐œ๐‘šand null tones on {๐’ฅ๐‘™ โˆ– ๐’ฏ๐‘›,๐œ๐‘š}. The C-T design is describedby the relationship of the pilot vectors at the second symbol tothose at the first symbol. For CDM-F, the pilot vectors at thesecond symbol are just

โˆšโˆ’1 times the corresponding pilotvectors at the first symbol. For FDM, we can have severalapproaches such as: (i) the pilot vectors at the second symbolare

โˆšโˆ’1 times those at the first symbol, (ii) the pilot toneswith indexes less than ๐‘/2 remain the same over two symbols,while the remaining pilots change polarities across the twosymbols, or (iii) for each antenna pair with mirror indexsets ๐’ฏ๐‘›,๐œ๐‘š and ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ , the antenna using ๐’ฏ๐‘›,๐œ๐‘š transmits thesame pilots on both symbols, while the other antenna changespolarities of pilots across the two symbols.

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H. [TFDM / CDM-T; Null / C-T] Design

In this design, pilots from different antennas are decoupledthrough TFDM and CDM-T, while mirror tone interferenceswithin each antenna are addressed by mirror null tones,and those across antennas are suppressed by mirror nulltones and C-T. The antennas are divided into ๐พ๐‘” groups.Pilot tones of different groups are disjoint via TFDM (orTDM). Each antenna group ๐‘š (for an even ๐‘Tx) consistsof an even number of antennas, say 2๐‘๐‘š, and uses ๐’ฏ๐‘›,๐œ๐‘šand ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ where ๐œ๐‘š โˆˆ {{1, 2, . . . ,๐‘€๐‘› โˆ’ 1} โˆ– {๐‘€๐‘›/2}},๐œ๐‘š โˆ•= ๐œ๐‘˜ if ๐‘š โˆ•= ๐‘˜, ๏ฟฝฬ„๏ฟฝ = ๐‘€๐‘› โˆ’ ๐‘š, ๐œ๐‘š + ๐œ๏ฟฝฬ„๏ฟฝ = ๐‘€๐‘›, and๐‘› โˆˆ {1, . . . , log2(๐‘/(2๐พ๐‘”๐ฟ0)}. ๐’ฏ๐‘›,๐œ๐‘š and ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ are eachdivided into two subsets of the same cardinality, denoted by๐’ฏ๐‘›,๐œ๐‘š,0, ๐’ฏ๐‘›,๐œ๐‘š,1, ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,0, and ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,1, such that ๐’ฏ๐‘›,๐œ๐‘š,0 and๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,1 form mirror pairs and so do ๐’ฏ๐‘›,๐œ๐‘š,1 and ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,0. Thisdivision need not be a splitting of even and odd elements asrequired in Section IV-F.

The first half of the antennas within group ๐‘š transmiton ๐’ฏ๐‘›,๐œ๐‘š,0 during the first ๐‘„๐‘š (โ‰ฅ ๐‘๐‘š) symbols and on๐’ฏ๐‘›,๐œ๐‘š,1 during the next ๐‘„๐‘š symbols. The other half of theantennas transmit on ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,0 during the first ๐‘„๐‘š symbols andon ๐’ฏ๐‘›,๐œ๏ฟฝฬ„๏ฟฝ,1 during the next ๐‘„๐‘š symbols. These two subsets ofantennas within each group are of TFDM type. The pilot forantenna ๐‘– of group ๐‘š on corresponding subcarrier [๐’ฏ๐‘›,โˆ—,โˆ—]๐‘˜ at๐‘™th symbol is given by ๐‘’๐‘—๐œ™๐‘–,๐‘˜๐‘’๐‘—2๐œ‹๐‘™๐‘–/๐‘„๐‘š๐‘Ž๐‘š,๐‘™,๐‘˜ where {๐œ™๐‘–,๐‘˜}are arbitrary and {๐‘Ž๐‘š,๐‘™,๐‘˜} are arbitrary constant amplitudesymbols, i.e., antennas transmitting on the same subcarrierfollow CDM-T design. Each group ๐‘š requires 2๐ฟ๐‘› tones over2๐‘๐‘š symbols, and setting ๐‘› = 0 in ๐ฟ๐‘› and ๐‘€๐‘› yields asmaller overhead.

For an odd ๐‘Tx, a dummy antenna can be fictitiouslyadded in the design to have an even ๐‘Tx. Two more-efficientalternatives are described below where pilots for the first๐‘Txโˆ’1 (even) antennas are developed according to the above-mentioned design. In the first alternative, the last antennatransmits pilots on ๐’ฏ๐‘›,0 with ๐‘› โ‰ฅ 1 over one symbol using C-F design. In the second alternative, the last antenna transmitsarbitrary constant amplitude pilots on ๐’ฏ๐‘›,๐‘€๐‘›+1 with ๐‘› โ‰ฅ 0over two symbols using C-T design, or on ๐’ฏ๐‘›,๐‘€๐‘›+1,๐‘’ with๐‘› โ‰ฅ 0 at the first symbol and on ๐’ฏ๐‘›,๐‘€๐‘›+1,๐‘œ at the secondsymbol (i.e., Null design over two symbols).

I. [TDM; Null] Design

In this design, pilots of different antennas are decoupled bymeans of TDM while mirror tone interferences are suppressedby means of null tones. All antennas transmit on the sameset of subcarriers, but each antenna transmits during a dif-ferent OFDM symbol. Thus, ๐พ = ๐‘Tx OFDM symbols arerequired, while data can be transmitted on other subcarriersin a pilot-data-multiplexed format. This design can handlea larger number of null guard tones than other designs.Let ๐’ฅ guard denote the tone index set for the null guardtones. First, obtain ๐’ฏ๐‘›,๐‘š such that {๐’ฏ๐‘›,๐‘š โˆฉ ๐’ฅ guard} = โˆ…,๐‘š โˆˆ {{1, 2, . . . ,๐‘€๐‘›โˆ’1}โˆ–{๐‘€๐‘›/2}}. With ๏ฟฝฬ„๏ฟฝ โ‰œ ๐‘€๐‘›โˆ’๐‘š, wehave ๐’ฏ๐‘›,๏ฟฝฬ„๏ฟฝ = {๐‘ โˆ’ ๐’ฏ๐‘›,๐‘š modulo ๐‘}. Then, the index sets

are given by

๐’ฅ pilot๐‘–,๐‘™ =

{ ๐’ฏ๐‘›,๐‘š, if ๐‘– = ๐‘™โˆ…, else

(100)

๐’ฅ null๐‘–,๐‘™ =

{ ๐’ฏ๐‘›,๏ฟฝฬ„๏ฟฝ, if ๐‘– = ๐‘™๐’ฏ๐‘›,๐‘š โˆช ๐’ฏ๐‘›,๏ฟฝฬ„๏ฟฝ, else

(101)

โ„๐‘™ = {0, 1, 2, . . . , ๐‘ โˆ’ 1} โˆ– {๐’ฏ๐‘›,๐‘š โˆช ๐’ฏ๐‘›,๏ฟฝฬ„๏ฟฝ โˆช ๐’ฅ guard}. (102)

The non-zero pilot tones can have arbitrary phases {๐œ™๐‘–,๐‘˜} as

๐’„๐‘–,๐‘–[๐‘˜] = ๐‘’๐‘—๐œ™๐‘–,๐‘˜ , ๐‘˜ โˆˆ ๐’ฅ pilot๐‘–,๐‘– , ๐‘– โˆˆ {0, 1, . . . , ๐‘Tx โˆ’ 1}. (103)

For minimum overhead and operability with largest numberof null guard tones, ๐‘€๐‘› should be set to ๐‘€0.

J. Other Designs

Other variations or combinations of the above designs arealso possible. For example, [CDM-F; C-T] and [FDM; C-T] designs can be combined as [CDM-F/FDM; C-T] whereantennas are divided into two groups such that there is nomirror pair across the two groups, and the first group applies[CDM-F; C-T] while the other group uses [FDM; C-T]. Othercombinations may also be possible, but for practical designsimplicity we skip further investigation in this direction.

V. SIMULATION RESULTS AND DISCUSSIONS

A. Parameter Setting

System parameters in the simulation are ๐‘Tx = 2, ๐‘Rx =2, ๐‘ = 64, 6 left and 6 right null guard tones, ๐‘€ -ary QAMwith ๐‘€ = 16, and a Rayleigh fading channel having anexponential power delay profile (3 dB per tap decay factor)with 4 sample-spaced taps. The FI I/Q imbalances are set to๐›ผ =

๐‘Ž๐ผ๐‘ก

๐‘Ž๐‘„๐‘ก

=๐‘Ž๐ผ๐‘Ÿ

๐‘Ž๐‘„๐‘Ÿ= 1.09648 (= 0.4 dB), and ฮ”๐œƒ = ๐œƒ๐ผ๐‘ก โˆ’ ๐œƒ๐‘„๐‘ก

= ๐œƒ๐ผ๐‘Ÿ โˆ’ ๐œƒ๐‘„๐‘Ÿ = 3 โˆ˜ 2. The FD I/Q imbalances are modeledby 3-tap filters (hence, ๐ฟ = 8) with discrete-time impulseresponses of [0.01, 0.9999, 0.01] and [0.015, 0.9998, 0.01] forthe transmit I and Q branches, and [0.012, 0.9997, 0.018],and [0.01, 0.9997, 0.02] for the receive I and Q branches.For performance comparison, we use the pilot design from[21] as SISO Reference. We use the design from [24] asMIMO Reference 1 and an FDM design from [11]3 as MIMOReference 2. In all methods, the estimators from Eqns. (16)and (17) are utilized and for BER results the maximumlikelihood (ML) detection (joint detection of mirror tones) isapplied. The energies of a non-zero pilot tone in the SISOreferences and the MIMO reference 1 are set to be the sameas the average bit energy of data. The total pilot energy iskept the same for all pilot designs in each case (SISO orMIMO). For the SISO results, the system parameters are thesame except for the number of antennas.

2These values are within typical ranges (e.g., see [4], [22], [27], [28]).3To illustrate the performance degradation when I/Q imbalance is not

considered in the pilot design, we pick a particular FDM design. Some ofthe pilot designs from [11] with 2๐‘Tx๐ฟ0 pilot tones may yield the sameestimation performance as the proposed ones.

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2262 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

5 10 15 20 25

10โˆ’3

10โˆ’2

(EK/N)/N

0 (dB)

Cha

nnel

Est

imat

ion

MS

E

Proposed (Theo.)SISO Ref. (Theo.)Proposed (Simu.)SISO Ref. (Simu.)

Fig. 3. Channel estimation MSE comparison of different pilot designs in aSISO OFDM system.

2 4 6 8 10 12 14

0.001

0.002

# of null guard tones

Cha

nnel

Est

imat

ion

MS

E

Proposed MIMO

Proposed SISO

MIMO Ref. 1

SISO Ref.

Fig. 4. Channel estimation MSE comparison of different pilot designs whilevarying the number of guard tones, ((๐ธ๐พ/๐‘)/๐‘0 = 21 dB)

B. Estimation and BER Performance

Fig. 3 shows the channel estimation MSEs (MSE๐’‘+MSE๐’’)from simulation and the theoretical MSEs ((20) for the ref-erence design and (30) for the proposed design) for a SISOsystem. The proposed design outperforms the SISO referencewhich experiences a flooring effect at high SNR. The nullingof some of the subcarriers results in a breakdown of the codingused in [21] to eliminate the mirror tone interferences anddegrades the channel estimation MSE. The theoretical MSEsmatch the simulation MSEs very well.

The effects of guard tones on the pilot designs are illustratedin Fig. 4 using the theoretical MSEs. The MSE degradationof the SISO reference pilot design is observed to be moresensitive to the number of null guard tones than the MIMOReference 1 which is due to different coding strategies adoptedin the pilot designs. The larger MSE level of MIMO pilot

10 15 20 25 3010

โˆ’4

10โˆ’3

10โˆ’2

10โˆ’1

100

(EK/N)/N

0 (dB)

Cha

nnel

Est

imat

ion

MS

E

Proposed (Theo.)

MIMO Ref. 1 (Theo.)

MIMO Ref. 2 (Theo.)

Proposed (Simu.)

MIMO Ref. 1 (Simu.)

MIMO Ref. 2 (Simu.)

Proposed (CRB)

Fig. 5. Channel estimation MSE comparison of different pilot designs in a2ร— 2 MIMO OFDM system.

TABLE IIIPILOT OVERHEAD COMPARISON

Design Overhead(# of Tones)

(MIMO) [TDM; TD/C-F] 2๐‘Tx๐ฟ0

(MIMO) [CDM-T; C-T] 2๐‘Tx๐ฟ0

(MIMO) [CDMF; Null] 2๐‘Tx๐ฟ0

(MIMO) [FDM; Null] 2๐‘Tx๐ฟ0

(MIMO) [TFDM; Null/C-F] 4๐ฟ0โŒˆ๐‘Tx/2โŒ‰(MIMO) [TFDM/CDM-T; Null/C-T] 2๐‘Tx๐ฟ0

(MIMO) [TDM; Null] 2๐‘Tx๐ฟ0

(MIMO) Design from [24] 2๐‘Tx๐‘(SISO) Design from [21] ๐‘

designs if compared to the SISO designs is due to the factthat the MSE presented is the sum of MSEs of all parametersunder estimation and that MSE increases with the number ofparameters under estimation.

Fig. 5 presents the channel estimation MSEs from sim-ulation, the theoretical MSEs, and the CRB from (31) forthe MIMO system. The proposed design outperforms bothreference designs. Although MIMO Reference 2 gives optimalestimation performance for systems without I/Q imbalances,it causes a substantial performance degradation in the pres-ence of I/Q impairments. MIMO Reference 1 has a slightperformance loss due to its non-optimality for systems withguard bands. The theoretical MSEs in Section III give an exactmatch to the simulation MSE results which show no noticeabledifference from the CRB. This is due to the fact that the mildfrequency-selectivity of the I/Q imbalance makes the whitenoise assumption used in the pilot design valid, and then theestimators become ML with the Gaussian signal model whichare known to approach the CRB.

In Fig. 6, the (uncoded) BER performances of differentpilot designs are compared for the MIMO system. Similarconclusions for the MSE performance apply for the BERperformance. Although the BER improvement is small, theproposed pilot designs reduce the pilot overhead greatly (see

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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2263

5 10 15 20 2510

โˆ’5

10โˆ’4

10โˆ’3

10โˆ’2

10โˆ’1

100

Eb/N

0 (dB)

BE

R

Proposed

MIMO Ref. 1

MIMO Ref. 2

No IQ

No IQ, Smart Rx

Fig. 6. BER comparison of different pilot designs for 16-QAM in a 2 ร— 2MIMO OFDM system.

Table III) which increases the data capacity/throughput, andcan also reduce the data detection latency if compared toMIMO Reference 1. When the receiver does not know theabsence of I/Q imbalance, its performance is the same asthe cases with I/Q imbalance. But when the receiver knowsthe absence of I/Q imbalance (denoted as the Smart Rx), itsperformance is better than those with I/Q imbalance exceptat high SNR where the ML receiver gains frequency diversityprovided by the I/Q imbalance.

Fig. 7 presents the MSE performance of the proposedpilot design under various FI I/Q imbalance levels. The low,medium, and high I/Q imbalance cases, respectively, corre-spond to (๐›ผ = 0.1 dB, ฮ”๐œƒ = 1โˆ˜), (๐›ผ = 0.7 dB, ฮ”๐œƒ = 10โˆ˜),and (๐›ผ = 2 dB, ฮ”๐œƒ = 30โˆ˜) 4. The no I/Q imbalance caserefers to (๐›ผ = 0 dB, ฮ”๐œƒ = 0โˆ˜) and without FD I/Q imbalance,while the receiver still considers there is I/Q imbalance andestimates in the same way as in the other cases. For the SmartRx case, the MSE drops proportional to the number of tapsestimated. From Fig. 7, it is clear that the channel estimationMSE is not affected by the FI I/Q imbalance level.

In Fig. 8, to show the effect of different levels of FDI/Q imbalance, we have evaluated an additional set of FDI/Q imbalance filters with impulse responses of [0.9999, 0.01]and [0.9999, 0.015] for the transmit I and Q branches, and[0.9998, 0.018], and [0.9999, 0.01] for the receive I and Qbranches. This corresponds to an equivalent channel with๐ฟ = 6 taps. We have also included the performance of theSmart Rx where the number of parameters under estimationis ๐ฟ = 4 as opposed to 2๐ฟ = 12 and 2๐ฟ = 16 in theother two cases. The MSE gaps are due to the differencesin the numbers of parameters under estimation. Hence, thefrequency-selectivity level of I/Q imbalance (more specifically,the lengths of the equivalent direct and mirror channels) canaffect the total channel estimation MSE.

4The values for the high case are outside of the current expected rangesfor hardware in use today (see [27], [28]). However, as semiconductordownscaling continues, higher values for I/Q imbalance such as these couldbe seen.

12 14 16 18 20 22 24 26 28 30 32

10โˆ’4

10โˆ’3

10โˆ’2

(EK/N)/N

0 (dB)

Cha

nnel

Est

imat

ion

MS

E

No IQ Im.(Theo)With IQ Im. (Theo)No IQ Im. (Simu)Low IQ Im. (Simu)Med IQ Im. (Simu)High IQ Im. (Simu)No IQ, Smart Rx (Simu)

Fig. 7. Effects of different levels of FI I/Q imbalance on a proposed pilotdesign in a 2ร— 2 MIMO OFDM system.

15 20 25 30

10โˆ’4

10โˆ’3

10โˆ’2

(EK/N)/N

0 (dB)

Cha

nnel

Est

imat

ion

MS

E

L = 8 (Theo.)L = 6 (Theo.)L = 4, Smart Rx (Theo.)L = 8 (Simu.)L = 6 (Simu.)L = 4, Smart Rx (Simu.)

Fig. 8. Effects of different levels of FD I/Q imbalance on a proposed pilotdesign in a 2ร— 2 MIMO OFDM system.

C. Comparison in Other Aspects

All of the proposed designs give the same estimation perfor-mance since they meet the criteria defined by (24). However, ifother system constraints or impairments are considered, therecould be advantages of one over another. For example, somedesigns such as [CDM-T; C-T] and [TDM; Null] are betterequipped to facilitate a larger null guard band than otherssince the gaps between used subcarriers is larger. However,they utilize more OFDM symbols, and thus would introducemore latency at the receiver.

The proposed designs can be applied to preamble as wellas pilot-data-multiplexed symbols because the design problemwas formulated so. This reduces the pilot overhead and thelatency at the receiver. The existing designs in [15], [16],[18]โ€“[20], [22], [24] only apply for preamble since no datatones are considered in the designs. The proposed designsare applicable for both SISO and MIMO OFDM systemswhile the existing methods such as [15]โ€“[22] only address forSISO OFDM systems. In term of the estimation performance,

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2264 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010

the proposed designs yield white-noise optimality and closeto ideal-optimal results for colored noise at the demodulatoroutput. The existing methods such as [19], [21], [22] onlyprovide a suboptimal estimation performance which degradesfor systems with guard bands while the proposed designscan overcome this shortfall as well. The comparison of pilotoverhead for several pilot designs is presented in Table III.Under the considered system setting, the proposed designsrequire 1/4 of the overhead of the SISO reference design,and 1/8 of that of the MIMO Reference 1.

VI. CONCLUSIONS

We have developed efficient pilot designs for estimationof the equivalent channel responses incorporating FI and FDtransmitter and receiver I/Q imbalances in MIMO OFDMsystems. The receive filter output noise samples in the pres-ence of FD receiver I/Q imbalance are colored and generallyunknown, and hence development of exactly-optimal pilotdesigns is impractical. However, in practice the frequency-selectivity of I/Q imbalance is very small and hence our pilotdesigns developed based on the white noise condition areobserved to yield essentially the same performance as theCRB. To be applicable in the pilot-data-multiplexed format,not only the data and pilot tones need to be disjoint butalso each type (pilot or data) should occupy only on mirrorsubcarrier pairs. The minimum number of pilot tones requiredfor the considered estimators is at least doubled if compared tothe systems without I/Q imbalance. To suppress inter-antennainterferences, the pilots of different transmit antennas need tosatisfy the condition in (42) as required in systems without I/Qimbalance but also an additional condition due to the mirrortone interference as given in (47).

We have also observed that the frequency-selectivity levelof I/Q imbalance can affect the total channel estimation MSEwhile the FI I/Q imbalance level does not. The proposed pilotdesigns are more efficient than the existing designs in termsof overhead, estimation performance, and general applicability(preamble or pilot-data-multiplexed setup, SISO or MIMO,with or without guardbands). Our MIMO pilot designs canalso be extended to OFDMA downlink systems and OFDM-based cooperative communication systems. For OFDMA up-links, the applicability of the proposed pilot designs dependson how resources are channelized for different users.

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