2252 ieee transactions on communications, vol. 58, โฆhxm025000/pilotiqtcom2010aug.pdfminn and...
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2252 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
Pilot Designs for Channel Estimation ofMIMO OFDM Systems with
Frequency-Dependent I/Q ImbalancesHlaing Minn, Senior Member, IEEE, and Daniel Munoz, Student Member, IEEE
AbstractโMultiple input multiple output (MIMO) orthogonalfrequency division multiplexing (OFDM) systems facilitate highdata rate wireless communications, and require reliable channelestimates to fully materialize their advantages. The semicon-ductor downscaling trend has exacerbated device impairmentssuch as inphase and quadrature (I/Q) imbalances which causeinter-carrier interferences in OFDM systems which cannot beremedied by increasing signal power. Different RF chains ofMIMO branches can cause different I/Q imbalances whichfurther complicates MIMO OFDM channel estimation. Thispaper proposes several pilot designs for the estimation of thecombined responses of MIMO frequency-selective channels andfrequency-dependent I/Q imbalances. The proposed designs re-quire much smaller pilot overhead than the existing designs, andalso provide estimation mean-squares error optimality (underwhite noise) and general applicability to preamble as well aspilot-data-multiplexed symbols in MIMO systems with or withoutnull guard tones. Performance analyses and simulation resultscorroborate advantages of the proposed designs.
Index TermsโChannel estimation, I/Q imbalance, MIMO,OFDM, pilot design.
I. INTRODUCTION
THE advances in semiconductor down-scaling have en-abled a substantial growth of signal processing power
but exacerbated the variations of the device characteristicsdue to more difficult control in the fabrication process [1].The inphase (I) and quadrature (Q) imbalance, which repre-sents mismatch between the I and Q branches, is a majorimpairment especially for direct-conversion transceivers witha high modulation order [2]. The higher data rate of next-generation wireless systems in a typical bandwidth limitedregime requires ๐ -ary quadrature amplitude modulation (๐ -QAM) with a large ๐ . Both the semiconductor down-scalingtrend and the need of a high modulation order underline theimportance of I/Q imbalance issue for next generation wirelesssystems.
A viable transmission technology for current and next gen-eration wireless systems is OFDM, while MIMO systems have
Paper approved by G. Bauch, the Editor for MIMO, Coding and Relayingof the IEEE Communications Society. Manuscript received March 12, 2009;revised July 23, 2009 and November 9, 2009.
The result for SISO systems was presented at IEEE WCNC 2009, the[TDM; Null] design at IEEE VTC 2009 (Fall), and [CDM-F; Null], [FDM;Null], [CDM-T; C-T] without Self-Mirror Tones, and [CDM-F or FDM; C-T]designs at IEEE Globecom 2009.
The authors are with the Dept. of Electrical Engineering, Universityof Texas at Dallas, Richardson, TX 75083 USA (e-mail: {hlaing.minn,djm072000}@utdallas.edu).
Digital Object Identifier 10.1109/TCOMM.2010.08.090150
become instrumental in providing higher data rate and betterreliability in diverse wireless channels. The I/Q impairment inOFDM causes inter-carrier interferences. In MIMO OFDM,due to different RF chains for different antennas, I/Q imbal-ances may differ across antennas as well as across frequency(especially for ultra-wideband systems) for each antenna. Fordata detection, the multitudes of I/Q imbalance and channelparameters or their combined responses need to be estimatedand compensated (see [3]โ[9] for representative I/Q imbalancecompensation methods). An efficient and practical way for theestimation is to transmit pilot tones. The main issue is how todesign these pilots while considering the interferences amongmirror tones and different transmit antennas as well as theestimation performance and overhead. The pilot designs maydepend on whether they are developed for separate estimationof I/Q imbalances and channel responses or estimation of thecombined responses. We consider the latter.
There are several pilot or training signal designs for MIMOOFDM channel estimation without I/Q imbalance (e.g., [10]โ[14]). There also exist several pilot designs for I/Q imbalanceestimation. The work in [15] applies two OFDM trainingsymbols to perform per-subcarrier estimation of frequency-dependent (FD) I/Q imbalance in single input single output(SISO) systems. The first training symbol has null tones at allnegative subcarrier indexes and the second symbol containsnull tones at all positive subcarrier indexes. This methodneither optimizes the pilot overhead (large overhead) norconsiders pilot-data-multiplexed symbols, MIMO OFDM, andexploitation of frequency-domain correlation. Similar draw-backs apply to [16] except it considers (mainly frequency-independent (FI)) transmitter I/Q imbalance only, and usesa different pilot design. It uses an even number of OFDMtraining symbols with non-zero pilots where the pilots at thenegative (positive) subcarrier indexes of the even symbols arethe same as (negatives of) the corresponding pilots at the oddtraining symbols. The same pilot design using two OFDMtraining symbols is applied in [17] for the FI receiver I/Qimbalance, but [17] requires an additional training symbol forchannel estimation, resulting in a larger overhead. It considerspilot-data-multiplexed symbols, but the other drawbacks stillhold. [18] applies a pilot design (a combination of the designsin [15] and [16]) which uses an even number of OFDMtraining symbols with null pilots at the negative subcarrierindexes of the first half of OFDM training symbols and atthe positive subcarrier indexes of the second half of OFDM
0090-6778/10$25.00 cโ 2010 IEEE
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2253
training symbols. It considers FI receiver I/Q imbalance only,and is also associated with the above drawbacks.
For estimating the frequency-domain combined responsesof the channel and FD I/Q imbalances, [19] presents two pilotdesigns using two OFDM training symbols. In the first design,the second OFDM training symbol is the 90 degree phaserotated version of the first symbol. The second design is thesame as the one in [16]. It also provides a design conditionfor multiple OFDM symbols. But it uses a large overhead,addresses neither pilot-data-multiplexed symbols nor MIMOOFDM, and does not exploit frequency-domain correlation.
In [20], the authors consider a time-domain estimationwhich exploits frequency-domain correlation. It transmitsequal-amplitude pilots on all subcarriers of one OFDM sym-bol. The mirror-tone interference is cancelled by designing thesum of the phases of the ๐th pilot and its mirror tone to be ๐๐.In this design the phases of the pilots at tone indexes 0 and๐/2 cannot be arbitrary as opposed to the statement in [20].Moreover, when null guard tones are added at the band edgesas in [21], the estimation optimality is no longer maintained.It does not address for pilot-data-multiplexed symbols andMIMO OFDM systems, and uses a large overhead. The pilotdesign in [22] uses all subcarriers of an OFDM symbol forpilots (large overhead), and is not applicable to systems witha pilot-data multiplexed format and/or with null guard tones.It only considers FI I/Q imbalance and requires a two stepapproach of estimation of I/Q parameters and compensation.
All of the above pilot designs for I/Q imbalance addressfor SISO OFDM systems only. Recently, [23] proposes a pilotdesign for the transmitter and receiver FI I/Q imbalances inMIMO-OFDM systems. Its extension to the FD I/Q imbalanceis considered in [24]. Their pilot design uses the design in [16]with two OFDM training symbols as a basic block which isrepeated according to Hadamard sequences for multiple trans-mit antennas. For ๐Tx transmit antennas, they require 2๐Tx
OFDM training symbols, and hence a large pilot overhead.Moreover, they are not applicable to pilot-data-multiplexedsymbols, and do not provide the estimation optimality sincethe frequency-domain correlation is not exploited. A pilotdesign for space-time block coded systems in single-carrierfrequency-flat fading channels is recently proposed in [25].But it uses a large pilot overhead (at least two space-timecode blocks), lacks the estimation optimality, and cannot beapplied to systems with more than two transmit antennas.
In brief, the existing pilot designs for MIMO OFDM chan-nel estimation do not consider the I/Q imbalance effects, whilethe existing pilot designs for the I/Q imbalance estimationdo not address the channel estimation optimality. All of thelatter designs require large pilot overheads, and most ofthem cannot be applied to MIMO systems and pilot-data-multiplexed symbols. In contrast, in this paper, we proposeseveral pilot designs for MIMO OFDM equivalent channelestimation in the presence of transmitter and receiver FDI/Q imbalances, which avoid the drawbacks of the existingmethods. Our designs require much less pilot overhead, andare more general and white-noise optimal in estimation.
The paper is organized as follows. Section II describes thesignal model and Section III presents pilot design criteria. InSection IV, we study the relationships of the design criteria to
)(tw
Equivalent Lowpass SISO System
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][krI)(tgIr
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nI nTtns )()(
Itt
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)(tg It
)2cos( Itc
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n
nTtns )()(Q
)(bp thChannel
)(tgQt
Bandpass SISO System
Channel)(th
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)(tgDT
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Fig. 1. Bandpass and equivalent lowpass SISO system models.
pilot symbols and structures, and propose several pilot designs.Performance evaluation and simulation results are discussed inSection V, and conclusions are provided in Section VI.
Notation: A bold capital (small) letter represents a matrix(a column vector). The superscripts โ, ๐ , and ๐ป representthe conjugate, the transpose, and the conjugate transposeoperations, respectively. [๐ฟ ]๐,๐ denotes the ๐-th row, ๐-thcolumn element of ๐ฟ . All indices start from 0. The all-one (all-zero) square matrix of size-๐, the ๐ ร ๐ all-zeromatrix, and the ๐ร ๐ identity matrix are denoted by 1๐ (0๐),0๐ร๐, and ๐ฐ๐, respectively. Tr[๐ฟ] denotes the trace of ๐ฟ .diag{๐} represents a diagonal matrix whose diagonal elementsare defined by ๐. The ๐-cyclic-forward-shifted version of ๐is denoted by ๐(๐). * and โ denote the convolution and theKronecker product, respectively. (๐)๐ represents ๐ modulo ๐ .โ๐โ denotes the integer part of ๐ while โ๐โ represents thesmallest integer greater than or equal to ๐ . ๐ญ denotes the๐ -point unitary discrete Fourier transform matrix and ๐๐ isthe ๐-th column of ๐ญ . The subcarrier indexes are modulo ๐where ๐ is the total number of OFDM subcarriers. We willoften use โ๐ to denote the subcarrier index (โ๐)๐ , and & torepresent the logical โandโ operation.
II. SIGNAL MODEL
First, consider a single antenna system with FI and FD I/Qimbalances at both transmitter and receiver sides as shown inFig. 1 where {๐๐ผ๐ก , ๐๐๐ก } and {๐๐ผ๐ก , ๐๐๐ก } represent the FI gainsand phase offsets of the I and Q branches at the transmitter(see [1]). The equivalent pulse shaping filters (i.e., the overallimpulse responses including DAC, amplifier, pulse shaping,and FD I/Q imbalances) for the I and Q branches of thetransmitter are denoted by ๐๐ผ๐ก (๐ก) and ๐๐๐ก (๐ก). These filtersinclude effects of the FD I/Q imbalances [4], [21]. The receiverside parameters are defined similarly with the subscript ๐กreplaced by ๐. An equivalent low-pass system is also presentedin Fig. 1. The channel impulse response โ(๐ก) is the low-pass-equivalent of the bandpass channel โbp(๐ก). The transmitsystem with I/Q imbalance can be viewed as the summationof two systems namely the direct system whose input is thesame as the transmitter input signal ๐ (๐ก) = ๐ ๐ผ(๐ก)+ ๐๐ ๐(๐ก) andthe mirror system whose input is ๐ โ(๐ก). The impulse responsesof the direct and mirror systems at the transmitter are denotedby ๐๐ท๐ (๐ก) and ๐๐๐ (๐ก), and those at the receiver are represented
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2254 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
][krj
][knj
1
,0
Tx
][N
illlj ky
)(tpij
)(tqij( )*
ni nTtns )()(
][kyij
AWGN
( )* )(, tgMjR
)(, tgD jR)(tw
Fig. 2. A simplified equivalent lowpass MIMO system model.
by ๐๐ท๐ (๐ก) and ๐๐๐ (๐ก). Then, they can be given as
๐๐ท๐ (๐ก) =1
2[๐๐ผ๐ก ๐
๐๐๐ผ๐ก ๐๐ผ๐ก (๐ก) + ๐๐๐ก ๐
๐๐๐๐ก ๐๐๐ก (๐ก)] (1)
๐๐๐ (๐ก) =1
2[๐๐ผ๐ก ๐
๐๐๐ผ๐ก ๐๐ผ๐ก (๐ก)โ ๐๐๐ก ๐
๐๐๐๐ก ๐๐๐ก (๐ก)] (2)
๐๐ท๐ (๐ก) =1
2[๐๐ผ๐๐
โ๐๐๐ผ๐๐๐ผ๐ (๐ก) + ๐๐๐ ๐
โ๐๐๐๐ ๐๐๐ (๐ก)] (3)
๐๐๐ (๐ก) =1
2[๐๐ผ๐๐
๐๐๐ผ๐๐๐ผ๐ (๐ก)โ ๐๐๐ ๐
๐๐๐๐ ๐๐๐ (๐ก)]. (4)
In MIMO systems, different RF chains for different antennasmay give rise to different I/Q imbalances. The above responsescorresponding to the ๐th transmit antenna and the ๐th receiveantenna are denoted by ๐๐ท๐,๐(๐ก), ๐
๐๐,๐(๐ก), ๐
๐ท๐ ,๐(๐ก), and ๐๐๐ ,๐(๐ก),
respectively. Then, a simplified low-pass-equivalent signalmodel for MIMO can be obtained as in Fig. 2 for the ๐threceive antenna where the impulse responses of the overalldirect channel ๐๐๐(๐ก) and the overall mirror channel ๐๐๐(๐ก) forthe ๐th transmit antenna and the ๐th receive antenna read as
๐๐๐(๐ก) = ๐๐ท๐,๐(๐ก) โ โ๐๐(๐ก) โ ๐๐ท๐ ,๐(๐ก)
+ (๐๐๐,๐(๐ก))โ โ โโ๐๐(๐ก) โ ๐๐๐ ,๐(๐ก) (5)
๐๐๐(๐ก) = ๐๐๐,๐(๐ก) โ โ๐๐(๐ก) โ ๐๐ท๐ ,๐(๐ก)
+ (๐๐ท๐,๐(๐ก))โ โ โโ๐๐(๐ก) โ ๐๐๐ ,๐(๐ก). (6)
The receive filter output signal corresponding to the receiveantenna ๐ and transmit signals {๐ ๐(๐ก)} is
๐๐(๐ก) =
๐Txโ1โ๐=0
(๐ ๐(๐ก) โ ๐๐(๐ก) + ๐ โ๐ (๐ก) โ ๐๐(๐ก)) + ๐๐(๐ก) (7)
where the complex Gaussian noise ๐๐(๐ก) is given by
๐๐(๐ก) = ๐ค(๐ก) โ ๐๐ท๐ ,๐(๐ก) + ๐คโ(๐ก) โ ๐๐๐ ,๐(๐ก). (8)
When MIMO channels are independent or their joint statis-tics are unknown, the logical approach to the estimation ofthe equivalent direct and mirror channels is to estimate themat each receive antenna independently. Hence, we just needto describe them for one receive antenna. In the following,we drop the receive antenna index ๐. We consider an OFDMsystem with a cyclic prefix (CP) interval (๐CP samples) longerthan the maximum span (๐ฟ samples) of {๐๐(๐ก)} and {๐๐(๐ก)}.
Now, consider a low-pass-equivalent discrete-time OFDMsystem with ๐ subcarriers. The channels are assumed to be
constant over ๐พ symbols. The discrete-time transmit trainingsignal from the ๐th transmit antenna during the ๐th symbol isdenoted by ๐ (๐)๐ [๐] with the integer index ๐ โ [โ๐CP, ๐ โ 1],and the CP samples ๐
(๐)๐ [๐] = ๐
(๐)๐ [๐ โ ๐] for ๐ โ
[โ๐CP,โ1]. Similar notations apply to the data signal ๐ฅ(๐)๐ [๐].The discrete-time versions of ๐๐(๐ก) and ๐๐(๐ก) are denoted by๐ฟร1 vectors ๐๐ and ๐๐, respectively. The time-domain ๐ ร1received signal vector after the CP removal at each receiveantenna for the ๐th OFDM symbol, denoted by ๐๐, can beexpressed in a general pilot-data multiplexed setup (whichincludes pilot only or data only symbols as special cases) as
๐๐ =
๐Txโ1โ๐=0
{(๐บ๐[๐] +๐ฟ๐[๐])๐๐ + (๐บโ๐ [๐] +๐ฟโ
๐ [๐])๐๐}+ ๐๐
(9)
where (๐, ๐)th element of the ๐ ร ๐ฟ signal matrix ๐บ๐[๐] (or๐ฟ๐[๐]) is ๐ (๐)๐ [๐ โ ๐] (or ๐ฅ(๐)๐ [๐ โ ๐]) with ๐ โ [0, ๐ โ 1]and ๐ โ [0, ๐ฟ โ 1]. The received signal vector for ๐พ OFDMsymbols is given by
๐ = ๐บ๐+ ๐บโ๐ +๐ฟ๐+๐ฟโ๐ + ๐ (10)
where ๐ = [๐๐0 ๐๐1 . . . ๐
๐๐พโ1]
๐ , ๐ = [๐๐0 ๐
๐1 . . .๐
๐๐Txโ1
]๐ ,๐ = [๐๐0 ๐
๐1 . . .๐
๐๐Txโ1
]๐ , ๐ = [๐๐0 ๐
๐1 . . .๐
๐๐พโ1]
๐ , the(๐, ๐)th submatrice of ๐บ and ๐ฟ are respectively given by ๐บ๐[๐]and ๐ฟ๐[๐], with ๐ โ [0,๐พ โ 1] and ๐ โ [0, ๐Tx โ 1]. Thecomplex Gaussian noise vectors {๐๐} are given by
๐๐ = ๐ฎ๐ ,๐ท๐๐ +๐ฎ๐ ,๐๐โ๐ (11)
where {๐๐} are independent and identically distributed (i.i.d.)random vectors, each consisting of i.i.d. circularly-symmetriccomplex Gaussian random variables with the variance ๐2๐ค. Let๐ denote the maximum of the numbers of taps of ๐๐ท๐ [๐] and๐๐๐ [๐]. Then, ๐ฎ๐ ,๐ท and ๐ฎ๐ ,๐ are ๐ร(๐+๐) matrices withtheir first rows given by [๐๐ท๐ [0], ๐
๐ท๐ [1], . . . , ๐
๐ท๐ [๐], 01ร๐โ1]
and [๐๐๐ [0], ๐๐๐ [1], . . . , ๐
๐๐ [๐], 01ร๐โ1], respectively, and
their next ๐th rows are cyclically ๐-right-shift versions of theircorresponding first rows.
Let ๐๐,๐[๐] and ๐๐,๐[๐], respectively, denote the pilot sym-bol and the data symbol on the ๐th subcarrier of the ๐thOFDM symbol from the ๐th transmit antenna. Define ฮ๐,๐ โdiag{๐๐,๐[0], ๐๐,๐[1], . . . , ๐๐,๐[๐ โ 1]} and ๐ซ๐,๐ โ diag{๐๐,๐[0],๐๐,๐[1], . . . , ๐๐,๐[๐ โ 1]} . Let ๐ญ๐ฟ denote the first ๐ฟ columnsof the ๐ร๐ unitary discrete Fourier transform (DFT) matrix๐ญ . Then the time domain signal matrices can be given as
๐บ๐[๐] =โ๐๐ญ๐ปฮ๐,๐๐ญ ๐ฟ, ๐ฟ๐[๐] =
โ๐๐ญ๐ป๐ซ๐,๐๐ญ ๐ฟ. (12)
Define ๐ท โ ๐ญ๐ญ ๐ . Then, we have ๐ท = ๐ท ๐ , ๐ท๐ท ๐ = ๐ฐ๐ ,๐ท๐ญ โ
๐ฟ = ๐ญ ๐ฟ, and
ฮฬ๐,๐ โ ๐ทฮ๐ป๐,๐๐ท (13)
= diag{๐โ๐,๐[0], ๐โ๐,๐[๐ โ 1], ๐โ๐,๐[๐ โ 2], . . . , ๐โ๐,๐[2], ๐โ๐,๐[1]}๏ฟฝฬ๏ฟฝ๐,๐ โ ๐ท๐ซ๐ป
๐,๐๐ท (14)
= diag{๐โ๐,๐[0], ๐โ๐,๐[๐ โ 1], ๐โ๐,๐[๐ โ 2], . . . , ๐โ๐,๐[2], ๐โ๐,๐[1]}.
From the above and the DFT of (9), the received symbol
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2255
๐ ๐[๐] on the ๐th subcarrier of the ๐th OFDM symbol reads as
๐ ๐[๐] =
๐Txโ1โ๐=0
{(๐๐,๐[๐] + ๐๐,๐[๐])๐๐[๐]
+(๐โ๐,๐[โ๐] + ๐โ๐,๐[โ๐]
)๐๐[๐]
}(15)
where ๐๐[๐] and ๐๐[๐] are the ๐th elements ofโ๐๐ญ ๐ฟ๐๐ andโ
๐๐ญ๐ฟ๐๐, respectively. Clearly, the I/Q imbalance introducesto the received ๐th subcarrier symbol an interference from themirror tone (i.e., index โ๐) with a FD coefficient of ๐๐[๐].This interference, if not properly taken into account, can causea significant performance degradation.
III. MIMO OFDM PILOT DESIGN CRITERIA
For a coherent detection, the direct channel ๐ and the mirrorchannel ๐ need to be estimated at the receiver. In practicalsystems, the statistics of the channel and the transceiverimperfections are unknown and they can be non-stationary aswell. This leads to the practical choice of least-squares typechannel estimators as considered in this paper. The estimatesof the direct and mirror CIR vectors are given by
๏ฟฝฬ๏ฟฝ =(๐บ๐ป๐บ
)โ1
๐บ๐ป๐ (16)
๏ฟฝฬ๏ฟฝ =(๐บ๐๐บโ
)โ1
๐บ๐ ๐. (17)
Our pilot designs (and the optimality criterion) will be basedon minimizing the channel estimation mean-square error(MSE). Substituting (10) into (16) and (17) gives
๏ฟฝฬ๏ฟฝ =(๐บ๐ป๐บ
)โ1
๐บ๐ป๐บ๐+(๐บ๐ป๐บ
)โ1
๐บ๐ป๐บโ๐
+(๐บ๐ป๐บ
)โ1
๐บ๐ป๐ฟ๐+(๐บ๐ป๐บ
)โ1
๐บ๐ป๐ฟโ๐
+(๐บ๐ป๐บ
)โ1
๐บ๐ป๐ (18)
๏ฟฝฬ๏ฟฝ =(๐บ๐๐บโ
)โ1
๐บ๐๐บโ๐ +(๐บ๐๐บโ
)โ1
๐บ๐๐บ๐
+(๐บ๐๐บโ
)โ1
๐บ๐๐ฟ๐+(๐บ๐๐บโ
)โ1
๐บ๐๐ฟโ๐
+(๐บ๐๐บโ
)โ1
๐บ๐๐. (19)
All five terms in each of the above equations can affect theMSE. The second to the fourth terms are interferences andtheir contributions to the MSE is minimized if they becomezeros (no interferences). The first term needs to yield theparameter under estimation so that its contribution to the MSEis minimized (zero). The last term is due to the noise andwhen its covariance matrix is unknown (for the most generalcase), minimization of its contribution to MSE is impractical.However, in practice, the noise term is almost white as willbe explained below. Under this condition, the first and thelast terms are exactly the same as the system in [11] andhence, the MSE-minimizing condition (due to the first andthe last terms) is the same as that in [11]. All of the abovementioned (practically feasible) MSE-minimizing conditionscan be elaborated as follows:
1) Estimation Identifiability Condition: The identifiabilityof ๐ and ๐ estimation requires that ๐บ๐ป๐บ is of full-rank.
2) Zero Cross Channel Interference Condition: The MSEdue to the second term (cross channel interference) in(18) and (19) is removed when ๐บ๐ป๐บโ = 0๐ฟ๐Tx .
3) Zero Data Interference Condition: The random datainterference is completely suppressed when ๐บ๐ป๐ฟ =0๐ฟ๐Tx and ๐บ๐ป๐ฟโ = 0๐ฟ๐Tx .
4) White Noise Optimality Condition: When the equivalentreceive-filter is a square-root Nyquist filter, the MSEdue to the noise is minimized when ๐บ๐ป๐บ = ๐ธ๐พ๐ฐ๐ฟ๐Tx
where ๐ธ๐พ is the training signal energy from a trans-mit antenna over ๐พ symbols (excluding CPs). The FIreceiver I/Q imbalance with a square-root raised cosinereceive filter represents this scenario.
5) For the scenario with FD receiver I/Q imbalance, thenoise covariance matrix is unknown a priori, and hence itis infeasible to develop optimal pilot designs. However,the frequency selectivity of the receiver I/Q imbalanceis typically very small (and the noise would be almostwhite) since the amplifiers/filters are designed to havea frequency-flat response within the transmission band(see also [4], [21], [27], [28]). A practical approachin this case is to assume frequency-flat receiver I/Qimbalance1 in the pilot designs which leads to therequirement of ๐บ๐ป๐บ = ๐ธ๐พ๐ฐ๐ฟ๐Tx .
When the identifiability and zero data interference condi-tions are met, the MSEs of ๐ and ๐ estimation become
MSE๐ = Tr
[(๐บ๐ป๐บ
)โ1
๐บ๐ป๐บโ๐ธ[๐๐๐ป ]๐บ๐๐บ(๐บ๐ป๐บ
)โ1
+(๐บ๐ป๐บ
)โ1
๐บ๐ป๐ช๐๐บ(๐บ๐ป๐บ
)โ1]
(20)
MSE๐ = Tr
[(๐บ๐๐บโ
)โ1
๐บ๐๐บ๐ธ[๐๐๐ป ]๐บ๐ป๐บโ(๐บ๐๐บโ
)โ1
+(๐บ๐๐บโ
)โ1
๐บ๐๐ช๐๐บโ(๐บ๐๐บโ
)โ1]
(21)
where the noise covariance matrix ๐ช๐ is given by
๐ช๐ = ๐2๐ค(๐ฎ๐ ,๐ท๐ฎ๐ป๐ ,๐ท +๐ฎ๐ ,๐๐ฎ๐ป
๐ ,๐ )โ ๐ฐ๐พ . (22)
If the zero cross channel interference condition is also met,the above MSE expressions become
MSE๐ = MSE๐ = Tr[(๐บ๐ป๐บ)โ1๐บ๐ป๐ช๐๐บ(๐บ
๐ป๐บ)โ1]
(23)
Additionally, if the demodulator output noise samples arewhite (i.e., ๐ช๐ = ๐2๐๐ฐ), the MSE expressions simplify to
MSE๐ = MSE๐ = ๐2๐Tr[(๐บ๐ป๐บ)โ1
].
The MIMO OFDM pilot design criterion satisfying theestimation identifiability, zero data interference condition, zerocross channel interference, and white noise optimality readsas (
๐บ๐ป๐บ)= ๐ธ๐พ๐ฐ๐Tx๐ฟ &
(๐บ๐ป๐บโ
)= 0๐Tx๐ฟ
๐บ๐ป๐ฟ = 0๐Tx๐ฟ & ๐บ๐ป๐ฟโ = 0๐Tx๐ฟ
}(24)
1The transmitter I/Q imbalances need not be frequency-flat since they donot affect the receiver noise statistics.
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2256 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
The above criterion can be detailed as
Condition 1 :
๐พโ1โ๐=0
๐บ๐ป๐ [๐]๐ฟ๐[๐] = 0๐ฟ, โ๐, ๐ (25)
Condition 2 :
๐พโ1โ๐=0
๐บ๐ป๐ [๐]๐ฟ
โ๐[๐] = 0๐ฟ, โ๐, ๐ (26)
Condition 3 :
๐พโ1โ๐=0
๐บ๐ป๐ [๐]๐บ๐[๐] = ๐ธ๐พ๐ฐ๐ฟ, โ๐ (27)
Condition 4 :
๐พโ1โ๐=0
๐บ๐ป๐ [๐]๐บ๐[๐] = 0๐ฟ, โ๐ โ= ๐ (28)
Condition 5 :๐พโ1โ๐=0
๐บ๐ป๐ [๐]๐บ
โ๐[๐] = 0๐ฟ, โ๐, ๐. (29)
The corresponding MSE becomes
MSE๐ = MSE๐ =๐2๐ค๐ธ2๐พ
Tr[(
๐บ๐ป((๐ฎ๐ ,๐ท๐ฎ๐ป
๐ ,๐ท
+๐ฎ๐ ,๐๐ฎ๐ป๐ ,๐ )โ ๐ฐ๐พ
)๐บ)โ1
]. (30)
Following [26], we can also have the Cramer-Rao lowerbound (CRB) for the estimation of [๐๐ , ๐๐ ]๐ (i.e., a theoret-ical lower bound for MSE๐ + MSE๐) as
CRB = Tr
[{([๐บ, ๐บโ]๐ป ๐ช๐
โ1 [๐บ, ๐บโ])โ1
}], (31)
which will be used as a benchmark for the performanceevaluation of the proposed pilot designs.
IV. MIMO OFDM PILOT DESIGNS
We will first investigate what characteristics of pilots willsatisfy each of the above conditions, separately. Using (12),(13) and (14), we obtain
๐บ๐ป๐ [๐]๐ฟ๐[๐] = ๐๐ญ๐ป
๐ฟฮ๐ป๐,๐๐ซ๐,๐๐ญ ๐ฟ (32)
๐บ๐ป๐ [๐]๐ฟ
โ๐[๐] = ๐๐ญ๐ป
๐ฟฮ๐ป๐,๐๏ฟฝฬ๏ฟฝ
๐ป
๐,๐๐ญ ๐ฟ (33)
๐บ๐ป๐ [๐]๐บ๐[๐] = ๐๐ญ๐ป
๐ฟฮ๐ป๐,๐ฮ๐,๐๐ญ ๐ฟ (34)
๐บ๐ป๐ [๐]๐บ
โ๐[๐] = ๐๐ญ๐ป
๐ฟฮ๐ป๐,๐ฮฬ
๐ป
๐,๐๐ญ ๐ฟ. (35)
As data are random, using (32) and (33), we can concludethat Condition-1 and 2 require
ฮ๐ป๐,๐๐ซ๐,๐ = 0๐ & ฮ๐ป
๐,๐๏ฟฝฬ๏ฟฝ๐ป
๐,๐ = 0๐ โ๐,๐, ๐. (36)
In other words, at each OFDM symbol, the pilot tone indexset for all antennas and the data tone index set for all antennasneed to be disjoint and each set is composed of pairs of mirrortones only. The indexes of a mirror pair is given by (๐,โ๐) for๐ = 0, 1, . . . , ๐/2. Note that ๐ = 0 is a self-mirror tone andso is ๐ = ๐/2. Define ๐ฅ๐ โ โช๐Txโ1
๐=0 ๐ฅ๐,๐ and โ๐ โ โช๐Txโ1๐=0 โ๐,๐
where ๐ฅ๐,๐ and โ๐,๐ denote the pilot (including null pilot) indexset and the data index set of ๐th antenna at ๐th OFDM symbol,respectively. Note that ๐ฅ๐,๐ can be separated into non-zero pilottone index set ๐ฅ pilot
๐,๐ and null pilot tone index set ๐ฅ null๐,๐ . Then,
Condition-1 and 2 require that if ๐ โ ๐ฅ๐ and ๐ โ โ๐, then(โ๐)๐ โ ๐ฅ๐, (โ๐)๐ โ โ๐, and ๐ โ= ๐.
Next, by using (12), Condition-3 becomes
๐พโ1โ๐=0
๐ญ๐ป๐ฟ ฮ
๐ป๐,๐ฮ๐,๐๐ญ๐ฟ = ๐ธ๐พ๐ฐ๐ฟ, โ๐ (37)
or equivalently, for ๐ โ {โ๐ฟ+ 1,โ๐ฟ+ 2, . . . , ๐ฟโ 1},๐โ1โ๐=0
(๐พโ1โ๐=0
โฃ๐๐,๐[๐]โฃ2)๐๐2๐๐๐/๐ = ๐ธ๐พ๐ฟ[๐]. (38)
For a typical FFT size ๐ which is a power of 2, define ๐ฟ0 =2โlog2(L)โ, ๐ฟ๐ = 2
๐๐ฟ0, and ๐๐ = ๐/๐ฟ๐. Then, Condition-3 issatisfied when
๐พโ1โ
๐=0
โฃ๐๐,๐[๐]โฃ2 =
log2(๐0)โ1โ
๐=0
๐๐โ1โ
๐=0
๐ฟ๐โ1โ
๐=0
๐๐,๐๐ฟ[๐ โ ๐๐๐ โ ๐๐,๐],
(39)
where {๐๐,๐} take on real non-negative values and satisfy
log2(๐0)โ1โ๐=0
๐๐โ1โ๐=0
๐๐,๐ = ๐ธ๐พ , ๐๐,๐ โฅ 0, ๐๐,๐ โ [0,๐๐ โ 1].(40)
Similarly, Condition-4 becomes
๐พโ1โ๐=0
๐ญ๐ป๐ฟฮ๐ป
๐,๐ฮ๐,๐๐ญ๐ฟ = 0๐ฟ, โ๐ โ= ๐ (41)
or equivalently, for ๐ โ {โ๐ฟ+ 1,โ๐ฟ+ 2, . . . , ๐ฟโ 1},๐โ1โ๐=0
๐พโ1โ๐=0
๐โ๐,๐[๐]๐๐,๐[๐]๐๐2๐๐๐/๐ = 0, โ๐ โ= ๐. (42)
With the definitions of ๐บ๐,๐[๐, ๐] โ ๐โ๐,๐[๐]๐๐,๐[๐] and{๐๐,๐[๐, ๐]} being the inverse DFT of {๐บ๐,๐[๐, ๐]}, a sufficientcondition for satisfying Condition-4 is either of the twofollowing conditions:
๐พโ1โ๐=0
๐บ๐,๐[๐, ๐] = 0, โ๐ โ= ๐, โ๐ (43)
๐๐,๐[๐, ๐] = 0,
๐ = 0, 1, . . . , ๐ฟโ 1, ๐ โ ๐ฟ+ 1, . . . , ๐ โ 1, โ๐ โ= ๐. (44)
Condition-5 can be expressed as
๐พโ1โ๐=0
๐ญ๐ป๐ฟฮ๐ป
๐,๐๐ญ๐ญ ๐ฮ๐ป๐,๐๐ญ
โ๐ฟ = 0๐ฟ, โ๐,๐. (45)
Then, Condition-5 becomes, for all ๐ and ๐,
๐พโ1โ๐=0
๐ญ๐ป๐ฟฮ๐ป
๐,๐ฮฬ๐ป
๐,๐๐ญ ๐ฟ = ๐ญ๐ป๐ฟ
(๐พโ1โ๐=0
ฮ๐ป๐,๐ฮฬ
๐ป
๐,๐
)๐ญ ๐ฟ = 0๐ฟ,
(46)
or equivalently, for ๐ โ {โ๐ฟ+ 1,โ๐ฟ+ 2, . . . , ๐ฟโ 1},๐โ1โ๐=0
(๐พโ1โ๐=0
๐โ๐,๐[๐]๐โ๐,๐[๐ โ ๐]
)๐๐2๐๐๐/๐ = 0, โ๐,๐. (47)
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2257
This mandates that either the vector{(โ๐พโ1๐=0 ๐โ๐,๐[๐]๐
โ๐,๐[๐ โ ๐]
): ๐ = 0, . . . , ๐ โ 1
}be in
the null space of the columns of ๐ญ๐ฟ and ๐ญ๐ป๐ฟ , โ๐,๐, ๐, as
๐พโ1โ๐=0
๐โ๐,๐[๐]๐โ๐,๐[๐ โ ๐] = 0, (48)
๐พโ1โ๐=0
๐โ๐,๐[๐]๐โ๐,๐[๐ โ ๐] = ๐๐2๐๐๐/๐ , ๐ โ {๐ฟ, . . . , ๐ โ ๐ฟ},
(49)
or, for ๐ = โ(๐ฟโ 1), . . . , (๐ฟโ 1),๐โ1โ๐=0
๐๐,๐[๐]๐๐,๐[๐ โ ๐]๐๐2๐๐๐/๐ = 0, โ๐,๐, ๐. (50)
By considering ๐ = ๐ in (50), one can easily find that 2๐ฟ0
pilot tones (may include null pilots) are necessary for onetransmit antenna.
Combining all of the above pilot characteristics, we presentseveral pilot designs which satisfy the five design criteria.These designs will be denoted by two terms โ how pilotsof different antennas are decoupled and how mirror toneinterferences are suppressed. The same pilot signal energyover ๐พ symbols for each antenna as required in Condition-3 will not be explicitly mentioned in the following designs.Without loss of generality, we will present unit-amplitudepilot symbols. The existing MIMO OFDM pilot designsfrom [11], namely time division multiplexing (TDM), codedivision multiplexing across time domain (CDM-T), CDMacross frequency domain (CDM-F), frequency division multi-plexing (FDM), and time and frequency division multiplexing(TFDM), will be extensively used in our designs. Due to spacelimitation, details of these existing designs are referred to[11]. How our pilot designs satisfy the five design criteriawill be briefly mentioned for the first design, but it shouldbe obvious and hence will be skipped for the other designs.Examples of our pilot designs are provided in Tables I and IIfor illustration of the concepts where parameters are chosen forthe convenience of the presentation. In some designs, severaloptions are presented by separating them with dashed lines.
A. [TDM; TD / C-F] Design
In this design, pilots of different antennas are decoupled byTDM design while mirror tone interferences are suppressed bymeans of time disjointness (TD) or code design in frequencydomain (C-F). The pilot and data index sets for the ๐th transmitantenna at the ๐th OFDM symbol are given by
๐ฅ๐,๐ =
{ {0,๐0/2,๐0, . . . , ๐ โ๐0/2}, if ๐ = ๐โ , else
(51)
โ๐ = {0, 1, 2, . . . , ๐ โ 1} โ ๐ฅ๐ (52)
where TDM pilot assignment for different antennas can beeasily noticed. Due to TDM, there is no mirror tone interfer-ence across antennas (i.e., mirror tone interference suppression
through TD). For each antenna, the pilots are given by
๐๐,๐[0] = ยฑ๐๐,๐[๐/2] = ยฑ1 or ยฑ ๐ (53)
๐๐,๐[๐] = ๐๐๐๐,๐ , arbitrary ๐๐,๐,
๐ โ {๐0/2,๐0, . . . , (๐ โ๐0)/2} (54)
๐๐,๐[๐] = (โ1)(2๐โ๐)/๐0๐2๐,๐[0]๐โ๐,๐[โ๐],
๐ โ {(๐ +๐0)/2, ๐/2 +๐0, . . . , ๐ โ๐0/2} (55)
๐๐,๐[๐] = 0, ๐ /โ ๐ฅ๐,๐. (56)
For example, {๐๐,๐[๐] : ๐ โ ๐ฅ๐,๐} can be either {ยฑ1, ๐1, ๐2,. . . , ๐๐ฟ0โ1, ยฑ1,โ๐โ๐ฟ0โ1, ๐โ๐ฟ0โ2, โ๐โ๐ฟ0โ3, . . . , ๐โ2,โ๐โ1}or {ยฑ๐, ๐1, ๐2, . . . , ๐๐ฟ0โ1, ยฑ๐, ๐โ๐ฟ0โ1, โ๐โ๐ฟ0โ2, ๐
โ๐ฟ0โ3,
. . . ,โ๐โ2, ๐โ1} where {๐๐} are arbitrary unit amplitude sym-bols. The above pilot codes across the frequency domain aredesigned such that the mirror interference becomes zero, i.e.,to satisfy (50) for ๐ = ๐. Note that (51) and (52) guaranteeConditions 1 and 2, while TDM fulfills Conditions 3 and 4.TDM and C-F guarantee Condition 5. This design requires2๐ฟ0 pilot tones in each symbol and a total of ๐พ = ๐Tx
symbols. ๐ฟ๐ and ๐๐ with ๐ > 0 can be used instead of ๐ฟ0
and ๐0, but they cost more overhead.
B. [CDM-T; C-T] Design Without Self-Mirror Tones
Pilots of different antennas are decoupled by CDM-T de-sign. Pilot mirror tone interferences are suppressed by codedesign across time domain (C-T). The pilot and data indexsets are given by
๐ฅ๐,๐ = {๐0
2,3๐0
2,5๐0
2, . . . , ๐ โ ๐0
2}, โ๐, ๐ (57)
โ๐ = {0, 1, 2, . . . , ๐ โ 1} โ ๐ฅ๐ (58)
where self-mirror tones (i.e., index 0 and ๐/2) are excludedfrom the pilot index set. Define
๐๐[๐] โ [๐๐,0[๐], ๐๐,1[๐], . . . , ๐๐,๐พโ1[๐]]๐ (59)
[๐๐]๐ โ ๐๐2๐๐๐/๐พ ,๐, ๐ โ {0, 1, . . . ,๐พ โ 1},๐พ โฅ 2(๐Tx + ๐), ๐ โ {0, 1, 2, . . .}. (60)
For ๐ โ {๐0/2, 3๐0/2, . . . , ๐/2 โ ๐0/2}, the pilots aregiven as
๐0[๐] = ๐๐๐0,๐ , ๐ โ ๐ฅ pilot0 (61)
๐0[โ๐] = ๐๐๐0,โ๐diag{๐1}๐โ0[๐] (62)
๐๐[๐] = ๐๐๐๐,๐diag{๐2๐}๐0[๐], ๐ = 1, . . . , ๐Tx โ 1 (63)
๐๐[โ๐] = ๐๐๐๐,โ๐diag{๐2๐+1}๐โ0[๐], ๐ = 1, . . . , ๐Tx โ 1(64)
where {๐๐,ยฑ๐} are arbitrary. In the above design, for ๐ โ{๐0/2, 3๐0/2, . . . , ๐/2 โ๐0/2}, {๐๐[๐]} for different an-tenna ๐ โ {0, 1, . . . , ๐Txโ1} are constructed using diag{๐2๐}while {๐๐[โ๐]} are based on diag{๐2๐+1}. Note that anypermutation of {๐ฃ๐ : ๐ = 1, . . . , 2๐Tx โ 1} can be associatedto {๐๐[๐]} in the above equations. Overhead consideration willset ๐พ = 2๐Tx. Although ๐๐ can be used in place of ๐0, itis not desirable due to larger overhead.
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2258 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
TABLE IEXAMPLES OF PROPOSED PILOT DESIGNS
Design antenna symbol Pilotsindex ๐ index ๐
[TDM; TD / C-F] ๐ฅ๐,๐ = {0, 2, 4, 6, 8, 10, 12, 14}๐ = 16, ๐ฟ0 = 4, ๐0 = 4, 0 0 ๐0,0[๐ โ ๐ฅ๐,๐] = {๐, ๐1, ๐2, ๐3, ๐, ๐
โ3 ,โ๐โ
2, ๐โ1}
๐พ = 2, ๐Tx = 2 1 1 ๐1,1[๐ โ ๐ฅ๐,๐] = {๐, ๐4, ๐5, ๐6,โ๐, ๐โ6 ,โ๐โ
5, ๐โ4}
[TDM; TD / C-F] ๐ฅ๐,๐ = {0, 2, 4, 6, 8, 10, 12, 14}๐ = 16, ๐ฟ0 = 4, ๐0 = 4, 0 0 ๐0,0[๐ โ ๐ฅ๐,๐] = {1, ๐1, ๐2, ๐3, 1,โ๐โ
3, ๐โ2 ,โ๐โ
1}๐พ = 2, ๐Tx = 2 1 1 ๐1,1[๐ โ ๐ฅ๐,๐] = {โ1, ๐4, ๐5, ๐6, 1,โ๐โ
6, ๐โ5 ,โ๐โ
4}[CDM-T; C-T] ๐ฅ๐,๐ = {2, 6, 10, 14}
without self-mirror 0 0 ๐0,0[๐ โ ๐ฅ๐,๐] = {๐1, ๐5, ๐โ5, ๐
โ1}
๐ = 16, ๐ฟ0 = 4, ๐0 = 4, 0 1 ๐1,0[๐ โ ๐ฅ๐,๐] = {๐2, ๐6, ๐โ6๐
๐๐/2, ๐โ2๐
๐๐/2}๐พ = 4, ๐Tx = 2 0 2 ๐2,0[๐ โ ๐ฅ๐,๐] = {๐3, ๐7,โ๐โ
7,โ๐โ3}
0 3 ๐3,0[๐ โ ๐ฅ๐,๐] = {๐4, ๐8, ๐โ8๐
โ๐๐/2, ๐โ4๐
โ๐๐/2}1 0 ๐0,1[๐ โ ๐ฅ๐,๐] = {๐1, ๐5, ๐
โ5, ๐
โ1}
1 1 ๐1,1[๐ โ ๐ฅ๐,๐] = {โ๐2,โ๐6, ๐โ6๐
๐๐/2, ๐โ2๐
๐๐/2}1 2 ๐2,1[๐ โ ๐ฅ๐,๐] = {๐3, ๐7,โ๐โ
7,โ๐โ3}
1 3 ๐3,1[๐ โ ๐ฅ๐,๐] = {โ๐4,โ๐8, ๐โ8๐
โ๐๐/2, ๐โ4๐
โ๐๐/2}[CDM-T; C-T] ๐ฅ๐,๐ = {0, 4, 8, 12}with self-mirror 0 0 ๐0,0[๐ โ ๐ฅ๐,๐] = {1, ๐1, 1, ๐
โ1}
๐ = 16, ๐ฟ0 = 4, ๐0 = 4, 0 1 ๐1,0[๐ โ ๐ฅ๐,๐] = {๐๐๐/4, ๐2, ๐๐๐/4, ๐โ
2๐๐๐/2}
๐พ = 4, ๐Tx = 2 0 2 ๐2,0[๐ โ ๐ฅ๐,๐] = {๐๐๐/2, ๐3, ๐๐๐/2,โ๐โ
3}0 3 ๐3,0[๐ โ ๐ฅ๐,๐] = {๐โ๐๐/4, ๐4, ๐
โ๐๐/4, ๐โ4๐
โ๐๐/2}1 0 ๐0,1[๐ โ ๐ฅ๐,๐] = {1, ๐1, 1, ๐
โ1}
1 1 ๐1,1[๐ โ ๐ฅ๐,๐] = {๐๐๐/4, ๐2, ๐๐๐/4, ๐โ
2๐๐๐/2}
1 2 ๐2,1[๐ โ ๐ฅ๐,๐] = {๐๐๐/2, ๐3, ๐๐๐/2,โ๐โ
3}1 3 ๐3,1[๐ โ ๐ฅ๐,๐] = {๐โ๐๐/4, ๐4, ๐
โ๐๐/4, ๐โ4๐
โ๐๐/2}[CDM-F; Null] ๐ฅ pilot
๐ = {1, 2, 9, 10}๐ = 16, ๐ฟ0 = 2, ๐0 = 8, ๐ฅ null
๐ = {6, 7, 14, 15}๐พ = 1, ๐Tx = 2 0 0 ๐0[๐ โ ๐ฅ pilot
๐ ] = {๐1, ๐2, ๐3, ๐4}1 0 ๐1[๐ โ ๐ฅ pilot
๐ ] = {๐1,โ๐2, ๐3,โ๐4}[CDM-F; Null] ๐ฅ pilot
๐ = {1, 5, 9, 13, 17, 21, 25, 29, }with equi-spaced pilots ๐ฅ null
๐ = {3, 7, 11, 15, 19, 23, 27, 31}๐ = 32, ๐ฟ0 = 2, ๐0 = 16, 0 0 ๐0[๐ โ ๐ฅ pilot
๐ ] = {๐1, ๐2, ๐3, ๐4, ๐5, ๐6, ๐7, ๐8}๐พ = 1, ๐Tx = 4 1 0 ๐1[๐ โ ๐ฅ pilot
๐ ] = {๐1๐๐๐/16, ๐2๐
๐5๐/16, ๐3๐๐9๐/16, ๐4๐
๐13๐/16 ,
๐4๐๐โ15๐/16 , ๐3๐
๐โ11๐/16, ๐2๐๐โ7๐/16, ๐1๐
๐โ3๐/16}2 0 ๐2[๐ โ ๐ฅ pilot
๐ ] = {๐1๐๐๐/8, ๐2๐
๐5๐/8, ๐3๐๐โ7๐/8, ๐4๐
๐โ3๐/8,
๐4๐๐๐/8, ๐3๐
๐5๐/8, ๐2๐๐โ7๐/8, ๐1๐
๐โ3๐/8}3 0 ๐3[๐ โ ๐ฅ pilot
๐ ] = {๐1๐๐3๐/8, ๐2๐
๐โ๐/8, ๐3๐๐โ3๐/8, ๐4๐
๐7๐/8,
๐4๐๐3๐/8, ๐3๐
๐โ๐/8, ๐2๐๐โ3๐/8, ๐1๐
๐7๐/8}[FDM; Null] ๐ฅ pilot
๐ = {1, 3, 9, 11}๐ = 16, ๐ฟ0 = 2, ๐0 = 8, ๐ฅ null
๐ = {5, 7, 13, 15}๐พ = 1, ๐Tx = 2 0 0 ๐0[๐ โ ๐ฅ pilot
๐ ] = {๐1, 0, ๐2, 0}1 0 ๐1[๐ โ ๐ฅ pilot
๐ ] = {0, ๐3, 0, ๐4}[TFDM; Null / C-F] ๐ฅ null
๐,๐ =โช๐ฅ๐,๐ where ๐ โ= ๐ and ๐ โ= ๐
๐ = 64, ๐ฟ0 = 2, ๐1 = 16, 0 0 ๐ฅ0,0 = ๐ฏ2,๐0,๐ = {1, 33, } ๐0,0[๐ โ ๐ฅ0,0] = {๐1, ๐2}๐พ = 2, ๐Tx = 3 0 1 ๐ฅ0,1 = ๐ฏ2,๐0,๐ = {17, 49} ๐0,1[๐ โ ๐ฅ0,1] = {๐3, ๐4}
1 0 ๐ฅ1,0 = ๐ฏ2,๐1,๐ = {15, 47, } ๐1,0[๐ โ ๐ฅ1,0] = {๐5, ๐6}1 1 ๐ฅ1,1 = ๐ฏ2,๐1,๐ = {31, 63} ๐1,1[๐ โ ๐ฅ1,1] = {๐7, ๐8}2 0 ๐ฅ2,0 = ๐ฏ2,๐1 = {0, 16, 32, 48} ๐2,0[๐ โ ๐ฅ2,0 ] = {ยฑโฃ๐1โฃ, ๐2,ยฑโฃ๐1โฃ,โ๐โ2}2 0 ๐ฅ2,0 = ๐ฏ2,๐1,๐ = {0, 32} ๐2,0[๐ โ ๐ฅ2,0] = {1, 1}
(alternative) 1 ๐ฅ2,1 = ๐ฏ2,๐1,๐ = {16, 48} ๐2,1[๐ โ ๐ฅ2,1] = {๐1, ๐2}2 0 ๐ฅ2,0 = ๐ฏ2,๐1/2,๐ = {8, 40} ๐2,0[๐ โ ๐ฅ2,0] = {๐9, ๐10}
(alternative) 1 ๐ฅ2,1 = ๐ฏ2,๐1/2,๐ = {24, 56} ๐2,1[๐ โ ๐ฅ2,1] = {๐11, ๐12}[TFDM / CDM-T; Null / C-T] ๐ฅ null
๐,๐ =โช๐ฅ๐,๐ where ๐ โ= ๐ and ๐ โ= ๐
๐ = 64, ๐ฟ0 = 4, ๐0 = 16, 0 0 ๐ฅ0,0 = ๐ฏ0,๐0,0 = {1, 17} ๐0,0[๐ โ ๐ฅ0,0] = {๐1, ๐2}๐พ = 4, ๐Tx = 5 0 1 ๐ฅ0,1 = ๐ฏ0,๐0,0 = {1, 17} ๐0,1[๐ โ ๐ฅ0,1] = {๐3, ๐4}
0 2 ๐ฅ0,2 = ๐ฏ0,๐0,1 = {33, 49} ๐0,2[๐ โ ๐ฅ0,2] = {๐5, ๐6}0 3 ๐ฅ0,3 = ๐ฏ0,๐0,1 = {33, 49} ๐0,3[๐ โ ๐ฅ0,2] = {๐7, ๐8}1 0 ๐ฅ1,0 = ๐ฏ0,๐0,0 = {1, 17} ๐1,0[๐ โ ๐ฅ1,0 ] = {๐1๐
๐๐1 ,โ๐2๐๐๐2}
1 1 ๐ฅ1,1 = ๐ฏ0,๐0,0 = {1, 17} ๐1,1[๐ โ ๐ฅ1,1 ] = {โ๐3๐๐๐1 , ๐4๐
๐๐2}1 2 ๐ฅ1,2 = ๐ฏ0,๐0,1 = {33, 49} ๐1,2[๐ โ ๐ฅ1,2] = {๐5๐
๐๐3 , ๐6๐๐๐4}
1 3 ๐ฅ1,3 = ๐ฏ0,๐0,1 = {33, 49} ๐1,3[๐ โ ๐ฅ1,2] = {โ๐7๐๐๐3 ,โ๐8๐
๐๐4}2 0 ๐ฅ2,0 = ๐ฏ0,๐15,0 = {15, 31} ๐2,0[๐ โ ๐ฅ2,0] = {๐1, ๐2}2 1 ๐ฅ2,1 = ๐ฏ0,๐15,0 = {15, 31} ๐2,1[๐ โ ๐ฅ2,1] = {๐3, ๐4}2 2 ๐ฅ2,2 = ๐ฏ0,๐15,1 = {47, 63} ๐2,2[๐ โ ๐ฅ2,2] = {๐5, ๐6}2 3 ๐ฅ2,3 = ๐ฏ0,๐15,1 = {47, 63} ๐2,3[๐ โ ๐ฅ2,2] = {๐7, ๐8}3 0 ๐ฅ3,0 = ๐ฏ0,๐15,0 = {15, 31} ๐3,0[๐ โ ๐ฅ3,0] = {๐1๐๐๐5 , ๐2๐
๐๐6}3 1 ๐ฅ3,1 = ๐ฏ0,๐15,0 = {15, 31} ๐3,1[๐ โ ๐ฅ3,1] = {โ๐3๐
๐๐5 ,โ๐4๐๐๐6}
3 2 ๐ฅ3,2 = ๐ฏ0,๐15,1 = {47, 63} ๐3,2[๐ โ ๐ฅ3,2] = {๐5๐๐๐7 , ๐6๐๐๐8}
3 3 ๐ฅ3,3 = ๐ฏ0,๐15,1 = {47, 63} ๐3,3[๐ โ ๐ฅ3,2] = {๐7๐๐๐7 , ๐8๐๐๐8}
4 0 ๐ฅ4,0 = ๐ฏ0,๐2,0 = {4, 20} ๐4,0[๐ โ ๐ฅ4,0] = {๐1, ๐2}4 1 ๐ฅ4,1 = ๐ฏ0,๐2,1 = {44, 60} ๐4,1[๐ โ ๐ฅ4,1] = {๐3, ๐4}4 ๐ฅ4,0 = ๐ฏ0,๐0/2 = {0, 8, 16, 24, 32, 40, 48, 56}
(alternative) 0 ๐4,0[๐ โ ๐ฅ4,0] = {ยฑโฃ๐1โฃ, ๐1, ๐2,ยฑโฃ๐0โฃ,โ๐โ2, ๐โ1,โ๐โ0}
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2259
TABLE IIEXAMPLES OF PROPOSED PILOT DESIGNS CONTโD.
Design antenna symbol Pilotsindex ๐ index ๐
[CDM-F; C-T] ๐ฅ๐ = {4, 12, 20, 28, 36, 44, 52, 60}๐ = 64, ๐ฟ0 = 4, ๐0 = 16 0 0 ๐0,0[๐ โ ๐ฅ๐] = {๐1, ๐2, ๐3, ๐4, ๐5, ๐6, ๐7, ๐8}
๐พ = 2, ๐Tx = 2 0 1 ๐0,1[๐ โ ๐ฅ๐] = {๐๐1, ๐๐2, ๐๐3, ๐๐4, ๐๐5, ๐๐6, ๐๐7, ๐๐8}1 0 ๐1,0[๐ โ ๐ฅ๐] = {๐1,โ๐2, ๐3,โ๐4, ๐5,โ๐6, ๐7,โ๐8}1 1 ๐1,1[๐ โ ๐ฅ๐] = {๐๐1,โ๐๐2, ๐๐3,โ๐๐4, ๐๐5,โ๐๐6, ๐๐7,โ๐๐8}
[FDM; C-T] ๐ฅ๐ = {4, 12, 20, 28, 36, 44, 52, 60}๐ = 64, ๐ฟ0 = 4, ๐0 = 16 0 0 ๐0,0[๐ โ ๐ฅ๐] = {๐1, 0, ๐2, 0, ๐3, 0, ๐4, 0}
๐พ = 2, ๐Tx = 2 0 1 ๐0,1[๐ โ ๐ฅ๐] = {๐๐1, 0, ๐๐2, 0, ๐๐3, 0, ๐๐4, 0}1 0 ๐1,0[๐ โ ๐ฅ๐] = {0, ๐5, 0, ๐6, 0, ๐7, 0, ๐8}1 1 ๐1,1[๐ โ ๐ฅ๐] = {0, ๐๐5, 0, ๐๐6, 0, ๐๐7, 0, ๐๐8}
[FDM; C-T] ๐ฅ๐ = {4, 12, 20, 28, 36, 44, 52, 60}(alternative) 0 0 ๐0,0[๐ โ ๐ฅ๐] = {๐1, 0, ๐2, 0, ๐3, 0, ๐4, 0}
๐ = 64, ๐ฟ0 = 4, ๐0 = 16 0 1 ๐0,1[๐ โ ๐ฅ๐] = {๐1, 0, ๐2, 0,โ๐3, 0,โ๐4, 0}๐พ = 2, ๐Tx = 2 1 0 ๐1,0[๐ โ ๐ฅ๐] = {0, ๐5, 0, ๐6, 0, ๐7, 0, ๐8}
1 1 ๐1,1[๐ โ ๐ฅ๐] = {0, ๐5, 0, ๐6, 0,โ๐7, 0,โ๐8}[FDM; C-T] ๐ฅ๐ = {4, 12, 20, 28, 36, 44, 52, 60}(alternative) 0 0 ๐0,0[๐ โ ๐ฅ๐] = {๐1, 0, ๐2, 0, ๐3, 0, ๐4, 0}
๐ = 64, ๐ฟ0 = 4, ๐0 = 16 0 1 ๐0,1[๐ โ ๐ฅ๐] = {๐1, 0, ๐2, 0, ๐3, 0, ๐4, 0}๐พ = 2, ๐Tx = 2 1 0 ๐1,0[๐ โ ๐ฅ๐] = {0, ๐5, 0, ๐6, 0, ๐7, 0, ๐8}
1 1 ๐1,1[๐ โ ๐ฅ๐] = {0,โ๐5, 0,โ๐6, 0,โ๐7, 0,โ๐8}[TDM; Null] ๐ฅ pilot
๐ = {1, 2, 9, 10}๐ = 16, ๐ฟ0 = 2, ๐0 = 8, ๐ฅ null
๐ = {6, 7, 14, 15}๐พ = 2, ๐Tx = 2 0 0 ๐0,0[๐ โ ๐ฅ pilot
๐ ] = {๐1, ๐2, ๐3, ๐4}1 1 ๐1,1[๐ โ ๐ฅ pilot
๐ ] = {๐1, ๐2, ๐3, ๐4}
C. [CDM-T; C-T] Design With Self-Mirror Tones
The C-T design in the above subsection uses ๐๐ and ๐๐ with๐ โ= ๐ for tones ๐ and โ๐. This approach becomes irrelevantfor self-mirror tones (i.e., for ๐ = (โ๐)๐ ). When the pilotscontain self-mirror tones, the index sets are given by
๐ฅ๐,๐ = {0,๐0, 2๐0, . . . , ๐ โ๐0}, โ๐, ๐ (65)
โ๐ = {0, 1, 2, . . . , ๐ โ 1} โ ๐ฅ๐. (66)
For non-self-mirror tones, the C-T design from the previoussubsection is applied with ๐พ defined below. At self-mirrortones ๐ = 0 and ๐/2, we present two designs. The first designis given by
๐0[๐] = ๐๐๐0,๐๐โฒ๐, ๐ โ {1, . . . ,๐พ โ 1} (67)
๐๐[๐] = ๐๐๐๐,๐diag{๐๐๐}๐โ0[๐], ๐ = 1, . . . , ๐Tx โ 1 (68)
where {๐๐,๐} are arbitrary, [๐โฒ๐]๐ โ
โ[๐๐]๐ and the indexes
๐ and ๐๐ satisfy i) (๐๐ + ๐๐ โ ๐)๐พ โ= 0, ๐๐,๐๐ โ{0, 1, . . . , ๐Txโ1},๐๐ โ= ๐๐ if ๐ โ= ๐, ii) (๐โ๐๐)๐พ โ= 0, iii)(๐๐โ๐๐)๐พ โ= 0, โ๐๐ โ= ๐๐ . For simplicity, we can set ๐ = 1in the above equations which yields 2 โค ๐, ๐ โค ๐พ/2 and๐พ = 2๐Tx. ๐พ can also be 2(๐Tx+๐) with ๐ โ {0, 1, 2, . . .}.
The second design at self-mirror tones is defined by
๐0[๐] = ๐๐๐0,๐๐๐, ๐ โ {1, . . . ,๐พ โ 1} & ๐ โ= ๐พ/2 (69)
๐๐[๐] = ๐๐๐๐,๐diag{๐๐๐}๐โ0[๐], ๐ = 1, . . . , ๐Tx โ 1 (70)
with the indexes ๐ and ๐๐ satisfying i) ๐๐ โ= ๐, (๐๐ โ๐)๐พ/2 โ= 0, ii) (๐๐ + ๐๐ โ 2๐)๐พ โ= 0, ๐๐,๐๐ โ{0, 1, . . . , ๐Tx โ 1},๐๐ โ= ๐๐ if ๐ โ= ๐, iii) (2๐โ๐๐)๐พ โ= 0,and {๐๐,๐} are arbitrary. By simply setting ๐ = 1, we have3 โค ๐ โค ๐พ/2 and ๐พ = 2(๐Tx + ๐) with ๐ โ {1, 2, . . .}.Using ๐๐ in place of ๐0 will require more overhead.
D. [CDM-F; Null] Design
This design uses ๐ ๐ฟ๐ constant amplitude pilot tones withthe index set ๐ฅ pilot
0,๐ = ๐ฅ pilot0 and ๐ ๐ฟ๐ null tones with the
index set ๐ฅ null0,๐ = ๐ฅ null
0 where ๐ฅ pilot0 and ๐ฅ null
0 are mirrorsof each other, and ๐ โฅ ๐Tx and ๐ ๐ฟ๐ โค ๐/2. Self-mirrortones cannot be used. Define ๐ฏ๐,๐ โ [๐, ๐ +๐๐, ๐ + 2๐๐,. . . , ๐+๐โ๐๐] which consists of cyclically equi-spaced ๐ฟ๐
indexes from [0, ๐ โ 1]. Then the index sets are given by
๐ฅ pilot0,๐ = ๐ฅ pilot
0 =
๐โ1โช๐=0
๐ฏ๐,๐๐ (71)
๐ฅ null0,๐ = ๐ฅ null
0 = {๐ โ ๐ฅ pilot0 } (72)
โ๐ = {0, 1, . . . , ๐ โ 1} โ {๐ฅ pilot0 โช ๐ฅ null
0 } (73)
where ๐๐ โ {{1, 2, . . . ,๐๐ โ 1} โ {๐๐/2}}, ๐๐ โ= ๐๐ if ๐ โ=๐, {๐๐} โฉ {๐๐ โ ๐๐} = โ , and ๐๐ โฅ 2๐Tx + 2.
Due to the mirror null tones, the I/Q imbalance inducedinterferences are completely suppressed. The pilots of differentantennas are decoupled by CDM-F design as
๐0[๐] = ๐๐๐๐ , arbitrary ๐๐, ๐ โ ๐ฅ pilot0 (74)
๐๐[๐] = ๐๐2๐๐๐
๐พ ๐0[๐], ๐ โ ๐ฏ๐,๐๐ , ๐ โ {1, . . . , ๐Tx โ 1} (75)
๐๐[๐] = 0, ๐ /โ ๐ฅ pilot0 , ๐ โ {0, . . . , ๐Tx โ 1}. (76)
If the elements of ๐ฅ pilot0 are cyclically equi-spaced, the pilots
can also be given by
๐0[๐] = ๐๐๐0,๐ , ๐ โ ๐ฅ pilot0 (77)
๐๐[๐] = ๐๐๐๐๐๐2๐๐๐๐/๐๐0[๐], ๐ โ {1, 2, . . . , ๐Tx โ 1},๐ฟ โค ๐1 โค ๐ฟ๐, ๐ฟ โค ๐๐+1 โ ๐๐ โค ๐ฟ๐ (78)
๐๐[๐] = 0, ๐ /โ ๐ฅ pilot0 , ๐ โ {0, 1, . . . , ๐Tx โ 1} (79)
where {๐๐, ๐0,๐} are arbitrary phases. The choice of ๐ = ๐Tx
and ๐ฟ๐ = ๐ฟ0 requires minimum overhead. Note that for ๐พ =1, the maximum number of transmit antennas this design cansupport is ๐
2๐ฟ0โ 1 since the null design cannot be applied to
๐ฏ1,0 which contains mirror pairs. However, the antenna ๐2๐ฟ0
can transmit pilots on ๐ฏ1,0 using C-F design.
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2260 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
E. [FDM; Null] Design
This design decouples pilots of different antennas throughFDM and suppresses mirror tone interferences by means ofmirror null tones. It uses ๐พ = 1 symbol with ๐Tx๐ฟ๐ โค ๐/2,and consists of ๐Tx๐ฟ๐ constant amplitude pilot tones withthe index set ๐ฅ pilot
0 = โช๐๐ฅ pilot0,๐ and ๐Tx๐ฟ๐ null tones with
the index set ๐ฅ null0 . The differences from [CDM-F; Null] are
(i) the constant amplitude pilots can be arbitrary within andacross antennas in this design while they are dependent acrossantennas in [CDM-F; Null] and (ii) a different antenna trans-mits its constant amplitude pilots only on distinct cyclicallyequi-spaced ๐ฟ๐ tones in this design while each antenna usesthe same ๐Tx๐ฟ๐ tones in [CDM-F; Null]. The index sets of[FDM; Null] design are defined by
๐ฅ pilot0,๐ = ๐ฏ๐,๐๐ , ๐๐ โ= ๐๐ if ๐ โ= ๐, (80)
๐ฅ null0,๐ = ๐ฅ null
0 = {๐ โ ๐ฅ pilot0 } (81)
โ๐ = {0, 1, . . . , ๐ โ 1} โ {๐ฅ pilot0 โช ๐ฅ null
0 } (82)
where ๐๐ โ {{1, 2, . . . ,๐๐ โ 1} โ {๐๐/2}}, {๐๐} โฉ {๐๐ โ๐๐} = โ . The pilot tones are given by
๐๐[๐] = ๐๐๐๐,๐ , ๐ โ ๐ฅ pilot0,๐ , ๐ โ {0, . . . , ๐Tx โ 1} (83)
๐๐[๐] = 0, ๐ /โ ๐ฅ pilot0,๐ (84)
where {๐๐,๐} are arbitrary. For ๐พ = 1, the maximum numberof transmit antennas this design can support is ๐
2๐ฟ0โ 1 which
is the same scenario as discussed in [CDM-F; Null] design.
F. [TFDM; Null / C-F] Design
In this design, pilots of different antennas are decoupled byTFDM design, while intra-antenna mirror tone interferencesare addressed by mirror null tones, and inter-antenna mirrortone interferences are suppressed by code design across fre-quency domain (C-F) or mirror null tones across differentantennas. This design uses ๐พ = 2 symbols over whichnon-zero pilots (with the index set ๐ฏ๐,๐๐) of each antenna๐ are spread out evenly. It requires ๐Tx๐ฟ๐ โค ๐ , and๐ฟ๐ โฅ 2๐ฟ0. The pilot index sets over two symbols for (2๐)thand (2๐+1)th antennas are chosen as ๐ฏ๐,๐2๐ and ๐ฏ๐,๐2๐+1 where๐๐ โ {{1, 2, . . . ,๐๐ โ 1} โ {๐๐/2} and ๐2๐ + ๐2๐+1 = ๐๐.If ๐Tx is an odd number, the pilot index set over the twosymbols for the last antenna (๐ = ๐Tx โ 1) is ๐ฏ๐,๐๐/2 whichconsists of mirror pairs excluding self-mirror tones. ๐ฏ๐,๐๐ isdivided into two decimated sets ๐ฏ๐,๐๐,๐ and ๐ฏ๐,๐๐,๐, consistingof even elements and odd elements (their values can be evenor odd) of ๐ฏ๐,๐๐ , respectively. For each antenna pairs 2๐ and2๐ + 1, ๐ฏ๐,๐2๐,๐ and ๐ฏ๐,๐2๐+1,๐ form mirror pairs, and so do๐ฏ๐,๐2๐,๐ and ๐ฏ๐,๐2๐+1,๐. Antennas 2๐ and 2๐+1 transmit pilotson ๐ฏ๐,๐2๐,๐ and ๐ฏ๐,๐2๐+1,๐, respectively, in the first symbol,and ๐ฏ๐,๐2๐,๐ and ๐ฏ๐,๐2๐+1,๐, respectively, in the second symbol.Mathematically, the index sets are given by
๐ฅ pilot0,2๐ = ๐ฏ๐,๐2๐,๐, ๐ฅ pilot
0,2๐+1 = ๐ฏ๐,๐2๐+1,๐, (85)
๐ฅ pilot1,2๐ = ๐ฏ๐,๐2๐,๐, ๐ฅ pilot
1,2๐+1 = ๐ฏ๐,๐2๐+1,๐ (86)
๐ฏ๐,๐2๐+1,๐ = {๐ โ ๐ฏ๐,๐2๐,๐}, (87)
๐ฏ๐,๐2๐+1,๐ = {๐ โ ๐ฏ๐,๐2๐,๐} (88)
๐ฅ pilot0,๐Txโ1 = ๐ฏ๐,๐๐/2,๐ & ๐ฅ pilot
1,๐Txโ1 = ๐ฏ๐,๐๐/2,๐, odd ๐Tx
(89)
๐ฏ๐,๐Txโ1,๐ = {๐ โ ๐ฏ๐,๐Txโ1,๐}, odd ๐Tx (90)
โ๐ = {0, . . . , ๐ โ 1} โ ๐ฅ pilot๐ , even ๐Tx (91)
โ๐ = {0, . . . , ๐ โ 1} โ {๐ฅ pilot๐ โช ๐ฏ๐,๐Txโ1}, odd ๐Tx.
(92)
For each antenna pair (2๐, 2๐+1), the C-F design is given by
๐2๐,0[[๐ฏ๐,๐2๐,๐]๐] = ๐๐๐0,๐ ,๐ = 0, 1, . . . , ๐ฟ๐โ1 โ 1 (93)
๐2๐+1,0[[๐ฏ๐,๐2๐+1,๐]๐ฟ๐โ1โ๐] = ๐๐2๐๐2๐/๐ฟ๐ ๐โ๐๐0,๐ ,
๐ โ {๐ฟ,๐ฟ+ 1, . . . , ๐ฟ๐โ1 โ ๐ฟ} (94)
๐2๐,1[[๐ฏ๐,๐2๐,๐]๐] = ๐๐๐1,๐ ,๐ = 0, 1, . . . , ๐ฟ๐โ1 โ 1 (95)
๐2๐+1,1[[๐ฏ๐,๐2๐+1,๐]๐ฟ๐โ1โ๐] = ๐๐2๐๐ผ(2๐+1)/๐ฟ๐ ๐โ๐๐1,๐ ,
๐ผ โ {๐ฟ,๐ฟ+ 1, . . . , ๐ฟ๐โ1 โ ๐ฟ} (96)
where {๐0,๐} and {๐1,๐} are arbitrary. For different antennapairs, {๐0,๐}, {๐1,๐}, ๐, and ๐ผ can be independently chosen.For an odd ๐Tx, ๐๐Txโ1,0[๐ โ ๐ฏ๐,๐๐/2,๐] and ๐๐Txโ1,1[๐ โ๐ฏ๐,๐๐/2,๐] can be set to arbitrary unit amplitude symbols. Ateach of the two symbols, this design uses ๐Tx๐ฟ๐โ1 pilottones for an even ๐Tx and (๐Tx โ 1)๐ฟ๐โ1 + ๐ฟ๐ pilot tones(including null tones) for an odd ๐Tx.
G. [CDM-F or FDM; C-T] Design
This design uses two OFDM symbols. In CDM-F, each an-tenna transmits ๐Tx๐ฟ๐ constant-amplitude pilot tones in eachsymbol, while in FDM each antenna transmits ๐ฟ๐ constant-amplitude pilot tones and (๐Txโ1)๐ฟ๐ null pilot tones in eachsymbol. These pilot indexes are all mirror-pairs (may includeself-mirror tones). The index sets are given by
๐ฅ๐ =
๐Txโ1โช๐=0
๐ฏ๐,๐๐ , ๐๐ โ {0, 1, . . . ,๐๐ โ 1}, (97)
โ๐ = {0, 1, 2, . . . , ๐ โ 1} โ ๐ฅ๐, (98)
๐ฅ๐,๐ =
{ ๐ฅ๐, CDM-F๐ฅ๐,๐ = ๐ฏ๐,๐๐ , FDM
(99)
where ๐๐ โ= ๐๐ if ๐ โ= ๐ and {๐๐} = {(๐๐ โ ๐๐)๐๐}.The choice of ๐ = 0 in ๐ฟ๐ and ๐๐ yields minimum pilot
overhead. The pilots from different antennas are decoupledby CDM-F or FDM, while the mirror tone interferences aresuppressed by C-T over two symbols. For CDM-F, eachantenna transmits constant amplitude pilots on ๐ฅ๐. For FDM,the ๐th antenna transmits constant amplitude pilots on ๐ฏ๐,๐๐and null tones on {๐ฅ๐ โ ๐ฏ๐,๐๐}. The C-T design is describedby the relationship of the pilot vectors at the second symbol tothose at the first symbol. For CDM-F, the pilot vectors at thesecond symbol are just
โโ1 times the corresponding pilotvectors at the first symbol. For FDM, we can have severalapproaches such as: (i) the pilot vectors at the second symbolare
โโ1 times those at the first symbol, (ii) the pilot toneswith indexes less than ๐/2 remain the same over two symbols,while the remaining pilots change polarities across the twosymbols, or (iii) for each antenna pair with mirror indexsets ๐ฏ๐,๐๐ and ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ , the antenna using ๐ฏ๐,๐๐ transmits thesame pilots on both symbols, while the other antenna changespolarities of pilots across the two symbols.
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2261
H. [TFDM / CDM-T; Null / C-T] Design
In this design, pilots from different antennas are decoupledthrough TFDM and CDM-T, while mirror tone interferenceswithin each antenna are addressed by mirror null tones,and those across antennas are suppressed by mirror nulltones and C-T. The antennas are divided into ๐พ๐ groups.Pilot tones of different groups are disjoint via TFDM (orTDM). Each antenna group ๐ (for an even ๐Tx) consistsof an even number of antennas, say 2๐๐, and uses ๐ฏ๐,๐๐and ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ where ๐๐ โ {{1, 2, . . . ,๐๐ โ 1} โ {๐๐/2}},๐๐ โ= ๐๐ if ๐ โ= ๐, ๏ฟฝฬ๏ฟฝ = ๐๐ โ ๐, ๐๐ + ๐๏ฟฝฬ๏ฟฝ = ๐๐, and๐ โ {1, . . . , log2(๐/(2๐พ๐๐ฟ0)}. ๐ฏ๐,๐๐ and ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ are eachdivided into two subsets of the same cardinality, denoted by๐ฏ๐,๐๐,0, ๐ฏ๐,๐๐,1, ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,0, and ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,1, such that ๐ฏ๐,๐๐,0 and๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,1 form mirror pairs and so do ๐ฏ๐,๐๐,1 and ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,0. Thisdivision need not be a splitting of even and odd elements asrequired in Section IV-F.
The first half of the antennas within group ๐ transmiton ๐ฏ๐,๐๐,0 during the first ๐๐ (โฅ ๐๐) symbols and on๐ฏ๐,๐๐,1 during the next ๐๐ symbols. The other half of theantennas transmit on ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,0 during the first ๐๐ symbols andon ๐ฏ๐,๐๏ฟฝฬ๏ฟฝ,1 during the next ๐๐ symbols. These two subsets ofantennas within each group are of TFDM type. The pilot forantenna ๐ of group ๐ on corresponding subcarrier [๐ฏ๐,โ,โ]๐ at๐th symbol is given by ๐๐๐๐,๐๐๐2๐๐๐/๐๐๐๐,๐,๐ where {๐๐,๐}are arbitrary and {๐๐,๐,๐} are arbitrary constant amplitudesymbols, i.e., antennas transmitting on the same subcarrierfollow CDM-T design. Each group ๐ requires 2๐ฟ๐ tones over2๐๐ symbols, and setting ๐ = 0 in ๐ฟ๐ and ๐๐ yields asmaller overhead.
For an odd ๐Tx, a dummy antenna can be fictitiouslyadded in the design to have an even ๐Tx. Two more-efficientalternatives are described below where pilots for the first๐Txโ1 (even) antennas are developed according to the above-mentioned design. In the first alternative, the last antennatransmits pilots on ๐ฏ๐,0 with ๐ โฅ 1 over one symbol using C-F design. In the second alternative, the last antenna transmitsarbitrary constant amplitude pilots on ๐ฏ๐,๐๐+1 with ๐ โฅ 0over two symbols using C-T design, or on ๐ฏ๐,๐๐+1,๐ with๐ โฅ 0 at the first symbol and on ๐ฏ๐,๐๐+1,๐ at the secondsymbol (i.e., Null design over two symbols).
I. [TDM; Null] Design
In this design, pilots of different antennas are decoupled bymeans of TDM while mirror tone interferences are suppressedby means of null tones. All antennas transmit on the sameset of subcarriers, but each antenna transmits during a dif-ferent OFDM symbol. Thus, ๐พ = ๐Tx OFDM symbols arerequired, while data can be transmitted on other subcarriersin a pilot-data-multiplexed format. This design can handlea larger number of null guard tones than other designs.Let ๐ฅ guard denote the tone index set for the null guardtones. First, obtain ๐ฏ๐,๐ such that {๐ฏ๐,๐ โฉ ๐ฅ guard} = โ ,๐ โ {{1, 2, . . . ,๐๐โ1}โ{๐๐/2}}. With ๏ฟฝฬ๏ฟฝ โ ๐๐โ๐, wehave ๐ฏ๐,๏ฟฝฬ๏ฟฝ = {๐ โ ๐ฏ๐,๐ modulo ๐}. Then, the index sets
are given by
๐ฅ pilot๐,๐ =
{ ๐ฏ๐,๐, if ๐ = ๐โ , else
(100)
๐ฅ null๐,๐ =
{ ๐ฏ๐,๏ฟฝฬ๏ฟฝ, if ๐ = ๐๐ฏ๐,๐ โช ๐ฏ๐,๏ฟฝฬ๏ฟฝ, else
(101)
โ๐ = {0, 1, 2, . . . , ๐ โ 1} โ {๐ฏ๐,๐ โช ๐ฏ๐,๏ฟฝฬ๏ฟฝ โช ๐ฅ guard}. (102)
The non-zero pilot tones can have arbitrary phases {๐๐,๐} as
๐๐,๐[๐] = ๐๐๐๐,๐ , ๐ โ ๐ฅ pilot๐,๐ , ๐ โ {0, 1, . . . , ๐Tx โ 1}. (103)
For minimum overhead and operability with largest numberof null guard tones, ๐๐ should be set to ๐0.
J. Other Designs
Other variations or combinations of the above designs arealso possible. For example, [CDM-F; C-T] and [FDM; C-T] designs can be combined as [CDM-F/FDM; C-T] whereantennas are divided into two groups such that there is nomirror pair across the two groups, and the first group applies[CDM-F; C-T] while the other group uses [FDM; C-T]. Othercombinations may also be possible, but for practical designsimplicity we skip further investigation in this direction.
V. SIMULATION RESULTS AND DISCUSSIONS
A. Parameter Setting
System parameters in the simulation are ๐Tx = 2, ๐Rx =2, ๐ = 64, 6 left and 6 right null guard tones, ๐ -ary QAMwith ๐ = 16, and a Rayleigh fading channel having anexponential power delay profile (3 dB per tap decay factor)with 4 sample-spaced taps. The FI I/Q imbalances are set to๐ผ =
๐๐ผ๐ก
๐๐๐ก
=๐๐ผ๐
๐๐๐= 1.09648 (= 0.4 dB), and ฮ๐ = ๐๐ผ๐ก โ ๐๐๐ก
= ๐๐ผ๐ โ ๐๐๐ = 3 โ 2. The FD I/Q imbalances are modeledby 3-tap filters (hence, ๐ฟ = 8) with discrete-time impulseresponses of [0.01, 0.9999, 0.01] and [0.015, 0.9998, 0.01] forthe transmit I and Q branches, and [0.012, 0.9997, 0.018],and [0.01, 0.9997, 0.02] for the receive I and Q branches.For performance comparison, we use the pilot design from[21] as SISO Reference. We use the design from [24] asMIMO Reference 1 and an FDM design from [11]3 as MIMOReference 2. In all methods, the estimators from Eqns. (16)and (17) are utilized and for BER results the maximumlikelihood (ML) detection (joint detection of mirror tones) isapplied. The energies of a non-zero pilot tone in the SISOreferences and the MIMO reference 1 are set to be the sameas the average bit energy of data. The total pilot energy iskept the same for all pilot designs in each case (SISO orMIMO). For the SISO results, the system parameters are thesame except for the number of antennas.
2These values are within typical ranges (e.g., see [4], [22], [27], [28]).3To illustrate the performance degradation when I/Q imbalance is not
considered in the pilot design, we pick a particular FDM design. Some ofthe pilot designs from [11] with 2๐Tx๐ฟ0 pilot tones may yield the sameestimation performance as the proposed ones.
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2262 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
5 10 15 20 25
10โ3
10โ2
(EK/N)/N
0 (dB)
Cha
nnel
Est
imat
ion
MS
E
Proposed (Theo.)SISO Ref. (Theo.)Proposed (Simu.)SISO Ref. (Simu.)
Fig. 3. Channel estimation MSE comparison of different pilot designs in aSISO OFDM system.
2 4 6 8 10 12 14
0.001
0.002
# of null guard tones
Cha
nnel
Est
imat
ion
MS
E
Proposed MIMO
Proposed SISO
MIMO Ref. 1
SISO Ref.
Fig. 4. Channel estimation MSE comparison of different pilot designs whilevarying the number of guard tones, ((๐ธ๐พ/๐)/๐0 = 21 dB)
B. Estimation and BER Performance
Fig. 3 shows the channel estimation MSEs (MSE๐+MSE๐)from simulation and the theoretical MSEs ((20) for the ref-erence design and (30) for the proposed design) for a SISOsystem. The proposed design outperforms the SISO referencewhich experiences a flooring effect at high SNR. The nullingof some of the subcarriers results in a breakdown of the codingused in [21] to eliminate the mirror tone interferences anddegrades the channel estimation MSE. The theoretical MSEsmatch the simulation MSEs very well.
The effects of guard tones on the pilot designs are illustratedin Fig. 4 using the theoretical MSEs. The MSE degradationof the SISO reference pilot design is observed to be moresensitive to the number of null guard tones than the MIMOReference 1 which is due to different coding strategies adoptedin the pilot designs. The larger MSE level of MIMO pilot
10 15 20 25 3010
โ4
10โ3
10โ2
10โ1
100
(EK/N)/N
0 (dB)
Cha
nnel
Est
imat
ion
MS
E
Proposed (Theo.)
MIMO Ref. 1 (Theo.)
MIMO Ref. 2 (Theo.)
Proposed (Simu.)
MIMO Ref. 1 (Simu.)
MIMO Ref. 2 (Simu.)
Proposed (CRB)
Fig. 5. Channel estimation MSE comparison of different pilot designs in a2ร 2 MIMO OFDM system.
TABLE IIIPILOT OVERHEAD COMPARISON
Design Overhead(# of Tones)
(MIMO) [TDM; TD/C-F] 2๐Tx๐ฟ0
(MIMO) [CDM-T; C-T] 2๐Tx๐ฟ0
(MIMO) [CDMF; Null] 2๐Tx๐ฟ0
(MIMO) [FDM; Null] 2๐Tx๐ฟ0
(MIMO) [TFDM; Null/C-F] 4๐ฟ0โ๐Tx/2โ(MIMO) [TFDM/CDM-T; Null/C-T] 2๐Tx๐ฟ0
(MIMO) [TDM; Null] 2๐Tx๐ฟ0
(MIMO) Design from [24] 2๐Tx๐(SISO) Design from [21] ๐
designs if compared to the SISO designs is due to the factthat the MSE presented is the sum of MSEs of all parametersunder estimation and that MSE increases with the number ofparameters under estimation.
Fig. 5 presents the channel estimation MSEs from sim-ulation, the theoretical MSEs, and the CRB from (31) forthe MIMO system. The proposed design outperforms bothreference designs. Although MIMO Reference 2 gives optimalestimation performance for systems without I/Q imbalances,it causes a substantial performance degradation in the pres-ence of I/Q impairments. MIMO Reference 1 has a slightperformance loss due to its non-optimality for systems withguard bands. The theoretical MSEs in Section III give an exactmatch to the simulation MSE results which show no noticeabledifference from the CRB. This is due to the fact that the mildfrequency-selectivity of the I/Q imbalance makes the whitenoise assumption used in the pilot design valid, and then theestimators become ML with the Gaussian signal model whichare known to approach the CRB.
In Fig. 6, the (uncoded) BER performances of differentpilot designs are compared for the MIMO system. Similarconclusions for the MSE performance apply for the BERperformance. Although the BER improvement is small, theproposed pilot designs reduce the pilot overhead greatly (see
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MINN and MUNOZ: PILOT DESIGNS FOR CHANNEL ESTIMATION OF MIMO OFDM SYSTEMS WITH FREQUENCY-DEPENDENT I/Q IMBALANCES 2263
5 10 15 20 2510
โ5
10โ4
10โ3
10โ2
10โ1
100
Eb/N
0 (dB)
BE
R
Proposed
MIMO Ref. 1
MIMO Ref. 2
No IQ
No IQ, Smart Rx
Fig. 6. BER comparison of different pilot designs for 16-QAM in a 2 ร 2MIMO OFDM system.
Table III) which increases the data capacity/throughput, andcan also reduce the data detection latency if compared toMIMO Reference 1. When the receiver does not know theabsence of I/Q imbalance, its performance is the same asthe cases with I/Q imbalance. But when the receiver knowsthe absence of I/Q imbalance (denoted as the Smart Rx), itsperformance is better than those with I/Q imbalance exceptat high SNR where the ML receiver gains frequency diversityprovided by the I/Q imbalance.
Fig. 7 presents the MSE performance of the proposedpilot design under various FI I/Q imbalance levels. The low,medium, and high I/Q imbalance cases, respectively, corre-spond to (๐ผ = 0.1 dB, ฮ๐ = 1โ), (๐ผ = 0.7 dB, ฮ๐ = 10โ),and (๐ผ = 2 dB, ฮ๐ = 30โ) 4. The no I/Q imbalance caserefers to (๐ผ = 0 dB, ฮ๐ = 0โ) and without FD I/Q imbalance,while the receiver still considers there is I/Q imbalance andestimates in the same way as in the other cases. For the SmartRx case, the MSE drops proportional to the number of tapsestimated. From Fig. 7, it is clear that the channel estimationMSE is not affected by the FI I/Q imbalance level.
In Fig. 8, to show the effect of different levels of FDI/Q imbalance, we have evaluated an additional set of FDI/Q imbalance filters with impulse responses of [0.9999, 0.01]and [0.9999, 0.015] for the transmit I and Q branches, and[0.9998, 0.018], and [0.9999, 0.01] for the receive I and Qbranches. This corresponds to an equivalent channel with๐ฟ = 6 taps. We have also included the performance of theSmart Rx where the number of parameters under estimationis ๐ฟ = 4 as opposed to 2๐ฟ = 12 and 2๐ฟ = 16 in theother two cases. The MSE gaps are due to the differencesin the numbers of parameters under estimation. Hence, thefrequency-selectivity level of I/Q imbalance (more specifically,the lengths of the equivalent direct and mirror channels) canaffect the total channel estimation MSE.
4The values for the high case are outside of the current expected rangesfor hardware in use today (see [27], [28]). However, as semiconductordownscaling continues, higher values for I/Q imbalance such as these couldbe seen.
12 14 16 18 20 22 24 26 28 30 32
10โ4
10โ3
10โ2
(EK/N)/N
0 (dB)
Cha
nnel
Est
imat
ion
MS
E
No IQ Im.(Theo)With IQ Im. (Theo)No IQ Im. (Simu)Low IQ Im. (Simu)Med IQ Im. (Simu)High IQ Im. (Simu)No IQ, Smart Rx (Simu)
Fig. 7. Effects of different levels of FI I/Q imbalance on a proposed pilotdesign in a 2ร 2 MIMO OFDM system.
15 20 25 30
10โ4
10โ3
10โ2
(EK/N)/N
0 (dB)
Cha
nnel
Est
imat
ion
MS
E
L = 8 (Theo.)L = 6 (Theo.)L = 4, Smart Rx (Theo.)L = 8 (Simu.)L = 6 (Simu.)L = 4, Smart Rx (Simu.)
Fig. 8. Effects of different levels of FD I/Q imbalance on a proposed pilotdesign in a 2ร 2 MIMO OFDM system.
C. Comparison in Other Aspects
All of the proposed designs give the same estimation perfor-mance since they meet the criteria defined by (24). However, ifother system constraints or impairments are considered, therecould be advantages of one over another. For example, somedesigns such as [CDM-T; C-T] and [TDM; Null] are betterequipped to facilitate a larger null guard band than otherssince the gaps between used subcarriers is larger. However,they utilize more OFDM symbols, and thus would introducemore latency at the receiver.
The proposed designs can be applied to preamble as wellas pilot-data-multiplexed symbols because the design problemwas formulated so. This reduces the pilot overhead and thelatency at the receiver. The existing designs in [15], [16],[18]โ[20], [22], [24] only apply for preamble since no datatones are considered in the designs. The proposed designsare applicable for both SISO and MIMO OFDM systemswhile the existing methods such as [15]โ[22] only address forSISO OFDM systems. In term of the estimation performance,
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2264 IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 58, NO. 8, AUGUST 2010
the proposed designs yield white-noise optimality and closeto ideal-optimal results for colored noise at the demodulatoroutput. The existing methods such as [19], [21], [22] onlyprovide a suboptimal estimation performance which degradesfor systems with guard bands while the proposed designscan overcome this shortfall as well. The comparison of pilotoverhead for several pilot designs is presented in Table III.Under the considered system setting, the proposed designsrequire 1/4 of the overhead of the SISO reference design,and 1/8 of that of the MIMO Reference 1.
VI. CONCLUSIONS
We have developed efficient pilot designs for estimationof the equivalent channel responses incorporating FI and FDtransmitter and receiver I/Q imbalances in MIMO OFDMsystems. The receive filter output noise samples in the pres-ence of FD receiver I/Q imbalance are colored and generallyunknown, and hence development of exactly-optimal pilotdesigns is impractical. However, in practice the frequency-selectivity of I/Q imbalance is very small and hence our pilotdesigns developed based on the white noise condition areobserved to yield essentially the same performance as theCRB. To be applicable in the pilot-data-multiplexed format,not only the data and pilot tones need to be disjoint butalso each type (pilot or data) should occupy only on mirrorsubcarrier pairs. The minimum number of pilot tones requiredfor the considered estimators is at least doubled if compared tothe systems without I/Q imbalance. To suppress inter-antennainterferences, the pilots of different transmit antennas need tosatisfy the condition in (42) as required in systems without I/Qimbalance but also an additional condition due to the mirrortone interference as given in (47).
We have also observed that the frequency-selectivity levelof I/Q imbalance can affect the total channel estimation MSEwhile the FI I/Q imbalance level does not. The proposed pilotdesigns are more efficient than the existing designs in termsof overhead, estimation performance, and general applicability(preamble or pilot-data-multiplexed setup, SISO or MIMO,with or without guardbands). Our MIMO pilot designs canalso be extended to OFDMA downlink systems and OFDM-based cooperative communication systems. For OFDMA up-links, the applicability of the proposed pilot designs dependson how resources are channelized for different users.
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