a novel method of single phase quasi z-source · pdf filea novel method of single phase quasi...

SSRG International Journal of Electronics and Communication Engineering (SSRG-IJECE) – volume1 issue7 Sep 2014

ISSN: 2348 – 8549 www.internationaljournalssrg.org 2

A Novel Method of Single Phase Quasi Z-Source Converter M.R.Rushikesh1 T.Shivashankar2

1M.Tech Scholar, 2Assoc.Professor, Department of EEE, AVR& SVR College of Engineering and Technology, India

ABSTRACT:AC-AC converter convert fixed AC-input voltage in to variable ac output voltage by using ac voltage controllers. The single-phase z-source PWM ac-ac converter in this the converter works in a discontinuous current mode and do not share the same ground. Due to this output voltage does not provides continuous sine wave. And another is single phase ac-ac converter based on quasi-z-source technology. This converter shares the same ground but commutation technique is not safe commutation technique and it uses the snubber circuit. Due to this harmonics are occurs and voltage, power factor will comes low. In order to overcome the above problems in the traditional AC-AC converters a new technology of AC-AC converter is proposed by using a Z-source converter with this new technology, the proposed converter offers the advantages over traditional AC-AC converter which operates in continuous current mode, safe commutation technique is used and no snubber circuit is used which provides good voltage regulation, improves power factor and provides less harmonics.

Keywords: Buck-Boost Capability, pulse width modulation (PWM), quasi-Z-source converter, safe-commutation, single-phase ac-ac converter, total harmonics distortion (THD)

I. INTRODUCTION

This template, The modified single phase quasi Z-source ac-ac converter [1], the conventional Z-source PWM ac–ac converters found in [2] and [3] have a significant drawback in that the input current is operated in a discontinuous current mode. When the input current operates in this discontinuous current mode, its waveform is non-sinusoidal, which increases the input current THD. Moreover, the peak of the input current in the discontinuous current mode is higher than it is in the continuous current mode. Another drawback is that the input voltage and the output voltage of the original Z-source PWM ac–ac

converter [3] do not share the same ground. In an effort to overcome the inconveniences of the traditional Z-source ac–ac converters, a single-phase quasi-Z-source ac–ac converter has recently been proposed in [4].In this project, a new technology of single phase quasi z source converter presenting. The presenting converter inherits all of the advantages of the traditional single-phase Z-source ac-ac converter it has buck-boost capabilities and can maintain or reverse the output phase angle all the while sharing the same ground. Moreover, the presenting ac-ac converter has the following unique advantages: a smaller converter size, an operation in the continuous current mode which enables special features such as a reduction in harmonic current, and improving power factor. A safe-commutation strategy is provided for the proposed converter that eliminates voltage spikes on the switches without the need for a snubbed circuit.

1.1 PROBLEM FORMULATION The For ac–ac power conversion, the most popular technologies are indirect ac–ac converters with a dc link, matrix converters, and direct pulse width modulation (PWM) ac–ac converters. However, they have some significant disadvantages, such as a low input power factor, a high total harmonic distortion (THD) in the source current, and poor power transfer efficiency. To overcome the above problems in ac controllers can be replaced by pulse width modulation (PWM) ac chopper controllers, which have the following features: provisions for a better power factor and efficiency, a low harmonic current in the line.

II A NOVEL METHOD FOR AC-AC CONVERTER The proposed converter is a new technology of single phase quasi z source converter without input or output filters is presented is as shown in fig 3.1. The proposed converter inherits all of the advantages of the traditional single-phase Z-source ac-ac converter;


ISSN: 2348 – 8549 www.internationaljournalssrg.org Page 13

it has buck-boost capabilities and can maintain or reverse the output phase angle all the while sharing the same ground. A safe-commutation strategy is provided for the proposed converter that eliminates voltage spikes on the switches without the need for a snubber circuit. The proposed converter consists of four igbt’s ,two capacitors and two inductors are used in converter as shown in fig 3.1 in below which in this the switches that is igbt’s are connected in the parallel .The operating principles are compared to those of a conventional single-phase Z-source ac-ac converter, are thoroughly outlined.

Fig 2.1 A novel method for ac-ac converter

2.1 COMMUTATION PROBLEM The two bidirectional switches, S1j and S2j (j=a, b), are able to block voltage and conduct current in both directions. Because true bidirectional switches are not available to date, they can be implemented by the connection of two diodes and two insulated gate bipolar transistors (IGBTs) in an anti-parallel (common emitter back to back) manner, as shown in Fig. 3.1. The single-phase quasi-Z-source ac–ac converters shown in [1],[2] and[3] have a commutation problem. A change in the current due to the PWM switching will result in current and voltage spikes. Note that the current spikes are generated by the short-circuit or shoot-through path; the voltage spikes are produced because of the current derivative of the inductance. Both types of spikes will destroy the switches due to stress. In order to understand the commutation problem found in single phase quasi-Z-sou2.2rce ac–ac converters, take the circuit shown in Fig. 3.1 as an example. Suppose that S1j (j = a, b) is turned ON and conducts current. After a time, we want to commutate the current to S2j (j = a, b). Theoretically, the switching must be instantaneous and simultaneous. For practical reasons, however, we have to take into account the finite switching times and delays found in the drive circuits and switches. Therefore, if S2j (j = a, b) is turned ON before S1j (j

= a, b) is turned OFF, a short-circuit path is established through S1j–C1–S2j –C2 causing current spikes that will destroy the devices. Similarly, if S1j (j1= a, b) is turned OFF before S2j (j = a, b) is turned ON, there will be a junction that connects inductors L1 and L2 resulting in voltage spikes that will destroy the switches. In some previous methods, a lossysnubber circuit is added for each switch in order to limit the voltage overshoot.

2.2 OPERATING PRINCIPLES The switching strategy of the modified single phase quasi-Z-source ac–ac converter in the in-phase mode where the input voltage and the output voltage are in the same phase, if the input voltage Vi>0, switches S1a and S2b are fully turned ON while S1b and S2a are modulated complementary to the dead time. If Vi<0, switches S1b and S2a are fully turned ON while S1a and S2b are modulated complementary to the dead time. In the out-of-phase mode in which the input voltage and the output voltage are in opposite phases, if Vi>0, switches S1b and S2a are fully turned ON while S1a and S2b are modulated complementary to the dead time. If Vi<0, switches S1a and S2b are fully turned ON while S1b and S2a are modulated complementary to the dead time. As indicated in Fig 3.2, D refers to the equivalent duty ratio and T is the switching period. Fig 3.3 to Fig 3.6 shows the operation states in the in phase mode when Vi> 0. Switches S1a and S2b are fully turned ON while S2b and S2a are modulated complementary to the dead time.

Fig 2.2 Switching pattern of the new technology of single phase quasi z-source converter (D is the duty ratio; T is the switching period).



2.3 OPERATING STATES STATES 1

In state 1, as shown in Fig. 3.3, S1a is turned ON and conducts the current during the increasingly positive cycle of the input voltage; S1b is turned ON and conducts the negative current from the load to the source, if possible; S2b is turned ON for commutation purposes. S1b is then turned OFF while S2a has not yet turned ON, and so there are two commutation states that occur.

Fig 2.3 operating state 1 STATES 2

If ii + iL2> 0, the current flows along a path from S1a, as shown in Fig. 3.4.


If ii+iL2<0, the current flows along a path from S2b as shown in Fig 3.5.


S2a is turned ON and conducts the current from the source to the load; S2b is turned ON and conducts the negative current from the load to the source, if possible; S1ais turned ON for commutation purposes. In these switching patterns, the current path is always

continuous regardless of the current direction. This eliminates the voltage spikes during the switching and commutation processes. The analysis when vi<0 is similar to that found when vi>0. The dotted line Fig 2.6 operating state 4

2.4SWITCHING CONTROL SEQUENCE: The below Table 1 explains the operation states of the converter during different modes for buck operation in the in-phase mode and for boost operation in the out-phase mode.

Mode v Switch "on" states State1 State2 Active Commutation Active Commutation

In –

phase

>0 S1a,S1b S2b S2a,S2b S1a

<0 S1a,S1b S2a S2a,S2b S1b

Out

of Phase

>0 S1a,S1b S2a S2a,S2b S1b

<0 S1a,S1b S2b S2a,S2b S1a

Table 1 The switching sequences for the operations of the a new technology of single phase quasi z-source converter.

2.5 CIRCUIT ANALYSIS A circuit analysis of the proposed single-phase quasi-Z-source ac–ac converter begins ith the following assumptions

The converter is operating in the continuous conduction mode; the parasitic resistances of L1 and L2 are the same and equal and denoted by rl. The equivalent series resistances of C1 and C2are the same and equal and denoted by RC; The on-resistance of the switches S1j and S2j (j = a, b) are the same and equal and denoted by rs and the switching frequency is more than the frequency of the input and output voltages. Ignoring the dead time effects, the modified single-phase quasi-Z-source ac–ac converter has two



operating states in one switching period, which are denoted as state 1 and state 2, as shown in Fig. 4(a) and (b), respectively. In state 1, S1j is turned ON and S2j is turned OFF, as shown in Fig. 4(a). The time interval of this state is DT, where D is the equivalent duty ratio ofS1j and T is the switching period, as shown in Fig. 3.2.

Fig 2.7 Equivalent circuits for the proposed converter in state1

퐿 00 퐿

0 00 0

0 00 0

퐶 00 퐶

푑푑푡⎣⎢⎢⎡ 푖

(푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

=

⎣⎢⎢⎢⎡−(푟 + 푟 ) −푟

−푟 −(푟 + 푟 + 푟 )−1 0 0 −1

1 0 0 1

− 0 0 0⎦

⎥⎥⎥⎤∗

⎣⎢⎢⎡푖 (푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

+푣 (푡)

000

(3.1)

Therefore, In state 2 S1j is turned OFF and S2j is turned ON, as shown in Fig 2.8. The time interval of this state is (1-D)*T.

Fig 2.8 Equivalent circuits for the proposed converter in state2.

퐿 00 퐿

0 00 0

0 00 0

퐶 00 퐶


(푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

=

⎣⎢⎢⎢⎡−(푟 + 푟 + 푟 ) −푟

−푟 −(푟 + 푟 )−1 0 0 −1

1 0 0 1

− 0 0 0⎦

⎥⎥⎥⎤∗

⎣⎢⎢⎡푖 (푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

+푣 (푡)

000

(3.2)

Therefore From eqn (3.1) and (3.2) the following average equation is obtained.

퐿 00 퐿

0 00 0

0 00 0

퐶 00 퐶


(푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

=

⎣⎢⎢⎢⎡−

(푟 + 푟 )− (1− 퐷)푟 −푟−푟 −(푟 + 푟 )− 퐷푟

−퐷 1− 퐷 1 −퐷 −퐷

퐷 −(1− 퐷) −(1 −퐷) 퐷

− 0 0 0 ⎦

⎥⎥⎥⎤∗

⎣⎢⎢⎡푖 (푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

+푣 (푡)

000

(3.3) In the Steady State, it is found that 퐿 00 퐿

0 00 0

0 00 0

퐶 00 퐶 ⎣

⎢⎢⎡ 푖

(푡)푖 (푡)푣 (푡)푣 (푡)⎦

⎥⎥⎤

=0000

(3.4)



Thus 푣

=퐷(2퐷 − 1)푅

(2퐷 − 1) 푅 + 푟 + (2퐷 − 2퐷 + 1)푟 +퐷(1 −퐷)푟 푣

푣

=(1 −퐷)(2퐷 − 1)푅 − 푟 − (1− 퐷)푟 − 퐷(1 −퐷)푟

(2퐷 − 1) 푅 + 푟 + (2퐷 − 2퐷 + 1)푟 +퐷(1 −퐷)푟 푣

푖 =퐷

2퐷 − 1푣푅

푖 =

(3.5) The Output voltage gain and inductor L2 current gain can be defined, respectively as

퐾 = = ∗ ( )( )

(3.6)

K_i=i_(L_2 )/i_i =(1-D)/D

퐾 = = (3.7)

It should be noted that the output profile in eqn (3.6) can be used for both resistive and inductive loading. The equations from (3.1) to (3.5) which lead to the output profile in (6) are unchanged when the load is inductive. In the ideal case (rs= rL= rC= 0 Ω) from (5), the following is obtained:

푣 =퐷

2퐷 − 1푣

푣 = 푣 (3.8)

푖 =퐷

2퐷 − 1푣푅

푖 =1 −퐷

2퐷 − 1푣푅

Table II shows the voltage and current equations for the ideal case (rs= rL= rC= 0 Ω) of the conventional single-phase Z-source ac–ac converters presented in [1] and [2] as well as for the single-

phase quasi-Z-source ac–ac converters. In Table II, Po is the output power and v0 is output voltage, i0 is the output current and D is the duty cycle ratio that is here different duty ratio is used for the different traditional converters is shown in table ii Ii,IL 1, IL 2, and Vi are the RMS values of the input current, inductor L1 current, inductor L2 current, and the input voltage, respectively. The quasi-Z-source network parameters are usually chosen according to the percentage of the current ripple through the inductor x%, and to the percentage of the voltage ripple across the capacitor y% [3], [4]. The parameter design of the conventional single-phase Z-source ac–ac converter [2],[3], the single-phase quasi-Z-source ac–ac converter [3], and the modified single-phase quasi-Z-source ac–ac converter.Istr, max,IS, RMS , and VCE, max are the peak current stress, the RMS current stress, and the collector-emitter voltage stress of the switches S1j, S2j(j=a, b) respectively. As shown in Table II, Conventional

Converter

Quasi-Z- Source Converter

Modified converter

푣푣 =

푣푣 퐷

2퐷 − 1 퐷

2퐷 − 1 퐷

2퐷 − 1 푣푣 퐷

2퐷 − 1 1− 퐷

2퐷 − 1 1 −퐷

2퐷 − 1

퐼 = 퐼 푝푣

푝푣

푝푣

퐼 푝푣

푝푣 1−퐷

퐷푝푣

Table II Governing equations of the conventional single-phase z-source Ac–AC converters

3.1 Simulation and results for boost converter with r-load Fig 3.1 shows the boost converter simulation circuit for R-Load which consists of four IGBT’s, two inductors and two capacitors. The converter operation is based on Multiple PWM technique .here in simulation circuit the parameters are having the values of inductors L1=L2=1mh, C1=C2=6.8µF and R=30 Ω



Fig 3.1 Simulation and results for boost converter with r-load

Fig 3.2 switching technique for boost converter with R-Load

Fig 3.3 Switching pulses for boost converter with R-Load

Fig 3.4 Boost converter output for R-Load The Fig 3.1 shows the simulation circuit for boost converter with R-load in this input voltage is given as 70 vrms, Fig 3.2 shows the switching technique which explains here the on states of the converter in this the switches S1a, S1b and s2b is in commutation state. Here the switching pulsesof this converter are shown in the Fig 3.3 and the Fig 3.4 shows the output voltage which shows the increase in voltage up to 100vrms.

3.2 Simulation and results for buck converter with r-load Fig 3.5 shows the buck converter simulation circuit for R-Load which consists of four IGBT’s, two inductors and two capacitors. The converter operation is based on Multiple PWM technique.

Fig 3.5 Simulation Circuit for Buck converter with R-Load



Fig 3.6 switching technique for buck converter with R-Load

Fig 3.7 Switching pulses for buck converter with R-load

Fig 3.8 Buck converter output for R-Load The Fig 3.5 shows the simulation circuit for buck converter with R-loadin this input voltage is given as 70 vrms, Fig 3.6 shows the switching technique which explains here the on states of the converter in this the switches S1a,S1band s2a is in commutation state. Here the switching pulsesof this converter are shown in the

Fig 3.7 and the Fig 3.8 shows the output voltage which shows the decrease in voltage up to 50vrms..

3.3 Simulation circuit and results for boost converter with rl-load The boost converter simulation circuit fig3.9 for RL-Load which consists of four IGBT’s, two inductors and two capacitors. The converter operation is based on Multiple PWM technique.

Fig 3.9 Simulation circuit for Boost converter with RL load

Fig 3.10 switching technique for boost converter with RL-Load

Fig 3.11 Switching pulses for boost converter with RL-Load



Fig 3.12 Boost converter output for RL-Load The Fig 3.9 shows the simulation circuit for boost converter with R-load in this input voltage is given as 70 vrms, Fig 3.10 shows the switching technique which explains here the on states of the converter in this the switches S1a,S1band s2b is in commutation state. Here the switching pulsesof this converter are shown in the Fig 3.11 and the Fig 3.12 shows the output voltage which shows the increase in voltage up to 100vrms.

3.4 1 Simulation circuit and results for buck converter with rl-load The buck converter simulation circuit fig 3.13 for RL-Load which consists of four IGBT’s, two inductors and two capacitors. The converter operation is based on Multiple PWM technique.

Fig 3.13Simulation circuit for Buck converter for RL-Load

Fig 3.14 switching technique for Buck converter in RL-Load

Fig 3.15 Switching pulses for buck converter with RL-Load

Fig 3.16 Buck converter output for RL-Load The Fig 3.13 shows the simulation circuit for buck converter with R-load in this input voltage is given as 70 vrms, Fig 3.14 shows the switching technique which explains here the on states of the converter in this the switches S1a,S1band s2a is in commutation state. Here the switching pulsesof this converter are shown in the Fig 3.15 and the Fig 3.16 shows the output voltage which shows the decrease in voltage up to 50vrms. 4.1 CONCLUSION A new type of AC-AC is proposed and corresponding simulated waveforms are done. The proposed converter has capability to reduces total harmonic distortion and improve power factor when



compared to traditional converter. The proposed converter provides the both buck and boost operation of voltages, this operation done by multiple pulse width modulation. Which operation is not observed in traditional AC-AC converters? Compared with traditional converter, the proposed converter shows the strong regulation capability to maintain the desired ac output voltage at low harmonics 2.15%, and power factor 0.99.The presented method is implemented by using MATLAB/Simulink software and simulation results are presented. 4.2 References

[1] Minh-Khai Nguyen and young-cheol Lim, “modified single phase quasi-z-source ac-ac converter,”IEEE power electron, vol 27, Jan 2012

[2] M. K. Nguyen, Y. G. Jung, and Y. C. Lim, “Single-phase ac–ac converter based on quasi-Z-source topology,” IEEE Trans. Power Electron., vol. 25,no. 8, pp. 2200–2210, Aug. 2010.

[3] Y.Tang, S. Xie, and C. Zhang, “Z-source ac–ac converters solving commutation problem,” IEEE Trans. Power Electron., vol. 22, no. 6, pp. 2146–2154, Nov. 2007.

[4] X.P.Fang, Z. Mian, and F. Z. Peng, “Single-phase Z-source PWMac–ac converters,” IEEE Power Electron. Lett. vol. 3, no. 4, pp. 121–124, Dec. 2005.

[5] F. Z. Peng, L. Chen, and F. Zhang, “Simple topologies of PWM ac– ac converters,” IEEE

[6] Fang z.peng,”z-source inverter”, IEEE Power Electron.In 2002. [7] M d Singh “ac-ac converters” Tata McGraw-hill Publishing

Author Profile:

T.siva shankar graduated from BVRIT, JNTU Hyderabad. Finished his M.Tech graduation from PDA engg college,Visweswaraiah technological university, belgaum. His area of Interest is machines.

T.Siva Shankar is an associate professor from A.V.R & S.V.R college of engineering and technology. Graduated from BVRIT, JNTU Hyderabad. Finished his M.Tech graduation from PDA engg college,Visweswaraiah technological university, belgaum. His area of Interest is machines.

M.R.Rushikesh graduated from SVCET, JNTU Anantapur. M.Tech from A.V.R & S.V.R college of engineering and technology,nandyal. His area of Interest is machines.

a novel method of single phase quasi z-source · pdf filea novel method of single phase quasi...

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