rf mems, bst, and gaas varactor system-level …response in complex modulation systems ... tunable...
Post on 23-Apr-2018
216 Views
Preview:
TRANSCRIPT
RF MEMS, BST, and GaAs Varactor System-LevelResponse in Complex Modulation Systems
Kamran Entesari,1 Gabriel M. Rebeiz2
1Department of Electrical and Computer Engineering, Texas A&M University,College Station, TX 778432Electrical and Computer Engineering Department, The University of California at San Diego,La Jolla, CA 92037
Received 19 January 2007; accepted 23 April 2007
ABSTRACT: This article presents the response of RF microelectromechanical systems (RF
MEMS), barium strontium titanate (BST), and gallium arsenide (GaAs)-based tunable filters
and reconfigurable matching networks to a wideband code-division-multiple-access signal cen-
tered at 1.95 GHz. The RF MEMS tunable filter and impedance tuner result in very low inter-
modulation distortion and spectral regrowth compared to their BST and GaAs counterparts.
The linearity of the BST and GaAs tunable networks improves considerably by using a series
combination of BST and GaAs varactors, but the RF MEMS-based networks still show the best
linearity of all three technologies. Also, it is shown that the reconfigurable networks, tuned
with capacitive RF MEMS can handle up to 1 W of RF power with no self-actuation. VVC 2007
Wiley Periodicals, Inc. Int J RF and Microwave CAE 00: 000–000, 2007.
Keywords: reconfigurable networks; RF microelectromechanical systems (RF MEMS); wideband
code-division-multiple-access (WCDMA); spectral regrowth; linearity
I. INTRODUCTION
One of the important applications of RF microelec-
tromechanical system (RF MEMS) devices is in
reconfigurable networks such as tunable filters and
impedance tuners [1–3], and the advantage of RF
MEMS devices is their excellent linearity [4]. From a
system-level point of view, linearity is a major issue
in wireless communications, especially for systems
which employ orthogonal types of modulation techni-
ques such as wideband code-division-multiple-access
(WCDMA) [5, 6].
The presence of two tones at the input of a non-
linear circuit generates third-order intermodulation
tones, and as a consequence, a WCDMA signal
results in spectral regrowth at the output port and a
nondesirable adjacent channel power leakage phe-
nomena (Fig. 1). The reason is that the abrupt phase
transitions in the WCDMA signal appear as ampli-
tude modulation in practice [7, 8], and therefore, the
WCDMA signal is very sensitive to circuit nonlinear-
ities in the RF front-end [5, 6]. It is therefore critical
to obtain RF reconfigurable circuits as linear as possi-
ble to avoid the adjacent channel interference in
WCDMA transceivers.
In this paper, two different multifunctional circuits
centered at 1.95 GHz, a tunable filter, and a tunable
matching network are designed using RF MEMS,
barium strontium titanate (BST), and gallium
arsenide (GaAs) varactors for a WCDMA front-end
centered at 1.95 GHz to investigate and compare their
response to a WCDMA input signal. Also, to improve
the linearity of the BST tunable networks, BST array-
Correspondence to: K. Entesari; e-mail: kentesar@ece.tamu.eduDOI 10.1002/mmce.20275Published online in Wiley InterScience (www.interscience.
wiley.com).
VVC 2007 Wiley Periodicals, Inc.
1
based varactors are employed instead of a single var-
actor and their response is compared with other tuna-
ble networks.
II. TUNABLE FILTERS IN WCDMATRANSCEIVERS
A. Tunable Filter Topology
Figure 2 shows the schematic of a two-pole band-
pass tunable filter suitable for wideband tuning at RF
frequencies [3, 9]. The filter is designed as a 3% Che-
byshev filter, with a center frequency of 1.95 GHz
(Table I). The inductors have a resistance of 0.15 Oresulting in a Q ¼ 200 at 1.95 GHz and a filter inser-
tion loss of 1.5 dB in the passband (the capacitors are
assumed lossless). By changing CR from 1.7 to 3.1
pF and keeping CM fixed, one can achieve a tuning
range of 1.7–2.2 GHz with a resonator match better
Figure 1. The effect of circuit nonlinearity on the spectral regrowth and adjacent channel inter-
ference of a WCDMA signal.
Figure 2. Schematic of a two-pole bandpass tunable fil-
ter at 1.95 GHz (CM: Matching capacitor, CR, LR: Resona-
tor capacitor and inductor).
TABLE I. Tunable Filter Element Values at 1.95
GHz for a 3% Chebyshev Design
Element Values
CR (pF) 2.3
CM (pF) 0.42
LR (nH) 2.48
R (O) 0.15
k 0.04
Figure 3. The simulated (a) insertion loss and (b) return
loss of the tunable filter covering the 1.7–2.2 GHz tuning
range (Qu ¼ 200 at 1.95 GHz, IL ¼ 1.0–1.8 dB). Simula-
tion is done using ADS [10].
TABLE II. Tunable Filter Characteristics for
Different Center Frequencies (Qu ¼ 200 at 1.95 GHz)
Frequency
(GHz) CR (pF) BW (%) IL (dB) BW (MHz)
1.7 3.1 3.2 1.8 54
1.8 2.7 3.3 1.7 60
1.95 2.3 3.0 1.5 58
2.1 2.0 3.4 1.2 71
2.2 1.7 3.5 1.0 77
2 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
than �13 dB (Fig. 3). Table II presents the band-
width, insertion loss, and the CR capacitor values for
the tunable filter at different frequencies. It can be
seen that the filter is slightly undercoupled at 1.7
GHz and slightly overcoupled at 2.2 GHz. A near-
perfect tunable filter response can be achieved if CM
is changed between 0.4 and 0.44 pF (not done for
simplicity).
To implement the tunable filter, CR is substituted
by RF MEMS switched capacitors, BST, and GaAs
varactors. Figures 4a and 4b show the side and top
views of the RF MEMS shunt capacitive switch,
respectively, and Figure 4c presents the electrical
model of the switch [1]. Table III provides the infor-
mation regarding physical dimensions and electrome-
chanical parameters of the switch.
The DC biasing electrode and the RF signal path
are identical in this device and result in the worst case
IIP3 compared to other types of capacitive switches. If
the DC biasing electrode is separated from the RF sig-
nal path, the RF MEMS switched capacitor shows bet-
ter linearity and higher IIP3 compared to the standard
design, but this is not considered here [1].
Figure 5a shows the C-V curve for a RF MEMS
capacitive switch [1]. The switch remains in the upstate
position for a bias voltage between �26 and 26 V with a
capacitance of 110 fF (0 V) � 148 fF (24 V). Above 26
V, the capacitive switch actuates to the downstate posi-
tion and results in a capacitance of 3.5 pF (capacitance
ratio of 30). The CAD-based model of a nonlinear
MEMS capacitor is shown in Figure 6. This model is
composed of three blocks: (A) electrostatic force gener-
ation, (B) the mechanical bridge, and (C) the variable-
gap parallel plate capacitor. This model is used to create
a behavioral model in ADS [4, 10].
Figure 5b shows the measured C-V curve for a
typical ferroelectric varactor [11]. The ferroelectric
varactor has a symmetric C-V curve and acts as a
variable capacitor for both positive and negative vol-
tages across the varactor. It provides a capacitance ra-
tio of 3.4 and is biased at 6 V (Cvar ¼ 1.8 pF) to
achieve maximum capacitance swing. The CAD-
based model of a nonlinear BST varactor is based on
fitting the mathematical Eq. (1) to the measured BST
C-V curve. This expression is modeled in ADS for
simulation purposes as
Figure 4. (a) Side view, (b) top view, and (c) the electrical model of a shunt capacitive RF
MEMS switch.
TABLE III. Physical Dimensions and
Electromechanical Parameters of the RF MEMS
Capacitive Switch
Physical dimensions
Bridge length, L (lm) 280
Bridge width, W (lm) 130
Bridge thickness, t (lm) 0.8
Air gap, g (lm) 2.0
Electrode width, w (lm) 160
Dielectric (er)–Roughness ratio (e)a 7.0–0.6
Dielectric thickness (lm) 0.2
Electromechanical parameters
Spring constant, k (N/m) 52
Pulldown voltage, Vp (V) 26
Mechanical resonant freq., f0m (kHz) 76
Mechanical quality factor, Qm 1.0
CMEMS, up (pF)/CMEMS, down (pF) 0.11/3.5
Switch inductance, LS (pH) 10
Switch resistance, RS (O)F 0.6
a Roughness ratio: Percentage of contact area between bridge
layer and dielectric layer in the downstate position. This is
between 0.5 and 0.7, and depends on the roughness of the dielec-
tric layer.
System-Level Responses in Complex Modulation Systems 3
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
CðVÞ ¼ aV= bþ V=cj jmð Þð Þ þ d ð1Þ
where a ¼ 2.3 3 10–12, b ¼ 0.9, c ¼ 8.5, m ¼ 1.14, d¼ 1.2 3 10–12 are constants found from mathematical
curve fitting.
Figure 5c shows the measured C-V curve for a
typical reversed-biased GaAs varactor [12]. The
GaAs varactor provides a capacitor ratio of 4 and is
biased at 6 V (Cvar ¼ 1.8 pF) to achieve maximum
voltage swing and linearity. The CAD-based model
of a nonlinear GaAs varactor is implemented by
employing a reverse-biased P-N semiconductor junc-
tion model in ADS and is fitted to the measured
GaAs C-V curve using
CðVÞ ¼ Cjo 1 þ V=Vj
� �� ��M ð2Þ
Figure 5. Capacitance vs. biasing voltage for a (a) RF MEMS (b) BST, and (c) GaAs varactors.
Figure 6. A CAD-based nonlinear model for the RF MEMS capacitive switch [4].
4 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
where Cjo ¼ 4.0 pF is the zero-bias junction capacitance,
Vj ¼ 5 V is the junction potential and M¼ 1 is the grad-
ing coefficient. Also, a comparison among the switching
speed of three different varactors shows that the switch-
ing speed for the existing RF MEMS switched capaci-
tors, BST, and GaAs varactors is in the order of 3–10 ls
[1], 1 ns [11], and 0.1 ns [12], respectively.
The tunable filters are now implemented using the
RF MEMS, BST, and GaAs devices (Fig. 7). In the
case of the RF MEMS design, the varactor CR is
implemented as a 3-bit switched capacitor, where
each RF MEMS switched capacitor is a series combi-
nation of the RF MEMS switch and a fixed capacitor
[3]. Table IV presents the different capacitance val-
ues and center frequencies vs. bit combinations. The
3-bit tuning approach is a practical solution in RF
MEMS, since it is very hard to fabricate a reliable
high-capacitance ratio RF MEMS analog varactor
[1]. Figures 7b and 7c show the tunable filter using
GaAs and BST varactors, respectively. Both GaAs
and BST varactors are biased at 6 V to provide the
filter response at the middle of tuning band (CR ¼ 2.3
pF and f0 ¼ 1.95 GHz). Table IV also shows the
value of BST and GaAs varactors and their biasing
voltages for different filter center frequencies. To
investigate the filter response to the WCDMA input
signal, it is assumed that all three filter center fre-
quencies are at 1.95 GHz.
Figure 7. Tunable filters implemented using (a) RF MEMS, (b) BST, and (c) GaAs devices. The
RF chokes (RFC) and bias resistors assumed to be ideal have no effect on the filter transfer function.
TABLE IV. MEMS Switch Combinations, BST and GaAs Varactor Values, and Biasing
Voltages for Different Filter Center Frequencies
f0 (GHz)
MEMS Design BST Design GaAs Design
S1-S2-S3 CR (pF) CVar (pF) CR (pF) VBias (V) CVar (pF) CR (pF) VBias (V)
1.7 ON-ON-ON 3.1 3.0 3.1 2.0 3.0 3.1 1.8
1.8 OFF-ON-ON 2.7 2.3 2.7 4.0 2.3 2.7 3.8
1.95 OFF-OFF-ON 2.3 1.8 2.3 6.0 1.8 2.3 6.0
2.1 ON-OFF-OFF 2.0 1.4 2.0 9.0 1.4 2.0 9.8
2.2 OFF-OFF-OFF 1.7 1.2 1.7 13.0 1.2 1.7 11.6
System-Level Responses in Complex Modulation Systems 5
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
Notice that for the BST and GaAs design, two
devices are used for CR to achieve reasonable values
for the tunable devices. Since the two devices are in
parallel, the resulting intermodulation products are
the same as a large device with twice the capacitance
value. A series combination of the BST (or GaAs)
varactor with a fixed capacitor is also used in the tun-
able filter topology to result in smaller voltage swing
across the BST (or GaAs) varactor and keep the de-
vice in the linear region as much as possible. This
will be discussed in Section B.
TABLE V. Voltage Swing Across a Single Varactor
Inside the Tunable Filter at node ‘‘a’’ for Different
Input Powers (f0 ¼ 1.95 GHz)
Pin (dBm) RF MEMS BST GaAs
10 3.2 2.4 2.4
20 10.0 �0.4 to 15a �0.4 to 15a
30 31.0 �17 to 36a �0.7 to 19b
a Distorted signal across Va.b GaAs design cannot handle >22 dBm. Voltage swing turns
on diodes completely.
Figure 8. Va(t) across the RF MEMS switch and the BST
and GaAs varactors for Pin ¼ 30 dBm at f0 ¼ 1.95 GHz.
Figure 9. Large-signal S21 for RF MEMS, BST, and GaAs tunable filter for different input
powers.
Figure 10. A tunable filter with (a) 3 3 1 and (b) 3 3 3 BST array varactors.
6 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
B. Simulation Results
Table V shows the voltage swing across the RF
MEMS, BST, and GaAs varactors inside the tunable
filter (Va in Fig. 7) for the 1.95 GHz single-tone exci-
tation, using harmonic balance simulation in ADS.
For Pin ¼ 30 dBm, Va ¼ 31 V peak (Vb ¼ 28.6 V due
to losses in the resonator) for the RF MEMS switch.
In this case, Va(r.m.s) ¼ 22 V, which is still lower than
the pulldown voltage of the switch (Vpulldown ¼ 26
V), and therefore the switch remains in the upstate
position, and self-biasing does not occur (as shown in
[1], If V(r.m.s) > Vpulldown, then the RF voltage can
pull down the switch even with no DC bias applied).
The voltage across the BST varactor is distorted for
Pin ¼ 30 dBm, since the signal experiences a huge
nonlinearity due to large capacitance variation of the
BST varactor.
For Pin ¼ 30 dBm, the voltage across the GaAs
varactor at nodes (a) and (b) is compressed and this is
due to the high voltage swing across the varactor
which pushes the varactor to the ohmic region
(Fig. 8). The large signal S-parameters of the filter
are simulated in ADS for different input power levels
(single tone excitation). The RF MEMS tunable filter
response does not change at all up to Pin ¼ 28 dBm
(Fig. 9). For Pin ¼ 30 dBm, the RF MEMS filter
response is slightly distorted due to the self-biasing
effect of the upstate MEMS bridges (Va(r.m.s) ¼ 22 V)
and the MEMS bridges are starting to pull down. For
the BST and GaAs tunable filter, the filter response is
distorted even at Pin ¼ 20 dBm because of the non-
linear behavior of the BST and GaAs varactors, andFigure 12. Large-signal S21 for BST (1 3 1), (3 3 1),
and (3 3 3) tunable filters for different input powers.
Figure 13. Simulated WCDMA spectrum at the output
of tunable filters for different input powers.
Figure 11. Simulated fundamental and IM3 output
power vs. input power for different BST topologies in the
2-pole tunable filter.
System-Level Responses in Complex Modulation Systems 7
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
both designs cannot handle 30 dBm of input power
(Fig. 9).
To improve the power handling and linearity of
the BST tunable filter, each BST varactor is substi-
tuted by a 3 3 1 and a 3 3 3 array of similar BST
varactors (Fig. 10). In the 3 3 1 case, each BST de-
vice is three times larger than the nominal value of
Figure 5, while in the 3 3 3 case, the same device in
Figure 5 is used. This topology divides the RF volt-
age across each BST device by a factor of 3 while
maintaining the same capacitance CR.
A two-tone intermodulation test is done in ADS
using two sinusoidal signals at f1 ¼ 1.95 GHz and f2¼ f1 þ Df (Df ¼ 1 MHz), and the simulated funda-
mental and IM3 powers for the 1 3 1, 3 3 1, and 3
3 3 BST-based tunable filters are shown in Figure 11.
As expected, the IIP3 level increases from 23 dBm
for 1 3 1 BST topology to 33 dBm for 3 3 1 and 3
3 3 BST topology. Figure 12 presents the large-sig-
nal S-parameters for 1 3 1, 3 3 1, and 3 3 3 BST-
based tunable filter at Pin ¼ 20 and 30 dBm. The
filter response for 3 3 1 and 3 3 3 BST topologies
has no distortion at Pin ¼ 20 dBm, and the power
handling of the 3 3 1 and 3 3 3 BST topology is
improved to 26 dBm without any shift in the large-
signal S-parameters.
The single-tone source is now substituted with a
WCDMA source at 1.95 GHz. The circuit is simu-
lated in ADS using envelope-simulation technique,
and the WCDMA spectrum is extracted at the output
of the tunable filter (Fig. 13). For Pin ¼ 10 dBm and
Va ¼ 2.4 V, the GaAs and 1 3 1 BST filters are
pushed to their nonlinear region while the 3 3 1 and
3 3 3 BST topologies and RF MEMS filters are still
in their linear region. The WCDMA output spectrum
for GaAs and 1 3 1 BST tunable filters shows a spec-
tral regrowth of more than 40 dB compared with their
MEMS counterpart at the adjacent channel. For Pin ¼20 dBm, the BST-tunable filters show 50 dB (1 3 1)
and 20 dB (3 3 1, 3 3 3) of spectral regrowth com-
pared to the input signal. For Pin ¼ 30 dBm, the RF
MEMS filter remains in the linear region and still fol-
lows the input signal, while both GaAs and BST fil-
ters show extremely nonlinear behavior. The output
spectrum of the GaAs filter is not shown in Figure
13c, because the GaAs filter response is totally dis-
torted at this power level and does not show a filter
response any more.
Figure 14 presents the simulated IIP3 for different
filter topologies at different filter center frequencies.
The GaAs and BST filters (1 3 1, 3 3 1, 3 3 3)
show similar behaviors: At f0 ¼ 1.7 GHz, the GaAs
and BST varactors are biased at 1.8 V and 2 V,
respectively, and because of the steep C-V curves at
these bias voltages, this results in the worst IIP3 com-
pared with other center frequencies. Also, the IIP3 is
improved by 10 dBm for (3 3 1, 3 3 3) BST topolo-
gies, compared with the 1 3 1 BST topology for all
center frequencies. The IIP3 for the RF MEMS filter
is simulated for Df ¼ 100 kHz and Df ¼ 1 MHz at
different filter center frequencies, and it shows
20 dBm improvement at Df ¼ 1 MHz compared to Df¼ 100 kHz.
This is because the intermodulation component
follows the mechanical response of the bridge, and
the IM3 level drops by 40 dB/decade for Df > f0m
(f0m ¼ 76 kHz), which is in agreement with theory
[4]. At f0 ¼ 2.2 GHz, all the MEMS switches are in
the upstate position, and this state results in the worst
IM3 products and the worst IIP3 level. At f0 ¼ 1.7
GHz, all the switches are in the downstate position
and this results in the highest IIP3 compared to other
filter center frequencies. Note for Df > 1 MHz, the
IIP3 of the tunable filter is >55 dBm for all tuning
conditions (IIP3 > 75 dBm for Df > 10 MHz).
Figure 14. Simulated IIP3 at different filter center fre-
quencies for different filter topologies.
Figure 15. The schematic of a WCDMA transmitter
front-end including a linear power amplifier and a tunable
matching network.
8 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
Figure 16. ZLoad impedance loci at the output of the power amplifier for (a) CS ¼ CL ¼ 5.7
pF, (b) CS ¼ CL ¼ 4.0 pF, (c) CS ¼ CL ¼ 7.4 pF. L ¼ 1.15 nH for all cases. (Smith chart im-
pedance ¼ 4 O). ZAmp impedance loci at the load (Zout) for (d) CS ¼ CL ¼ 5.7 pF, (e) CS ¼ CL
¼ 4.0 pF, (f) CS ¼ CL ¼ 7.4 pF. L ¼ 1.15 nH for all cases (Smith chart impedance ¼ 50 O).
System-Level Responses in Complex Modulation Systems 9
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
III. IMPEDANCE TUNER IN WCDMATRANSCEIVERS
A. General Matching Network Topology
Figure 15 presents a power amplifier with a reconfig-
urable matching network for wireless applications.
The power amplifier is assumed ideal with 20 dB of
gain and an input impedance of 50 O. The output im-
pedance, ZAmp, and the antenna impedance, ZAnt, are
assumed 4 and 50 O, respectively. To match the 4 Oto the 50 O impedance, a CS-L-CL p-network is used
(CS ¼ CL ¼ 5.7 pF, L ¼ 1.15 nH). In practice, ZAmp
and ZAnt are variable impedances, and a tunable
matching network is necessary to match ZAmp and
ZAnt for different impedance values.
The impedance tuner is designed for two cases:
1. ZAmp ¼ 4 O and ZAnt varies between 15 6
j50 and 100 6 j50.
Figure 16a shows ZLoad for ZAnt ¼ 15 6
j50 to 100 6 j50, for CS ¼ CL ¼ 5.7 pF, and
L ¼ 1.15 nH. All the impedance loci inside
the G ¼ 10 dB circle are matched to 4 O with
Figure 17. (a) RF MEMS, (b) BST, and (c) GaAs tunable matching networks.
TABLE VI. MEMS Switch Combinations, BST and GaAs Varactor Values, and Biasing
Voltages for Different Matching Capacitors
MEMS Design BST Design GaAs Design
S1-S2 CS,L (pF) CVar (pF) CS,L (pF) VBias (V) CVar (pF) CS,L (pF) VBias (V)
ON-ON 7.4 3.0 7.4 0.5 3.0 7.4 0.6
OFF-ON 5.7 2.5 5.7 3.3 2.5 5.7 3.0
OFF-OFF 4.0 1.8 4.0 15.5 1.8 4.0 12.5
10 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
a return loss better than 10 dB, but many of
the impedance loci are outside this circle. By
changing the capacitors to CS, CL ¼ 4.0 pF
(Fig. 16b) and 7.4 pF (Fig. 16c), the matching
range is extended to cover virtually as much
as possible antenna impedances.
2. ZAnt ¼ 50 O, and ZAmp varies between 2 6 j8and 8 6 j8.
Figure 16d shows the power amplifier output
impedance locations after the matching net-
work (Zout) for ZAmp ¼ 2 6 j8 to 8 6 j8, and
CS ¼ CL ¼ 5.7 pF, L ¼ 1.15 nH. All the imped-
ance loci inside the G¼ 10 dB circle are
matched to 50 O with a return loss better than
10 dB, but many of the impedance loci are out-
side this circle. To extend the matching range,
the capacitors CL and CS are changed while the
series inductor is fixed. Figures 16e and 16f
show Zout for CS ¼ CL ¼ 4.0 and 7.4 pF,
respectively.
B. RF MEMS, BST, and GaAsImpedance Tuners
CS and CL are substituted with the three different
types of varactors presented in Figure 5. Figure 17a
shows the RF MEMS impedance tuner, and each
varactor (CS, CL) is implemented as a 2-bit switched
capacitor.
Table VI presents the different capacitance values
vs. bit combinations. Figures 17b and 17c present the
impedance tuner using GaAs and BST varactors,
respectively. By changing the biasing voltage of the
GaAs or BST devices, the capacitors CS and CL vary
between 4.0 and 7.4 pF. Table VI also shows the
value of BST and GaAs varactors and their biasing
voltages for CS ¼ CL ¼ 4.0 pF, CS ¼ CL ¼ 5.7 pF,
and CS ¼ CL ¼ 7.4 pF. To investigate the effect of
the impedance tuner nonlinearity on the WCDMA
signal, we consider the nominal case where CS ¼ CL
¼ 5.7 pF (ZAnt ¼ 50 O, ZAmp ¼ 4 O).
TABLE VII. Voltage Swing Across the Varactors at Nodes ‘‘a’’ and ‘‘b’’ in Figure 17
Pout (dBm)
RF MEMS BST GaAs
Va (V) Vb (V) Va (V) Vb (V) Va (V) Vb (V)
10 0.3 1 0.2 0.75 0.2 0.75
20 0.9 3.2 0.7 2.5 0.7 2.5
30 2.7 9.2 2.3 �3.3 to 12.8a 2.8 �0.8 to 9.8b
a Distorted signal across Vb.b Voltage swing turns on GaAs diodes completely.
Figure 18. Impedance tuners with (a) 3 3 1 and (b) 3 3 3 BST array varactors.
System-Level Responses in Complex Modulation Systems 11
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
C. Simulation Results
Table VII shows the simulated voltage swing across
the RF MEMS, BST, and GaAs varactors (using har-
monic balance simulation in ADS) inside the recon-
figurable network (Va and Vb in Fig. 17) for a 1.95-
GHz single-tone excitation for different output power
levels. Vb is larger than Va, because node ‘‘b’’ is a
higher impedance node. As a result, varactors con-
nected to node b are pushed to the nonlinear region
more strongly. For Pout ¼ 30 dBm, the voltage across
the GaAs varactor at node b is compressed due to the
high voltage swing across the varactor which pushes
the varactor to the ohmic region. The voltage across
the BST varactor at node b is also distorted due to the
nonlinearly of the BST varactor. For the RF MEMS
capacitive switch, Vb reaches to 9.2 V (Vb(r.m.s) ¼ 6.5
V), which is much lower than the pulldown voltage
of the switch (Vpulldown ¼ 26 V).
To improve the linearity of the BST impedance
tuner, each BST varactor is substituted by 3 3 1 and
3 3 3 arrays as was done previously (Fig. 18).
Table VIII presents the voltage swing across a single
BST device connected to node ‘‘a’’ or ‘‘b.’’ For Pout
¼ 30 dBm, the 3 3 1 and 3 3 3 topologies show no
distortion in Vb. The 3 3 1 and 3 3 3 topologies also
show a 10-dB improvement in IIP3 compared to the
1 3 1 topology (not shown).
The single-tone source is now substituted with a
WCDMA source at 1.95 GHz. The circuit is simu-
lated in ADS using envelope-simulation technique,
and the WCDMA spectrum is extracted at the output
of the impedance tuner (Fig. 19). For Pout ¼ 10 dBm,
all three tuners are in the linear region and the spec-
tral regrowth at the adjacent channel is negligible
(not shown). For Pout ¼ 20 dBm, the GaAs and 1 3
1 BST array tuners are pushed to their nonlinear
region while the 3 3 1, 3 3 3 BST topology and RF
MEMS tuner stay in the linear region. The output
spectrum for GaAs and 1 3 1 BST tuners shows a
spectral regrowth of 25 and 20 dB compared to the
input signal, respectively. For Pout ¼ 30 dBm, the
GaAs and 1 3 1 BST tuners show large spectral
regrowth (50–25 dB) while the RF MEMS tuner still
stays in the linear region with no spectral regrowth.
CONCLUSION
This paper presented the spectral regrowth for RF
MEMS, BST, and GaAs tunable filters and reconfig-
urable matching networks with WCDMA excitation.
The RF MEMS tunable filter and impedance tuner
have the best linearity and lowest spectral regrowth
compared with their GaAs and BST counterparts.
The (3 3 3, 3 3 1) BST topologies improve the line-
arity by 10 dB, but still show significant spectral
regrowth at 20 dBm (tunable filter) and 30 dBm (tun-
able matching network). As a result, RF MEMS front
ends are excellent candidates for complex modulation
reconfigurable systems and can handle 1 W of power
in narrowband tunable filters.
REFERENCES
1. G.M. Rebeiz, RF MEMS theory, design, and technol-
ogy, Wiley, New York, 2003.
TABLE VIII. Voltage Swing Across a Single Varactor
Inside BST Arrays at Nodes ‘‘a’’ and ‘‘b’’
Pout
(dBm)
BST (1 3 1) BST (3 3 3) BST (3 3 1)
Va (V) Vb (V) Va (V) Vb (V) Va (V) Vb (V)
20 0.7 2.5 0.2 0.8 0.2 0.8
30 2.3 �3.3 to
12.8
0.7 2.6 0.7 2.6
Figure 19. Simulated WCDMA spectrum at the output
of the RF MEMS, GaAs, and BST impedance tuners.
12 Entesari and Rebeiz
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
2. T. Vaha-Heikkila, J. Varis, J. Touvinen, and G.M.
Rebeiz, A 20–50 GHz RF MEMS single-stub imped-
ance tuner, IEEE Microwave Wireless Compon Lett
15 (2005), 205–207.
3. K. Entesari and G.M. Rebeiz, A differential 4-bit 6.5-
10 GHz RF MEMS tunable filter, IEEE Trans Micro-
wave Theory Tech 53 (2005), 1103–1110.
4. L. Dussopt and G.M. Rebeiz, Intermodulation distor-
tion and power handling in RF MEMS switches, var-
actors and tunable filters, IEEE Trans Microwave
Theory Tech 51 (2003), 1247–1256.
5. B. Razavi, RF microelectronics, Prentice Hall, Upper
Saddle River, NJ, 1998.
6. T.H. Lee, The design of CMOS radio-frequency inte-
grated circuits, Cambridge University Press, Cam-
bridge, 2001.
7. D.H. Morais and K. Feher, The effects of filtering
and limiting on the performance of QPSK, OQPSK,
and MSK signals, IEEE Trans Commun 26 (1980),
1999–2009.
8. J.F. Sevic and J. Staudinger, Simulation of adjacent
channel power for digital wireless communication
systems, Microwave J (1996), 66–80.
9. K. Entesari, T. Vaha-Heikkila, and G.M. Rebeiz, Min-
iaturized differential filters for C and Ku-band appli-
cations, 33rd European Microwave Conference, Mu-
nich, Germany, 2003, pp. 227–230.
10. ADS, Agilent Technology, Palo Alto, CA, USA.
11. A. Tombak, J. Maria, F.T. Ayguavives, Z.Z. Jin, G.
Stauf, A.I. Kingo, and A. Mortazawi, Voltage-con-
trolled RF filter employing thin-film Barium-Stron-
tium-Titanate tunable capacitors, IEEE Trans Micro-
wave Theory Tech 51 (2005), 462–467.
12. A.R. Brown and G.M. Rebeiz, A varactor tuned RF
filter, IEEE Trans Microwave Theory Tech 48 (2000),
1157–1160.
BIOGRAPHIES
Kamran Entesari received the B.S.
degree in electrical engineering from
Sharif University of Technology, Tehran,
Iran, in 1995, the M.S. degree in electrical
engineering from Tehran Polytechnic Uni-
versity, Tehran, Iran, in 1999, and the
Ph.D. degree from the University of
Michigan, Ann Arbor, in 2005. In 2006,
he joined the Department of Electrical
and Computer Engineering at Texas A&M University where he is
currently an assistant professor. His research interests include
design of radio frequency/microwave/millimeter-wave integrated
circuits and systems, microelectromechanical systems (MEMS)
for microwave/millimeter-wave applications, related front-end
analog electronic circuits and antennas, microwave filters and pas-
sive components, and active and passive sensors. He is a member
of IEEE Microwave Theory and Techniques Society.
Gabriel M. Rebeiz is a professor of elec-
trical engineering at the University of Cal-
ifornia, San Diego. He received his Ph.D.
(1988) from the California Institute of
Technology. He has contributed to planar
mm-wave and THz antennas and imaging
arrays from 1988 to 1998, and to the de-
velopment of RF MEMS from 1996 to
present. He is the author of the book, RF
MEMS: Theory, Design and Technology, Wiley (2003). Prof.
Rebeiz’ group recently developed the fastest mm-wave SiGe
switch to date (70 ps), and 6–18 GHz and 30–50 GHz 8-element
phased array receivers and transmitters on a single chip, making
them the most complex RFICs ever built at this frequency range.
Prof. Rebeiz is an IEEE Fellow, an NSF Presidential Young In-
vestigator, an URSI Koga Gold Medal Recipient, an IEEE MTT
Distinguished Young Engineer (2003), and the 1998 Eta-Kappa-
Nu Professor of the Year and the 1998 Amoco Teaching Award
given to be the best undergraduate teacher at the University of
Michigan.
System-Level Responses in Complex Modulation Systems 13
International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce
top related