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Nest Watch:

Proximity Sensor Antenna

Catherine Boosales

Spring 2001

Faculty Supervisor: Dr. Thomas Weller


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Abstract

The Nest Watch Wireless Sensor group project set out to design and build a sensor to

monitor activity in a birds nest. This paper gives a brief description of the overall

project and then directs focus to the 2.45 GHz patch antenna that was part of the

proximity sensor of the system. Basic theory centered on the design improvements of an

aperture couple-fed microstrip patch antenna is presented followed by an outline of the

design steps. Lastly, data comparisons are made and conclusions drawn.

Introduction

With an overall goal of producing a wireless sensor that will monitor activity in a

birds nest and transmit the data to a remote location for collection, two teams of students

from two different universities collaborated on the project. Each student at the University

of South Florida (USF) had two roles to play in terms of the overall project. One of these

roles involved a system level responsibility of some type that would require substantial

interaction with other team members, some of which were from the other university

involved, Tennessee Technological University (TTU). The second role involved the

analysis and design of one part of the overall sensor. The production of the antenna for

the proximity sensor and the integration and calibration of the same sensor will be

presented in this report.

Before further details about the antenna or the proximity sensor are discussed,

some description of the overall project, The Nest Watch Wireless Sensor, is necessary.

USF was responsible for the production of three main parts of the sensor, the sensor card

itself, the transmitter and the receiver. The sensor card is simply a proximity sensor with

a temperature sensor and a light intensity sensor mounted to it. The proximity sensor

works at a frequency of 2.45 GHz and will be described in more detail in the next section.

The data from the sensor card is sent via a 915 MHz signal from the transmitter card,

which would be attached to the sensor card, to the receiver. Once the signal is received,

the information is processed and then displayed on a web page designed by TTU. The


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student team members at TTU also designed the signal processing parts of the transmitter

and receiver as well as the software that is used for the data collection. Students from

both universities were responsible for inter-university communication where necessary.

As an illustration of team roles, figure 1 demonstrates how the different parts fit together

to form the collective project. The boxes in green represent USF and those in purple are

for TTU.

Figure 1: University Team Assignments

On a slightly smaller scale, the baseline proximity sensor was assembled and built

by a team of 4 students and then tested to obtain calibration data. This first sensor card

consisted of a surface mount voltage controlled oscillator to generate the 2.45 GHz

signal, a quadrature hybrid coupler to equally split the signal into two (a reference signal

and a target signal), a surface mount mixer to serve as a phase detector and a patch

antenna to send and receive the target signal. In addition, an amplifier was also produced

to be used in the event that one of the signals wasnt strong enough for detection thereby

requiring amplification. This card was the safe approach first design produced to

ensure that a working sensor would be produced within the time frame of a semester.

Further analysis and design of many of the individual components of the system were to

be completed by the students as their individual projects.

Proximity

Baseband

DC ower

Temperature

Light Intensity

RF TX Antenna

USF USF USF

USF

TTU

TTU

TTU

Baseband

DC Power

Signal

ProcessingRF RXAntenna Web

Posting

TTU TTU TTUUSF USF


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The portion of the project presented in this paper contained two initial goals and

the addition of a third, as the semester progressed. The first and foremost goal was that

of producing a suitable antenna for the proximity sensor. This antenna was to transmit

and receive at 2.45 GHz with a return loss of better than 10 dB. The safe-approach

version of this antenna was to be probe-fed from the edge of the coupler on the sensor

card through the ground plane of the antenna. Once the baseline antenna was produced,

additional modifications to improve performance were to take place. The specifications

of the second antenna were to be determined during the semester via feedback from

various sources particularly including the calibration team with the overall goal of

improving sensor performance. One such improvement suggested at the onset, was an

antenna that was feed via a magnetically coupled feed line through a small aperture in the

ground plane of the sensor card as pictured in figure 2.

Figure 2: Example drawing

of an aperture-coupled feed

antenna.

Once the initial probe-fed design was produced and tested, the motivation for

producing this type of couple-fed antenna changed from that of improving polarization

purity and return loss (by improving input impedance match) to the necessary goal of

bringing the antenna closer to the sensor card for logistical reasons which will be

explained later. The production approach for the antennas began with the performance of

lab exercise 12 from the Wireless Circuits and Systems Design Course1 with

modifications to feed characteristics and resonant frequency. The second design began

with calculations by hand and then continued or altered in ADS-Linecalc and PCAAD.

These calculations were then utilized to create a layout in ADS to allow for a Momentum

Simulation. Lastly, the layouts were submitted for milling and then tested, and tuned as

necessary, after production.


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The second initial goal was to produce valid calibration data for the proximity

sensor so that analysis could be performed. This data was to help determine how to

distinguish between different types of activities in the nest. This calibration was to be

performed initially on the coaxial part sensor that was set up and then on each successive

sensor card to be produced. Calibration involved the testing and recording of output

voltages for a range of set-ups selected to simulate bird nest activity such as the presence

of eggs or that of a bird or two. This information was then relayed to TTU for reference

when designing the baseband part of the transmitting system.

As the second stage antenna design depends very highly on the placement of the

feed line on the circuit, it was necessary to become considerably more involved with the

sensor integration team than originally intended. In order to determine if the antenna

design to be produced was going to function as planned, it was necessary to produce the

new sensor card as well. As the semester progressed, additional time was spent on the

production of the second and third stage sensor cards. The successful fabrication of these

two cards quickly developed into the third goal of this particular project.

Theory

The Nest Watch Wireless Sensor utilizes three different antennas to operate. Two

of these antennas are designed to operate at a frequency of 915 MHz for transmission and

reception of data. The third antenna is part of the proximity sensor and as such will

simultaneously transmit and receive a signal at a frequency of 2.45 GHz. The VCO on

the sensor card generates a signal that is split into two equal signals. Both of the new

signals are of the same frequency and phase. One of the signals gets sent directly to the

mixer as the reference signal. The second signal is transmitted from the antenna towards

the target (in this case, a resident of the nest). The signal will bounce off the target (if

there is one present) and return to the antenna still at the same frequency, but with a

different phase. This new signal is received and then sent back through the coupler to the

other input of the mixer. The mixer will combine the two signals (the reference signal

and the target signal) to produce a DC voltage, as is always the case when two signals of

the same frequency are mixed, that will vary based on the phase difference. Since the


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sensor card will be placed below the birds nest, a microstrip patch antenna was chosen as

the design of the antenna because it will radiate a relatively broad beam that is normal to

the surface of the patch. This will direct the signal in the correct direction for occupant

detection. In addition, the patch antenna will resonate effectively at /2, which at 2.45

GHz would be of a fairly small size and therefore as unobtrusive to the bird as possible.

Due to simplicity and the need for a baseline antenna to operate with, a probe-fed patch

was designed. The equivalent circuit for such a patch is depicted in Figure 3, where the

parallel RLC represents the patch and the series inductor stands for the inductance in the

probe feed-line2.

Figure 3: Equivalent Circuit for Probe-fed Patch Antenna

The probe-fed (or line-fed) patch antenna only has one design parameter that can

be varied to optimize input impedance. The location of the probe determines the

magnitude of the real part of the input impedance and as such, cannot be located in the

center of the patch (where the impedance is /4 away from a virtual open and is therefore

seen as a virtual short). This would be where Zin is zero and therefore not matched to the

line. In addition, the probe cannot be at the edge because in a /2 patch, the both edges

are virtual opens and the real impedance is at a maximum. An additional advantage to a

probe-fed design is the lack of reliance on the balance of the circuit to complete the

design. As the probe has to extend off of the back of the patch, the location of the probe

on the sensor card is unimportant thereby eliminating many integration issues.

Disadvantages to the probe feed, while numerous in some situations, are fairly

limited in this application. The first and most obvious is a mechanical difficulty with the

manner in which the antenna is attached to the sensor card. In the probe feed

configuration, the patch extends from the back of the sensor card by about 3 centimeters

as can be seen in figure 4. While this doesnt seem like a very large distance, it would


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pose a problem should this initial design need to be placed unobtrusively underneath a

birds nest. The nest would literally have to be balanced on the antenna. By utilizing an

aperture couple fed antenna where the feed line is on the face of the sensor card with the

rest of the circuit and the antenna is flat against the back of the card with a ground plane

containing a small hole (or aperture) between them, the mechanical difficulties of the

probe-fed antenna are alleviated.

Figure 4: Initial Probe-fed antenna pictured

as connected to the back of the sensor card.

A second disadvantage of the probe-fed antenna has to do with polarization.

While a probe feed will excite the dominant mode of the antenna element, the asymmetry

of the feed in relation to the patch (as necessary for impedance matching) will generate

some higher-order modes. These additional modes of excitation will produce cross-

polarized radiation, degrading polarization purity.

Lastly, the probe-fed patch antenna traditionally has a very narrow band width.

This is due primarily to the location of the feed network and the resonating structure on

the same substrate. In order to improve bandwidth, it is possible to use a thicker substrate

with a lower dielectric. Unfortunately, with the feed set up as a probe (the same would

be true of an inset microstrip feed), increasing the thickness or dielectric of the substrate

would also be increasing the thickness of the feed substrate thereby increasing spurious

feed radiation in the circuit. This will affect overall antenna performance by increasing

loss of power and possibly affecting radiation efficiency, which is why in circuits where

radiation losses are detrimental, thin, high dielectric substrates are advantageous (such as

GaAs). Since the supplies available for this design did not include substrates of varying


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thickness and dielectric, this disadvantage could not be accounted for and was

subsequently uncorrectable in the final design.

The second iteration of antenna design involved the alteration of the feed

mechanism through which the antenna receives its power. This mechanism, an aperture-

coupled microstrip antenna, was first proposed by D.M. Pozar in the mid-eighties3. As

mentioned in the introduction, this design consists of a patch antenna on one layer of

substrate with the feed line (or network) on a second layer of substrate with a mutual

ground plane between them. The ground plane contains a hole or aperture that is

electrically small as compared with the wavelength of operation. This hole allows the

coupling of the equivalent magnetic current of the feed line through the aperture to

stimulate the dominant H-field of the patch4. If the dominant mode of the patch has an

electric field of zero and a maximum magnetic field centered directly below the patch,

then placement of the slot in this location would give a maximum coupling factor5. Pozar

demonstrated this in his initial proposal of the design. An equivalent circuit

representation of this mechanism is pictured in figure 5 where the series RLC represents

the patch, the parallel inductor is the aperture and the shunt capacitor stands for the feed

line6.

Figure 5: Equivalent Circuit for Aperture-coupled Patch Antenna

There are many advantages to an antenna of this feed type, not all of which were

necessary for this particular application. As mentioned earlier, this feed arrangement

requires the antenna and the circuit to be on separate substrates. This makes it possible to

utilize a thick, low dielectric constant substrate to maximize radiation and bandwidth onthe antenna side and a thin, high dielectric constant substrate (such as GaAs) to optimize

the circuit side. In addition, this configuration reduces the competition for real estate on

the circuit board between circuit devices and the antenna. Since the ground plane

effectively shields the two sides from each other, there is reduced spurious radiation

affecting the antenna from the feed lines or active devices of the circuit.


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Other advantages include a reduction in the size of the back lobes of the radiation

pattern because the aperture is smaller than the resonant size and a reduction in higher

order modes due to the symmetrical excitation resulting from the aperture location in the

center of the patch. This design also gives four degrees of freedom in design

optimization: the slot size, the feed substrate parameters, the feed line width and the

location of the slot. Impedance matching is achieved through the adjustment of the

length of the slot and the width of the feed line.

As an example of how adjusting these parameters affects antenna performance, it

should be noted that increasing the stub length from 10% of the patch length to 50% of

the patch length will rotate the trace around the smith chart to allow for the tuning of the

reactive part of the impedance as pictured in figure 6a. Adjusting the aperture length

from 22.5% of the patch length to 35% of the patch length allows for the tuning of the

resistive part of the impedance and will increase the coupling factor as illustrated in

figure 6b7.


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Figure 6a: Input impedance asa function of stub length.

Percentages are of patch length.

10%

50%

40%

20%


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These charts show how the impedance changes as a function of changing one (of

four possible) parameter. Lap is the length of the aperture and Ls is the length of the slot.

As in the case of a probe-fed patch as well, the resonant frequency is primarily

determined by the length of the patch. In an aperture coupled microstrip patch antenna,

the resonant frequency decreases and the resonant resistance increases with increasing

slot length. This relationship could come in very handy if patch size was limited but slot

length was not (within reason) and resonant resistance was unimportant. This would

allow you to increase the length of the slot and decrease the length of the patch and still

maintain the same resonant frequency while only increasing resonant resistance.

Figure 6b: Input impedance asa function of aperture length.

Percentages are of patch length.

30%

25%

35%

22.5%


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Design Process

As previously mentioned, the first goal of this project was to produce a patch

antenna to be utilized as part of the safe approach sensor that was being produced. To

this end, calculations to determine the initial patch dimensions were made as shown in

figure 7. The only substrate available for this design was FR-4 which has a dielectric

constant of 4.3 and a dielectric loss tangent of 0.022. Additionally, the thickness used for

calculations was 0.1574 cm.

rfd

C

= mmm 59059.0

3.410945.2

1080.3==

=

mm505.292= = length of patch width of patch = length x 1.25 = 23.604mm

Figure 7: Probe-fed Patch Initial Calculations

Once the dimensions were calculated by hand, all the information was entered into an

antenna design program called PCAAD. This allowed the determination of the optimal

probe-to-edge distance for the feed line. This dimension is the one design parameter that

can be altered to affect the antennas effective match to the 50 feed line of the probe

(and ultimately the sensor circuit, as well). The distance that was chosen based on the

best match was 1.108 cm from the edge. Calculations determined that this location

should give an impedance value between 50.5+j15 and 49.3+j7 for the frequency of

interest. A rendition of the patch layout that was submitted for milling can be seen in

figure 8 on the next page. This antenna was milled, soldered and tested, the results of

which will be discussed in the next section.


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.

The next step in the design process involved the acquisition of knowledge of the

Momentum features in Agilents Advanced Design System (ADS). By performing a

previously tested procedure involving the definition and simulation of a shunt stub

layout, it was possible to gain initial exposure to this program. In the process, new but

critical concepts such as how to define a substrate or edit a port quickly became clear.

These steps were explored further in the process of designing the second generation of

antennas. As a further investigation into these features, the probe-fed patch designed

above was successfully simulated and the results compared to actual data to be discussed

in the next section. Some alterations of the size of the patch had to be made to account

for the lack of precision in placement of the probe feed (it had to be measured and drilled

by hand). In order to adjust the input impedance to improve return loss, 0.5 mm were

manually (estimated) sliced off the length closest to the probe. This action was intended

not only to slightly increase the resonant frequency (i.e. Small patch) but also effectively

decreased the probe-to-edge distance thereby increasing the impedance from around 46

to 50. The results of this effort will also be discussed in the next section.

23.68 mm

29.6

0mm

11.84 mm 11.84 mm

11.0

8mm

Figure 8: Microstrip Patch

Antenna Version 1


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The second antenna design process began with the PCAAD program and the

dimensions of the previous patch. In this program, there is an option for designing a

rectangular aperture coupled patch. When selecting this option (located under the

microstrip heading), the user is prompted for the necessary parameters. These included

the antenna substrate dielectric constant thickness, the patch length and width, the feed

substrate dielectric and thickness (which in our case was the same as the antenna

substrate), the slot length and width, and the feed line width and stub length. A diagram

of these dimensions as defined by the PCAAD program is pictured in figure 9.

Once the parameters were keyed in correctly, PCAAD will estimate the design

frequency and use that as the default center frequency over which to analyze the input

impedance of the antenna (over a sweep range with this frequency at the center) 8. The

program will also compute and display the estimated bandwidth, the radiation efficiency,

the front-to-back ratio and the directivity. In this case, the only parameter that couldnt

be altered was that of the feed line width which was set at 50 for the circuit of the

Figure 9: AntennaDimensions for PCAAD

program

Patch length

Patch width

Stub length

Feed line width

Slot length

Slot width

A erture


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sensor card. If a different impedance was needed for the antenna, a matching network of

some type would have to be included in the sensor card layout. To avoid additional

integration difficulties, it was most convenient to match the antenna to 50 as was done

with the probe-fed patch. Once dimensions that appeared to give the correct response

were simulated in PCAAD, the same dimensions were then drawn in the layout program

of ADS. Initial attempts to match the ADS results to the PCAAD results proved much

more discouraging than helpful. To alleviate this frustration, PCAAD was virtually

abandoned for the remainder of the design process.

The initial patch design that was created for simulation consisted of an assortment

of arbitrarily chosen lengths (kind of a guess). This was done because it was initially

unclear as to how PCAAD had defined the dimensions that were working, and because

additional practical exposure to the substrate/layer definitions in ADS was necessary. In

order to define a substrate, ADS asks that all the necessary parameters are keyed in and

then the program will compute the substrate over a given frequency range. This newly

created definition can then be saved for future use. The more difficult part of this

substrate defining process was to assign the definitions to the correct layers so that ADS

knows how this design is assembled. As can be seen in figure 10, the substrate layers had

to be defined in terms of the layers on which the design was drawn.

In this design, the patch was on the 1st

cond layer, the feed line was on the

second cond layer (cond2) and the slot was on one of the pcvia layers. By telling ADS

AIR

FR4

FR4

AIR

Figure 10: Definitionof design layers (lines

are copper)

Red is antenna, black is ground plane and gold is feed line


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how each layer was related to both the substrates and the air, it was possible to

successfully simulate a multi-layered magnetically coupled design.

Since an impedance maximum will occur at wavelength multiples, an

impedance minimum and therefore a current maximum will occur wavelength from the

maximum point. This would indicate that if the maximum current in the feed line should

be located at the aperture opening, then the feed line should be terminated in an RF open

wavelength past the slot. In simulation, this didnt appear to work. The simulations

where the feed line ended wavelength past the slot produced resonant frequencies with

unacceptable return losses (less than 8 dB). After repeated attempts at correcting this

problem by altering the slot dimensions and the patch dimensions, it became clear that

regardless of what the theory dictates, in simulation, this arrangement wasnt producing

even moderately acceptable results. Furthermore, the original antenna attempt, which

was based on a misinterpretation of the PCAAD arrangement, produced the best results to

this point.

It was also discovered in this original design attempt that, contrary to the correct

information in PCAAD, results were more predictable when the long side of the slot was

perpendicular to the patch. It quickly became apparent in this process that a much

smaller patch needed to be utilized when using this method in order to keep the resonant

frequency in the right frequency range.

With all this in mind, the final attempts began to approach the arena of suitable

designs. Narrowing the slot produced better return loss but tended to increase resonant

frequency so the patch length had to be reduced to compensate. The last iteration

involved the scaling of the parts in relation to each other to maintain the same match, but

to increase the resonant frequency. This actual layout of this design can be pictured in

figure 10a while the current distribution simulation can be seen in figure 10b showing

maximum current in the slot where the coupling is occurring.


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After the simulations were producing acceptable results, the actual layout of the

antenna needed to be produced. The first challenge involved creating an antenna and

feed network without the sensor card so that testing could be accomplished on the

antenna alone. As may be expected, while the above design simulates easily, the desire

to test it in the lab illuminated some additional challenges. The feed network would be

milled on FR4 substrate as usual with the added difficulty of requiring an accurately

placed slot (or hole) in the ground plane on the reverse side. This proved to be somewhat

of a challenge to our current milling capabilities, but was accomplished in a timely

manner. The antenna itself had to be milled on single-sided FR4 so that no ground plane

would be present. As was discovered the hard way, the layout for this part of the design

needed to be a mirror image of the original layout as it would be flipped over and

attached to the back of the feed network card. In addition, there needed to be a way to

attach an edge-mount connector to the edge of the board. This required cutting away a

few millimeters of the antennas substrate so that the connector could make contact with

the ground plane.

Figure 11b: Aperture Coupled Patch

Antenna Current Distribution Simulation(green is higher current levels)

Figure 11a: Aperture Coupled Patch

Antenna Layout Used for Simulations


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In addition to the desire to test the antenna individually, there was also the priority

of producing a working sensor card. Since the feed line for this antenna is now a part of

the sensor card circuit, successful integration became the next step. To this end, two

cards were produced (in addition to the safe-approach card). The first one

implemented the coupler that was designed by another student and can be viewed in

figure 11a. The second card used all the same components of the safe-approach card

with alterations in layout to accommodate the new antenna feed line and to allow for the

insertion of an amplifier should one be necessary and can be viewed in figure 11b.

The last step to the design process involved the tuning of the new aperture

coupled patch antenna after milling. The resonant frequency was not where it needed to

be so initially some copper tape was added to the patch length to reduce the frequency.These pieces of tape worked very well, so they were measured and the dimensions added

to the size of the patch. Additional patched were produced for those that were already

milled and the ones that had not be produced yet were altered. The results of the antenna

tests will be displayed and discussed in the next section. The results of the actual sensor

Figure 12a: Sensor card layout

with new coupler (red is

aperture in ground plane)

Figure 12b: Sensor card layout

with only a feed line added (blue

is antenna on back of card).


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card will not be included in this report as the card was sent to TTU before anything more

than an initial test was completed.

Measurement Methods and Simulated Results

In terms of the testing of the proximity sensor antenna there are two main concerns. One

is resonant frequency as 2.45 GHz was chosen as the operating frequency because it is in

the range of frequencies not really regulated by the FCC for institutional usage. If we

operate too far from 2.45 GHz, we would be in danger of breaking a few laws. The

second main concern is that of return loss. The antenna is designed to operate at very low

power (partially to avoid cooked eggs) and as such, needs an antenna that will transmit

and receive a vast majority of the usable power. Knowing that amplification can make up

for some of the loss of power on the return path, the necessary return loss was set at 10

dB.

The first iteration of antennas, which was the probe-fed patch, resonated at 2.43

GHz with a return loss of only 8.7 dB. This antenna was used in conjunction with the

original safe-approach sensor card and valid calibration data was obtained which was

the goal for that antenna. However, the voltages produced by the mixer during this trial

were considerably lower than those produced by the first coaxial sensor circuit that we

measured. This would be because the signal was too weak which would be the direct

effect if the return loss was not optimum. The results of this measurement made with the

HP8714 Vector Network Analyzer can be seen below in figure 12.

Lowest Reflection at Resonant Frequency

2.43GHz

Figure 13: VNA

generated plot of S11.


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In addition, this measurement was made with the Anritsu Scorpion, which can

measure up to 6 GHz to allow viewing of any additional resonances due to other patch

dimensions besides the length. This data was saved in .S2P format and then imported

into ADS to allow simple comparisons to the simulations. Figures 13a and b show this

comparison and as can be clearly seen, the response of the measured data is a little closer

to the design value because of the removal of the thin strip of copper that was mentioned

earlier. In addition, it is apparent that there are three resonant frequencies within the

range in which we are measuring but the 2nd and 3rd are far away from the design

frequency so not in any danger of interfering. The smith chart plot also shows that the

marker frequencies (1st resonance) are closer to the center of the chart indicating a better

match than the other two frequencies. (Note: In all of the following plots, red is measured

data and blue is simulated data)

Figure 14a: Comparison of Measured data with Simulations for Probe-fed Patch

m 2freq=2.428GHzdB(S(1,1))=-9.435

m 1freq=2.337GHzdB(S(3,3))=-10.807

0 1 2 3 4 5 6

-4 0

-3 0

-2 0

-1 0

0

10

Frequency (GHz)

Refle

ction

(dB) m 2m 1

Probe-fed P atch Simulated Data vs. M easurements


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m1freq=2.337E9HzS(3,3)=0.288 / 172.253impedance = Z0 * (0.554 + j0.047)

m2freq=2.428E9HzS(1,1)=0.337 / 122.032impedance = Z0 * (0.602 + j0.389)

freq (10.00kHz to 6.000GHz)

S(1,1

)

m2

S(3,3

)m1

Probe-fed Patch Meas ured Data vs. Simulat ion

Figure 14b: Smith chart plot of measured data versus simulations for probe-fed patch

The second iteration on which measurements were made was the first aperture

couple fed antenna produced (separate from the sensor card). The same measurements

were made with the Scorpion data compared to the simulated data shown in figures 14a

and 14b.

Figure 15a: Comparison of Measured data with Simulations for Couple-fed Patch

m2freq=2.721GHzdB(S(1,1))=-9.942

m1freq=2.450GHzdB(S(3,3))=-17.190

0 1 2 3 4 5 6

-20

-15

-10

-5

0

5

Frequency (GHz)

Reflection

(dB)

m2

m1

Initial Couple-fed Patch Simulated Data vs. Measurements


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m2

freq=2.721E9HzS(1,1)=0.318 / -9.351impedance = Z0 * (1.899 - j0.219

m 1

freq=2.450E9HzS(3,3)=0.138 / -19.267impedance = Z0 * (1.294 - j0.120


S(1,1

)m 2

S(3,3

) m 1

Initial Couple-fed Patch Measured Data vs. Simulation

Figure 15b: Smith chart plot of measured data versus simulations for couple-fed patch

As may be apparent from the plots, the resonant frequency is quite a bit off from

the design value and with not nearly as good of a return loss value. As mentioned before,

to correct for this, small pieces of copper tape were used to increase the length of the

patch to hopefully bring the frequency closer to the design value. Additionally, it can be

observed from the log-mag format plot, that there was a slight calibration flaw where the

value of S11 goes greater than zero. This error was corrected for in the next set of

comparisons.

Finally, after the copper tape additions had been replaced by a newly milled patch

design (larger by the copper tape amount) a second set of measurements were made with

the Scorpion but with a better calibration. These measurements are compared to

simulated data in figures 15a and 15b.


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m2freq=2.473GHzdB(S(1,1))=-34.957

m1freq=2.450GHzdB(S(3,3))=-17.190

0 1 2 3 4 5 6

-4 0

-3 0

-2 0

-1 0

0

10

Frequency (GHz)

Reflection

(dB)

m2

m1

Final Couple-fed Patch Measured Data vs. Simulation 1

Figure 16a: Comparison of measured data with original design simulations for the

tuned couple-fed patch.

m 2freq=2.473E9HzS(1,1)=0.018 / 158.407impedance = Z0 * (0.967 + j0.013

m1freq=2.450E9HzS(3,3)=0.138 / -19.267impedance = Z0 * (1.294 - j0.120


S(1,1

) m2

S(3,3

) m1

Final Couple-fed Patch Meas ured Data vs. Simulation 1

Figure 16b: Smith chart plot of measured data versus original simulations for second

couple-fed patch


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It is worth noting from the above displays, that the actual antenna response, while

not quite at 2.45 GHz, was better matched to 50W than was the simulation. M2 in the

above plot is virtually in the center of the smith chart with the display indicating that the

impedance is equal to Zo*(0.967 + j0.013). An ideal load would be Zo*(1+j0.0). This

match explains why the return loss of the actual antenna is almost doubled that of the

original simulation.

The fact that the antenna that was increased in length came closer to resonating at

the same frequency as the simulation than did the original design (which had the same

dimensions as the simulated patch) leads one to believe that one or more of the hand-

keyed parameters might be incorrect. It was noted that without the ground plane on the

FR4 (as was the case with the antenna layout), it is obvious that different sheets of

substrate are of different colors (one was yellow, the other white). If the dielectric

constant was, in reality, different than designed with, it could significantly change the

necessary design. If the dielectric is increased, it would increase the resonant frequency

by reducing the size of the wavelength in the dielectric. Because of this, it is apparent

that the substrate used for milling the antennas could quite possibly have had a higher

constant that 4.3.

Conclusions and Recommendations

As the design and testing process drew to a close, it became quite clear how important it

was to have left some room for flexibility in the original design schedule. Uncontrollable

difficulties, such as unexplainable glitches in the simulation programs, arose that delayed

the production process by weeks. In the case of the antenna and the sensor card, this

prevented extensive testing of the final product. Overall, this particular design process

was very successful. The antennas were produced on schedule (for the most part) and to

specifications. As this was a group project, one of the original goals was to stick to the

individual schedule as closely as possible so as to not hold up anyone elses design

production. This particular goal was met without too much difficulty. In addition, and

more importantly, the final antenna was completed with the antenna residing directly on

the back of the ground plane of the sensor card, resonating at 2.45 GHz and giving a


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return loss of considerably better than 10 dB. Further investigation into this design

process could involve altering the design to affect radiation pattern and investigating how

this affects the ability of the sensor to monitor activity in a nest. In addition, some time

needs to be invested into determining what the voltage output values of the sensor card

are actually indicating. Had the sensor card that included this new improved antenna

been available for testing, some in-depth calibration trials could have been completed

towards this end.

Acknowledgements

The author would like to thank a few people without whose help and patience, this

project would not have been possible. The first is Dr. Rudolph Henning for not only

lending all of his antenna books but also for allowing the freedom to work on my own

schedule. The second two people who should be acknowledged here are Cheevin

Chulikavit and Marc Ejgird both of whom accepted our confused scheduling without

complaint and who completed circuit milling overnight as needed.

1T. Weller, Wireless Circuits and Systems Laboratory #12: Dipole and Patch Antennas, University of

South Florida, w12-exp-010101, 2001.2 D.M. Pozar, Microstrip Antennas, Proc. IEEE, Vol. 80, No. 1, pp. 79-81, January 1992.3

D.M. Pozar, Microstrip antenna aperture-coupled to a microstrip-line,Electronic Letters, Vol. 21, No.

2, pp. 49-50, Jan. 1985.4

H.A. Bethe, Theory of Diffractions by Small Holes, Physical Review, Vol. 66, pp. 163-182, 1944.5

C. A. Balanis,Antenna Theory: Analysis and Design, Second Edition, John Wiley & Sons, Inc. 1997.6

D.M. Pozar, Microstrip Antennas, Proc. IEEE, Vol. 80, No. 1, pp. 79-81, January 1992.7

P.L. Sullivan and D.H. Schaubert, Analysis of an aperture coupled microstrip antenna, RADC-TR-85-

274, Rome Air Development Center, Feb. 1986.8

D.M. Pozar, Personal Computer Aided Antenna Design for Windows: version 3.0,Antenna Design

Associates, Inc. Leverett, MA 1996.

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