b achelor s t hesis - cernorf example, the 100 vk supply voltage of the klystron needs a 10 5...

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Karlsruhe University of Applied Sciences Department of Electrical Sciences and Information Technology Electrical Engineering – Sensor Technology

BACHELOR’S THESIS

by

Philip Maus

Phase and amplitude stability of a pulsed RF system on the example of the

CLIC drive beam LINAC Workplace: CERN Supervisor at the workplace: Dr. Steffen Doebert Instructor: Prof. Dr. Michael Bantel Co-instructor: Prof. Dr.-Ing. Ulrich Grünhaupt Closing date: 01.03.2017 Bachelorthesis-Nr.: 297 Processed in the time from 01.11.2016 to 01.03.2017

Contents

1 Motivation 1

1.1 CERN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.2 CLIC study . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.3 CLIC Drive Beam injector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2 Pulsed radio frequency signals 5

2.1 General Radio frequency Information . . . . . . . . . . . . . . . . . . . . . . . . . . 52.2 RF signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52.3 Power and gain units . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52.4 Amplitude-pulse modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62.5 Transmission line characterization . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

3 Phase and amplitude measurements 8

3.1 Measurement aims . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83.2 Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

3.2.1 Hardware setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93.2.2 Software setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

3.3 Delay Generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103.3.1 Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103.3.2 Rise and fall time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

3.4 Signal Generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123.4.1 Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123.4.2 Comparison of SG392 and R&S SMB100A . . . . . . . . . . . . . . . . . . . 13

3.5 Amplier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193.5.1 Bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193.5.2 Gain characterization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 203.5.3 Pulse shape and rise and fall time . . . . . . . . . . . . . . . . . . . . . . . . 223.5.4 Amplitude attop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

3.6 Data acquisition system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 253.6.1 AD8302 measurement board . . . . . . . . . . . . . . . . . . . . . . . . . . . 253.6.2 Measurement setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263.6.3 Amplitude attop comparison AD8302 measurement board and power meter 273.6.4 Amplitude attop comparison AD8302 measurement board and AD8302-

EVALZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 283.6.5 Phase attop comparison AD8302 measurement board and AD8302-EVALZ 303.6.6 Bit noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 333.6.7 AD8302-EVALZ and measurement board layout comparison . . . . . . . . . 34

4 Measurement conclusion and setup recommendation 35

List of Figures 37

References 38

Appendix 39

Philip Maus Page I Bachelor's Thesis

Nomenclature

BCD Binary Coded Decimal

CERN Conseil Européen pour la Recherche Nucléaire

CLIC Compact Linear Collider

CW Continuous Wave

DBI Drive Beam Injector

DBL Drive Beam Linac

FFT Fast Fourier Transform

GPIB General Purpose Interface Bus

LDO Linear Dropout

LHC Large Hadron Collider

LINAC Linear Accelerator

ML Mismatch Loss

PETS Power Extraction and Transfer Structures

RF Radio Frequency

RL Return Loss

SBB Single Sideband

SHB Sub Harmonic Buncher

TTL Transistor-Transistor Logic

VSWR Voltage Standing Wave Ratio

Philip Maus Page II Bachelor's Thesis

1 Motivation

1.1 CERN

The European Organization for Nuclear Research (French: Organisation Européenne pour laRecherche Nucléaire), known as Conseil Européen pour la Recherche Nucléaire (CERN), is aEuropean research organization that operates the largest particle physics laboratory in the world.Established in 1954, the organization is based in a northwest suburb of Geneva on the Franco-Swissborder. CERN represents a global international organisation with 22 member states, mainly fromEurope, associate member states, and observer states.CERN was ocially founded on the 29th of September 1954 by the twelve founding Member Statesand Geneva was picked as its location. In 1957 the rst particle accelerator, the Synchrocyclotron(SC), was taken in operation, and running until 1990, mainly focused on particle physics [1]. CERNhas about 2,250 sta members, but with students, fellows, PhDs, users, sub-contractors and visit-ing scientists, the number of people on site can climb up to 13,000 if all facilities are running.Presently protons and ions are accelerated in up to ve stages and injected into the 27 km longcircular collider Large Hadron Collider (LHC), where they collide inside four detectors - ALICE,ATLAS, CMS and LHCb - with a center of mass energy of 13TeV. With the help of this largestaccelerator in the world, signicant scientic progress like the renement of the standard modelwith the discovery of the Higgs boson in 2012 and the investigation of supersymmetry is possible.Additionally, CERN features smaller projects like the Antiproton Decelerator and the on-line Iso-tope mass separator ISOLDE.Future planning and research studies for the lab include two major accelerator projects:

FCC, a 100 km long circular collider, which could allow proton collisions with up to 100TeV.

CLIC, a 50 km long linear electron-positron collider for complimentary investigations of theLHC ndings

1.2 CLIC study

The particle physics community worldwide has expressed a need for further studies of the LHCresults by a complementary lepton collider in the TeV-range. This would allow, for example, tostudy the properties of the Higgs-Boson in detail with pointlike particles. Circular lepton collidersin the TeV energy range are not possible within a reasonable size due to synchrotron radiationlosses. Therefore a future high energy lepton collider would use two linear accelerators head-on tocollide electrons and positrons in a central detector [2].

Figure 1: Artist view of the Compact Linear Collider1

Philip Maus Page 1 Bachelor's Thesis

The Compact Linear Collider (CLIC) study is an international collaboration working on aproject for a machine to perform electron(e−)-positron(e+)-head-on-collisions at energies up to3TeV. An artist view of the CLIC project implemented in the Geneva region can be seen in gure1. A 50 km long deep underground tunnel would house the accelerator with a central collisionpoint on the existing CERN site.

Figure 2: CLIC layout at 3TeV2

The main objective is to build at aordable cost and at a reasonable size a linear collider for theMulti-TeV-range. Therefore, the acceleration gradient, the amount of energy that can be addedto the main beam per meter, is one of the key parameters. For the CLIC study an acceleratinggradient of 100MV/m using normal-conducting structures at a frequency of 12GHz was chosen.

Figure 3: CLIC two-beam scheme3

1Source: http://clic-study.web.cern.ch, access: 07.02.20172Source: [2], g. 2.1, p. 9


Since there are no ecient microwave power sources at 12GHz available, a two beam schemewas invented. A low-energy but high-current drive beam is generated to serve as a power sourcefor the high-energy, low-current main beam in a kind of transformer scheme. The layout of the3TeV version of CLIC can be seen in gure 2. The high quality main beams are generated andpre-accelerated in the injector Linear Accelerators (LINACs) and damping rings and then furtheraccelerated and transported to the far end turnarounds (TA). In the main LINAC the electronsand positrons are accelerated to their nal energy using the 12GHz microwave power directlygenerated by decelerating the drive beam running in parallel to it. These drive beams are generatedeciently with a low frequency and current which is then accumulated in a series of delay loopsand combiner rings to reach the high drive beam current and bunch repetition frequency. Figure3 shows the CLIC two-beam scheme, including the Power Extraction and Transfer Structures(PETS) and accelerator structures, which couple the drive and main beam and provide the energytransfer. The main reason to use this scheme is a superior eciency and lower cost comparedto alternative power sources such as klystrons. With this layout, CLIC reaches an accelerationgradient of 100MV/m and center-of-mass collisions with an energy of 3TeV and a luminosity of2× 1034 cm−2s−1.

1.3 CLIC Drive Beam injector

The CLIC drive beam accelerator (g. 4) consists of the Drive Beam Injector (DBI) and two DriveBeam Linacs (DBLs). The drive beam injector is composed of a thermionic electron source, 3 SubHarmonic Bunchers (SHBs), a pre-buncher, and several acceleration structures.

Figure 4: CLIC Drive Beam injector4

3Source: [2], g. 2.6, p. 274Steen Doebert: "High-eciency L-band klystron development for the CLIC Drive Beam", CLIC workshop,

CERN, 18-22.01.2016


In the electron source the DC electron beam is produced from a thermionic cathode. Thefollowing buncher cavities group ("bunch") the electrons to be accelerated by RF later on. Eachelectron bunch has an energy of 140 keV, a length of 3mm, and a charge qb = 8.4 nC. Afterwardsthe electrons are accelerated in the 1GHz accelerating structures up to 50MeV. The pulsed RadioFrequency (RF) power for this acceleration is provided by 1GHz, 20MWmodulator-klystron units,one per acceleration structure. A klystron is an RF amplier based on a linear-beam vacuum tube.The high voltage modulator supplies the acceleration voltage to this tube. A DC electron beamgets modulated with an input signal, the modulation enhances in a drift space, and nally thepower gets coupled out into a waveguide.

Due to the fact that the main beam of CLIC is powered by the drive beam, the quality andstability of the main beam depends on the current, phase, and energy stability of the drive beam.There are very stringent requirements on the beam stability of the main beam which are thentranslated into requirements of the amplitude and phase stability of the klystron units of the drivebeam. The drive beam current stability is predened by the electron gun, while the drive beamphase and energy quality is dened by the klystron output phase and amplitude. Since the klystronis an amplier the output stability depends heavily on the input signal and the supply voltages.For example, the 100 kV supply voltage of the klystron needs a 10−5 stability. The specicationfor the RF drive signal to the klystron is an amplitude stability of 0.1% and a phase stability of0.05 degrees pulse to pulse.

The subject of this thesis is the generation and pre-amplication of the pulsed RF klystron inputsignal of the CLIC drive beam accelerator, with the above specications.


2 Pulsed radio frequency signals

2.1 General Radio frequency Information

Radio Frequency (RF) is the frequency range of the electromagnetic waves in the spectrum from3 kHz to 300GHz [3]. Which equals, according to the formula λ = c/f , with λ as the wavelength, cas the speed of light, and f as the frequency, the radio spectrum containing the wavelengths from100 km to 1µm.

Figure 5: Radio frequency spectrum5

The radio spectrum can be split up in smaller parts (g. 5), from Low Frequency used forradionavigation over Middle and High (AM broadcasting, aeronautical radionavigation) up toUltra and Super High Frequency (satellite communication) [4].

2.2 RF signals

An RF signal can be generally dened as:

E = E0 · cos(ωt+ φ) (1)

With E being the amplitude, ω as the angular frequency and φ as the phase [5].Formula 1 refers to an ideal RF signal. In the real world a signal can be rather described as:

E = (E0 + E(t)) · cos(ωt+ (φ+ φ(t))) (2)

With E(t) representing amplitude uctuations and φ(t) as phase noise. Further details and mea-surements concerning phase noise can be found in chapter 3.4.2.

2.3 Power and gain units

A common relative power unit used in RF signal generation is dB.

power in dB = 10 · log(P1

P2

)(3)

RF units require a logarithmic scale due to the wide ranges of powers and voltages used. Due toP ∼ V 2, voltage in dB is dened as

voltage in dB = 20 · log(V1V2

)(4)

All the following dB-statements will refer to the power, so formula 3 will be used.

In case absolute power values are needed, a power measurement can be referenced to 1 milliwatt:

power in dBm = 10 · log(

P

1mW

)(5)

5Source: http://sss-mag.com/spectrum.html, access: 11.11.2016


2.4 Amplitude-pulse modulation

Analog amplitude-pulse modulation is a type of modulation in which the carrier wave c(t) is varieddepending on the amplitude of a control signal s(t). The output power varies between noise fors(t) `low' and a maximum power level for s(t) `high'.

For the reason that the setup and measurements described in this thesis are based on amplitude-pulsed RF signals, some of the characteristics are elaborated in the following and will be furtherindicated as quality criteria.

Flattop Flattop is a term used to describe the quality of a pulsed signal. Hereby a pulse getsrated by the atness of its top, concerning aspects like increase and decrease of the signal levelover the pulse length (attop variation), ripples and unevenness, and repeating irregularities likenoise frequencies and harmonics.

Repeatability To achieve a good circuit performance it is necessary to guarantee high attoprepeatability of the pulsed signal. Repeatability is often even more important than the pure pulseshape and characteristics. A constant variance can be either counterbalanced or the circuits canbe adjusted to it, while unpredictable irregularities worsen the system's behavior.The attop repeatibility is assessed with standard deviation measurements. The standard deviationσ describes the dierence of a population x, consisting of n values, relative to its average valuex = 1

n

∑ni=1(xi) and is dened as

σ =

√√√√ 1

n·

n∑i=1

(x− x2) (6)

Method denition To calculate the standard deviation, every recorded pulse, with its pulselength of 150 µs, is split up in 15 parts. For each of those 10 µs segments the average value over thispart of the pulse is computed. The standard deviation over all measured pulses is determined withthese mean values over 10µs, therefore resulting in 15 absolute std values for the 150 µs pulses.

2.5 Transmission line characterization

The process of RF signal transmission includes, unlike DC or lower frequencies signal transfers,some additional challenges.

Reective coecient For RF circuits used in the measurement process of this thesis a termi-nation resistance of 50Ω is dened as the standard value and therefore used for every cable, plug,and device. In a perfect transmission process all components have the same impedance. But inreal application inaccuracies occur. Therefore the reective coecient Γ is introduced.

Γ =Z1 − Z2

Z1 + Z2

(7)

Z1 is the impedance of the source and Z2 the impedance of the load. The principle is comparableto Fresnel's law.


The reective coecient can also be dened as the ratio of the complex amplitude between thereected wave Vref and the incoming wave Vin.

Γ =VrefVin

(8)

VSWR Voltage Standing Wave Ratio (VSWR) is another method to characterize the qualityof a power transmission. In case a reection occurs on the interface between source and loadcaused by mismatched impedances, inaccuracies in resistors, or irregularities in cables and plugs,the reected signal interferes with the source signal, either constructive or destructive, leading totime- and distance-depending peaks and valleys in the voltage level along the line. VSWR allowsmeasurements on these aberrations by determining the ratio of the highest to the lowest voltagevalue along the transmission line. In case of a perfect transmission and no reections occurring,VSWR is 1.0, or expressed in another way, 1:1. The VSWR is dened as:

V SWR =|Vmax||Vmin|

(9)

With Vmax being the maximum and Vmin the minimum of the standing wave. Assuming themaximum is the superposition of the incoming and the reected wave, Vmax = Vin + Vref , whileVmin is a result of destructive interference, Vmin = Vin−Vref . Combining those results with formula9 and substituting with formula 8 results in the denition:

V SWR =1 + |Γ|1− |Γ|

(10)

Return loss and Mismatch loss The Return Loss (RL) describes the loss of signal power ofthe reected signal compared to the initially inserted one.

RL = 10 · log(Pin

Pref

)(11)

Combined with formula 8 and with a factor of 20, due to Γ referring to the voltage

RL = −20 · log(Γ) (12)

As opposed to the RL the Mismatch Loss (ML) determines the amount of power which is lackingat the end of the transmission line in dB.

ML = 10 · log(

Pin

Pin − Pref

)(13)

Following the same train of thought and steps as for the RL, ML can also be dened as

ML = −20 · log(1− Γ2) (14)


3 Phase and amplitude measurements

3.1 Measurement aims

The task of this thesis is the generation and pre-amplication of the pulsed RF klystron inputsignal of the CLIC drive beam. It can be assumed that the klystron is not inuencing the qualityfactors of the RF signals, therefore, the requirements demanded at the output of the klystron applyfor the klystron input signal.

The CLIC drive beam accelerator structure requires an RF signal with a frequency of 1GHz,a pulse time of 150µs, and a repetition rate of 50Hz. A pulse amplitude stability of 0.1% andphase stability of 0.05 degree is necessary. The klystron needs a pre-amplied RF input signalwith a power of 200W.

The generation and pre-amplication of this pulsed RF signal requires the characterization ofall devices and components, the calibration and automation of the measurement process, thecomparison of dierent devices to identify the most suitable setup, the ensuring of the requiredmeasurement resolution to gain reliable data, and the analysis of the measurement results.

The nal aim of this thesis is to provide a recommendation concerning the choice of devices,identify the suitable input power levels, provide a small and cost-ecient measurement setup,and determine the occurrent phase and amplitude pulse stability based on the obtained data, sothe system can be integrated in the CLIC drive beam structure, fullling the requirements of theklystron input signal.


3.2 Setup

3.2.1 Hardware setup

Figure 6: Phase and amplitude stability measurement hardware setup

Figure 6 shows the hardware setup for phase and amplitude stability measurements. A delaygenerator serves as the initial trigger, creating sharp 150µs, 5V pulses with a repetition rate of50Hz. These are provided to the RF signal generator, which is therewith amplitude-modulatingan internally generated Continuous Wave (CW) 1GHz signal. The output is transferred to thepulsed power amplier, which is controlled via a Transistor-Transistor Logic (TTL) control signalprovided by the delay generator. The pulsed RF amplier output signal is recorded either by apeak power meter, which performs an absolute measurement, or a combination of a phase and gainmeasurement board and an oscilloscope. The measurement board allows relative measurements,therefore the RF signal generator output signal is used as a reference.

3.2.2 Software setup

General Purpose Interface Bus (GPIB), also known as IEEE 488, is a highly popular and versa-tile interface standard. Although by now surpassed by other technologies, since rst registeredin 1978, it is still frequently used as the main peripheral bus for microcomputers and device-computer-communication. It is an 8-bit parallel master-slave bus using TTL-logic levels and has a1Mb/s peak transfer rate. Devices are associated with a 5 bit Binary Coded Decimal (BCD) codeand the number of instruments is limited to 15 [6].

The delay generator, signal generator, peak power meter, and oscilloscope are initialized, con-trolled and read out via GPIB and a script written in MATLAB. This allows the user to executemeasurements automatically and gather and process the data without relying on numerous devicesand scripts.


3.3 Delay Generator

The delay generator fullls two main tasks

Trigger generation for RF signal generator pulse modulation

Creation of the TTL control signal for the pulse amplier

A Stanford DG645 Digital Delay Generator is used for this cause. It is a remotely controllable 4channel delay generator creating pulses with a rate from 100µHz to 10MHz with a resolution of1µHz. The outputs are characterized with a source impedance of 50Ω and a voltage accuracy of100mV + 5% of the pulse amplitude [7].

3.3.1 Timing

Figure 7 shows the timing of the delay generator with the pulsed signal used for amplitude modu-lation of the 1GHz carrier frequency and triggering of the measurement device in channel 1, andthe TTL control signal for the amplier in channel 2.

Figure 7: Delay generator DG645 timing overview

The amplier requires a negative TTL logic and has to be activated 5 µs before and deactivated5µs after the active pulse from the delay generator. Further information concerning the ampliercan be found in section 3.5.

3.3.2 Rise and fall time

The DG645 rise and fall time measurement for a 150 µs long 5V pulse with a 50Hz repetition ratecan be seen in gure 8. Due to the fact that signal generator external pulse modulation triggerhas with a TTL logic, not the standard rise and fall time (10 to 90% of voltage and inverse), butthe time, after which the opposite TTL level is reached, is relevant. TTL input levels are for `high'2-5V and for `low' 0-0.8V. The time it takes the delay generator to switch its output level from10% of the maximum voltage (5V) to the lower edge of the TTL `high' level is called ∆trise_TTL.It is measured as 0.7 ns. And ∆tfall_TTL (90% of maximum voltage to 0.8V) is determined by themeasurement as 2.2 ns.


(a) Rise time, 1 ns/div (b) Fall time, 1 ns/div

Figure 8: Delay generator DG645 edges characterization

Compared to the 150µs pulse length and the SG392, which has in amplitude-pulse modulationmode an RF rise/fall time of 20 ns and a turn on/o delay of 60 ns [8], the DG645 rise and falltimes are negligible.


3.4 Signal Generator

A signal generator or synthesizer is an active RF device to create RF signals. It allows the userto generate a CW RF signal with adjustable frequency, amplitude, and phase shift. The signalcan be modied with several analog and digital modulations. Due to this, a synthesizer forms thesource of the RF setup. The quality of the synthesizer, therefore, strongly denes the quality ofthe RF signal, concerning phase, and amplitude stability.

The RF signal generator amplitude-pulse-modulates an internally generated carrier frequency of1GHz with the pulsed signal it receives from the delay generator. A description of the theoreticalbackground of pulse modulation and its characteristics can be found in section 2.4.

3.4.1 Timing

This section gives an overview over the timing concerning the amplitude-modulated output signalof the signal generator SG392.

Figure 9: Trigger and RF signal generator outpu signalmodulation timing

Figure 10: SG392 output for 1GHz

Figure 9 displays in diagram 2 the timing of the amplitude-modulated signal generator outputsignal. An internally created CW signal is modulated with the trigger signal generated by thedelay generator (diagram 1). The SG392 RF output signal varies between noise and denedoutput power. The duty cycle, the percentage of time in which the signal output is activated, fora pulse length of 150 µs and a repetition rate of 50Hz, is 0.75%. Figure 10 displays the output ofthe SG392 recorded with a peak power meter, with a carrier frequency of 1GHz and an outputpower of 10 dBm. Compared to the output of the delay generator (paragraph 3.3.2) the rise timeincreases to 80 ns. The power fall requires around 1µs to a power level of -15 dBm and around200 µs to settle at the noise level, slightly exceeding the characteristics of the signal generatoroutput.


3.4.2 Comparison of SG392 and R&S SMB100A

For the generation of the amplitude-modulated signal two dierent signal generators are used:

SG392, a DC to 2.025GHz RF signal generator made by Stanford Research Systems

SMB100A, a Rohde&Schwarz RF and Microwave Signal Generator with a range from 100 kHzup to 40GHz

Both signal generators are installed in the same setup, which is displayed in gure 6. Theyboth amplitude-modulate an internally generated 1GHz carrier frequency with the pulsed signalgenerated by the DG645 and are remotely controlled via GPIB. All measurement data used andplotted in this chapter is recorded with a Boonton4500B peak power meter.

Amplitude attop Due to the fact that the setup requires the best attop behavior, dierentfactors have to be considered. One of them is the RF signal generator output power. Recordedare 100 measurements per power level and RF signal generator, each with the same parameters,to identify potential uctuations in the output power.

Figure 11 displays the behavior of the SMB100A concerning the variation of the output powerin a range from 5 to -15 dBm.

(a) 5 dBm output power (b) -5 dBm output power (c) -15 dBm output power

Figure 11: SMB100A attop output power comparison

It can be seen that the SMB100A has the lowest pulse-to-pulse attop standard deviation for-5 dBm output power. Figure 11b shows the 100 traces recorded being highly parallel and with adispersion of 0.02 dB. Compared to that the records for 5 dBm (g. 11a) are distributed in a widerrange.

Figure 12 gives an overview over the power output of the SG392 for the three power levels 5,-5 and -15 dBm. The SG392 shows the same behavior as the SMB100A concerning low outputpowers (compare g.11c), namely a rise in the attop standard deviation due to measurementnoise. For 5 and -5 dBm input power the signal stability is comparable.


(a) 5 dBm output power (b) -5 dBm output power (c) -15 dBm output power

Figure 12: SG392 attop output power comparison

A wider overview on the inuence of the signal generator output power on the attop stabilityis given in gure 13. It displays the standard deviation of the pulse attop over the time afterthe trigger in µs for the two signal generators, SG392 and R&S SMB100A, for output powers in arange from -15 to 5 dBm.

The absolute amplitude standard deviation stdabsdBmis determined following the method clari-

ed in paragraph 2.4. With the help of formula 5 the power levels Pmean (output power in Watt),P+ (upper power level of std in Watt) and P− (lower power level of std in Watt) can be dened:

Pmean = 10∧(dBmmean

10

)(15)

P+ = Pmean +stdP,abs

2= 10∧

(dBmmean + stddBm,abs/2

10

)(16)

P− = Pmean −stdP,abs

2= 10∧

(dBmmean − stddBm,abs/2

10

)(17)

, dBmmean being the mean power level in dBm and stdabsdBmbeing the absolute standard deviation

value.

The output power standard deviation in % is dened as

Pstd,% =∆P

Pmean

· 100 =P+ − P−Pmean

· 100 (18)

Combining formulas 15 to 18, the output power std in % is dened as

Pstd,% =

[10∧

(stddBm,abs

20

)− 10∧

(−stddBm,abs

20

)]· 100 (19)

The SMB100A 5dBm input power result was omitted from this diagram due to reaching anstd level of over 1.1%.


Figure 13: SG392 and SMB100A attop stability comparison

For the two power levels -5 and -10 dBm, both signal generators create a signal with a com-parable standard deviation concerning the pulse attop, and for both the std doubles to 0.2%for -15 dBm output power due to increasing inuence of the measurement noise. Although theSMB100A has the ability to generate higher output powers (max. +30 dBm [9]; SG392: max.+16.5 dBm, [8]), higher inaccuracies occur for the power levels over 0 dBm than for the Stanforddevice. The optimal working point concerning amplitude attop stability can be dened for theSG392 between 0 and 5 dBm and for the SMB100A between -5 and -10 dBm output power. TheStanford device outperforms the Rohde& Schwarz with a lowest standard deviation value of 0.055%compared to 0.105%.

Phase noise

An ideal clock signal can be described as

V (t) = A · sin(2πf0t) (20)

It has a constant clock frequency and constant rise and fall time. This ideal signal only exists intextbooks. In reality, characteristics like frequency, signal period, and rise and fall time vary. Ajitter, a deviation in time from the ideal reference clock, is existing. The signal can be describedas

V (t) = [A+ E(t)] · sin(2πf0t+ φ(t)) (21)

, with E(t) being the amplitude uctuations and φ(t) representing the phase uctuations. Am-plitude variations can be well-controlled and do not strongly inuence the system, while phasevariations have a much higher impact on the functionality. Therefore a phase noise analysis isnecessary to characterize the available signal generators. One source of phase noise is long-termfrequency instability, including variations occurring over a time frame of days, months or evenyears. The report will not consider those due to their minimal impact on the measurement results(SG392: Rubidium timebase aging: <0.001 ppm/year [8]) and focus on short-term variations.


Phase noise can be split up in two components:

random phase noise

discrete phase noise

To distinguish the dierent kinds of phase noise and quantify its value, a Fast Fourier Transform(FFT) of the source signal is necessary.

A typical RF oscillator output spectrum is shown in gure 14. The spectrum is symmetricalaround the carrier frequency f0. An ideal signal would have a dirac pulse as spectrum, but due tonoise, the amplitude decreases gradually with the dierence between f and f0 increasing. Whilerandom phase noise is caused by thermal or other random reasons and is therefore describablewith a Gaussian distribution, discrete noise creates spurs caused by harmonic frequencies or mixerproducts. These can be clearly identied in the spectrum.

Figure 14: Output spectrum of a typical RF oscillator6

The term phase noise measurement refers to the Single Sideband (SBB) phase noise, generallydenoted as L(f), and dened as the ratio of the power density at a specic oset frequency fromthe carrier to the total power of the signal. Concerning the spectrum plot L(f) is dened as thepower, corresponding to the area under the curve for a 1Hz bandwidth, in relation to the totalpower, equivalent to the total area under the curve. Its unit is dBc/Hz (decibels relative to outputlevel per Hertz) [10].

The three existing phase noise measurement techniques are

direct measurement with a spectrum analyzer

the phase detector technique, where a phase dierence is transformed into a voltage

the two-channel cross-correlation method, which was used for the following measurements

The two-channel cross-correlation method combines two single-channel reference sources and per-forms cross-correlation between the outputs of each channel. This means that, while the deviceunder test's noise NDUT is unaected by this measurement method, the two internal noises N1 and

6Source: [10], page 594, gure 12.13


N2, generated by the two channels, are diminished by the cross-correlation operation due to thefact that they are incoherent. The measured phase noise Nmeas can be dened as:

Nmeas = NDUT + (N1 +N2)/M7 (22)

M is the number of correlations. Thanks to the cross-correlation, this method achieves superiorresults at the cost of a rising number of correlations requiring a longer measurement time [11].

Figure 15 displays the phase noise comparison for the signal generators SG392 and SMB100Agenerating a 10 dBm, 1GHz, 150 µs, 50Hz repetition rate pulse, recorded with the two-channelcross-correlation method with a Keysight E5052B signal source analyzer. The settings are a fre-quency range from 1Hz to 10MHz oset from the carrier frequency, averaging of 8 and correlationof 10, which equals a noise reduction of 5 dB.

(a) 1-500Hz oset from f0 (b) 1Hz-10MHz oset from f0

Figure 15: SG392 and R&S SMB100A phase noise measurement, correlation 10, averaging 8

The entire spectrum of the phase noise measurement for both signal generators can be seen ingure 15b. It is notable that the SG392 phase noise is higher than the R&S SMB100A one overthe entire spectrum, varying from 3dB dierence at 100Hz up to 16 dB at 10 kHz oset to thecarrier frequency. Both measurements show the same curve shape, a linear drop from 1 to 100Hz,a plateau between 10 and 200 kHz. It is followed by a drastic decrease, attening out from 6MHzon at a level of -158 dBc/Hz for the SMB100A, a level which is according to the E5052B datasheet( [12]), not yet the bottom noise of -170 dBc/Hz, a value which is recorded with a correlationof 1. As a main dierence the SG392 phase noise in a frequency range close to the carrier frequencyexceeds the one of the Rohde&Schwarz by 15 dB.The Rohde&Schwarz signal generator shows a denser appearance of spurs with a higher amplitude.To gain a closer insight in this observation and for further investigation of the cause and repetitionrate of the spurs gure 15a shows the oset frequency range from 1 to 500Hz in a non-logarithmicscale. The R&S SMB100A spurs can be clearly identied as being caused by an internal harmonicsor ringing frequency of the oscillator of 150Hz. At 150Hz and all its multiples a spur occurs, withthe amplitude 20 dB, cut in half for each following spur. A second less intense harmonics frequencyof 8Hz can be identied. The SG392 has a ringing frequency of 47Hz with an intensity comparable

7Source: [11], page 38


to the smaller harmonics of the R&S SMB100A with a value of 3 dB, which is negligible for osetfrequencies higher 200 kHz.The SG392 slightly underperforms compared to the phase noise value specied in the manual (-80 dBc/Hz at 10Hz, -102 at 1 kHz, -116 at 20 kHz and -130 dBc/Hz at 1MHz [8]), a discrepancywhich can be explained by the fact that a correlation value of 10 is not high enough to even out theentire internal noise of the measurement device. The same eect occurs for the Rohde & Schwarzdevice with a specied phase noise of -131 dBc/Hz for 20 kHz [9].

Conclusion The SG392 has a higher phase noise level but generates a cleaner signal concern-ing phase noise spurs. It can be assumed that the Stanford device's output signal is mainlyinuenced by random phase noise, which has a Gaussian distribution, while the Rohde&Schwarzdevice shows discrete phase noise. For the CLIC drive beam klystron RF input signal one majorrequirement is the amplitude attop stability. In this aspect the SG392 signal generator outper-forms the SMB100A. For the nal signal generator choice, the amplitude and phase stability afterthe amplier will be additionally taken in consideration.


3.5 Amplier

An amplier is an active RF device amplifying the RF signal power by a certain factor, called gain.The amplication factor depends on the input signal frequency and power. Also amplication andphase noises created vary with the input power. Therefore a precise characterization and a smartchoice of the input power is necessary. In case the working point is at a higher amplication factorthan required for the circuit, the signal can be attenuated afterwards.

The amplier used in the setup is the Microwave Amps L-Band 400W solid state pulsed power am-plier AM10, with a maximum output power of 400W. It has a frequency range of 995 - 1005MHz,a small signal gain of 59 dB min, and a specied output power of +57 dBm. The duty cycle islimited to 5% and the pulse rise time is about 4 µs [13]. The amplier needs an AC TTL pulsecontrol signal, which activates the amplier 5 µs before the RF pulse starts and deactivates it 5 µsafter the RF pulse ends. It is provided by the DG645 (g. 7 shows the timing).

3.5.1 Bandwidth

The AM10 bandwidth measurements, recorded for a pulsed input signal with 0 dBm power level,150 µs pulse length and 50Hz repetition rate and with the Boonton4500B peak power meter, aredisplayed in gure 16.

Figure 16: Pulsed power amplier AM10 bandwidth

The left diagram shows the overview from 900 to 1150MHz carrier frequency, which indicatesa maximum gain of 58.2 dB at 1027MHz. A closer look on the ocial frequency range, which isfrom 995 - 1005MHz , is given in the right diagram. It points out the fact that the maximumgain is not in the ocial range, but this range shows a constant rise of the amplier gain. For thecenter frequency of 1GHz the gain is 57.55 dB, a value which does not reach the advised 59 dB ofsmall signal gain. This could be caused by the fact that the measurements were only made withone input power, not guaranteeing an optimum performance. Therefore a gain characterizationis necessary. In the given frequency range of ±5MHz, the amplication varies by 0.2 dB, a valuewhich is due to the focus on repeatability of no big signicance. The half power points, thefrequencies on which the amplier gain drops to half of its peak value, indicate an bandwidth


unbalance, with fhalfpointlow being at 962MHz, 65MHz under the peak frequency, and fhalfpointhighequaling 1128MHz, 101MHz over the peak frequency.

3.5.2 Gain characterization

To determine the amplier gain, the setup shown in gure 6 is used. Based on the results fromsection 3.4.2, the signal generator SG392C is chosen as signal source, thanks to its lower amplitudeattop standard deviation.

In order to measure gain precisely, cable and plug attenuation has to be considered:

the output of the signal generator is recorded with the peak power sensor 57318 for theBoonton4500B, which has, according to its manual [14], a VSWR of 1.15 at 1GHz. Applyingthe formulas 10 and 14, this equals an ML of -0.02 dB.

the signal generator and the amplier are connected with a low power RF cable with a MLof -0.59 dB and the amplier is connected to the peak power meter with a 60-dB-couplerand a ST18A cable with -0.322 dB ML. All those values are determined with an AgilentTechnologies E5071C Network Analyzer.

Signal generator output For reliable measurement results a signal generator output powercharacterization is essential. Therefore a measurement of the SG392 output power, in a rangefrom -16 dBm to 16 dBm with a step size of 0.1 dBm, was recorded with the Boonton4500B. Theresults can be seen in gure 17. The signal generator output signal is as expected a linear graphover the input power levels. One often characterizes the quality of a signal generator by checkingits linearity. The linear regression is computed as

linRegSigGenout = 0.997 · SigGenin + 0.106 [dBm] (23)

Amplier gain With knowledge of the signal generator output power and the amplier outputpower the amplier gain can be calculated. Hereby the losses caused by cables, couplers andattenuators have to be considered.

AmpGain = Ampout − SigGenout − Losses; (24)

The results of the former measurement and the amplier gain calculation according to this formulacan be seen in gure 17.


Figure 17: Signal generator SG392 output, power amplier AM10 output and AM10 gain

The amplier output power, measured with the Boonton4500B in a range of input power levelsfrom -16 to +16 dBm with an increment of 0.1 dBm, shows a linear rise up to -10 dBm input power.In the following, the power saturates and reaches its peak at 0 dBm input with 58 dBm outputpower and declines to about 56 dBm for higher input power levels. The amplier therewith exceedsits specied output power of 57 dBm [13].The gain reaches its highest level of 65.2 dB at -13.2 dBm input power, which is 6 dB higher thanthe specied power gain of 59 dB min [13]. From this point on the amplier gain decreases steadilyand from 0dBm on linearly, due to the saturation of the amplier output signal.

The klystron following the AM10 pulsed power amplier requires an input power of 200W. This,therefore, is the minimum output power the amplier has to provide. In case a setting with a higheroutput power can be identied as the better working point, the signal can easily be attenuated.For stability it is expected to be of advantage to work in the saturation region of the amplier.Therefore two power levels will be taken under further investigation:

200W, which equals 53.01 dBm and requires an amplier input power of -12.2 dBm

400W, corresponding to 56.02 dBm and is created by an amplier input power of -7.6 dBm


Signal generator output stability Figure 18 shows the absolute linear error of the signal gen-erator output signal (SigGenout) with respect to its linear regression (linRegSigGenout), dependenton the signal generator input signal.

absLinErrSigGenout = SigGenout − linRegSigGenout (25)

It is notable that the dierence uctuates between ±0.02 dB for input values from −10 to +15 dbm,which equals 0.46%. For the higher input levels the linear error peaks in 0.06 dB, correspondingto 1.39%, thanks to higher amplitude noise created by the signal generator.

Figure 18: Absolute linear error of signal generator SG392 output over signal generator input

3.5.3 Pulse shape and rise and fall time

Figure 19 shows the pulse shape of the signal generator (blue) and the power amplier outputsignal (green) for a generator input power of -12.2 dBm recorded with the AD8302 evaluationboard (see later in chapter 3.6). The amplier output power is -6.5 dBm, a value which ts thepower amplier characterization in gure 17, considering the output is attenuated by 60 dB. Thegenerator signal has a rise/fall time of 20 ns. The power amplier output signal also has a falltime of 20 ns, but a rise time of around 5 µs, which is analogue to the specication given in themanual [13]. Figure 20 gives a closer look on the amplier output power rise, which shows ringing.


Figure 19: Signal generator SG392 (blue) andpower amplier AM10 output pulse (green)

Figure 20: Power amplier AM10 output sig-nal rise time

3.5.4 Amplitude attop

Comparable to section 3.4.2 the amplitude attop is determined as well for the power amplieroutput signal. Recorded are 100 measurements per power level and RF signal generator, each withthe same parameters, to identify potential uctuations in the output power.

Figure 21 shows the amplier output power for the two power levels, 200 and 400W, for bothsignal generators SG392C and SMB100A. As well for 200W (53 dBm) as for 400W (56 dBm) theRohde&Schwarz device shows a higher attenuation then the Stanford one. The formerly mentionedringing in the power amplier output signal rise is not visible as the rst and last 5 µs are cut o,due to the fact that the focus is on the attop stability.

(a) SG392C 200W (b) SG392C 400W


(c) SMB100A 200W (d) SMB100A 400W

Figure 21: Power amplier AM10 output power attop comparison with SMB100A input

It is notable that all four measurements show the same attop shape, including a local peakafter 50 µs. The output power does not inuence the attop variation, the dierence between thehighest and the lowest value of the pulse, which is for all pulses 0.5 dBm. The trace distributionof the SMB100A, a characteristic which relates to the attop stability, is wider then the oneof the SG392 signal generator. This observation is conrmed by the attop standard deviationmeasurements for SG392C and SMB100A for ve input power levels from -15 to 5 dBm(g. 22).

Figure 22: Power amplier AM10 output power stability comparison for SG392 and SMB100Ainput

The settings 5 dBm input for the SMB100A and 0, -7.6 and -12.2 dBm input power for theSG392 reach an amplitude attop standard deviation of 0.1% as needed. All those cases provideat least 200W amplier output power.

The Microwave Amps solid state pulsed power amplier AM10 fullls all the given requirementsby the CLIC Drive Beam structure. It is able to provide the necessary gain and output power at1GHz carrier frequency with the necessary amplitude pulse stability.


3.6 Data acquisition system

For the data acquisition system to be later used in the CLIC project, a compact and cost-ecientsolution is necessary. Therefore a commercially available AD8302 gain and phase detector wastested.

3.6.1 AD8302 measurement board

AD8302 The AD8302 is a DC to 2.7GHz RF Gain and Phase Detector. It determines theamplitude and phase dierence of two RF input signals and delivers two DC sensor signals. Thetwo input signals are required to be of the same frequency and of an amplitude in a range from-60 to 0 dBm with an amplitude dierence of not more than 30 dB. Further information about theAD8302 can be found in the data sheet [15].

Measurement board For the AD8302, a measurement board designed before my arrival atCERN was supposed to be assembled, calibrated and tested for its performance. The schematics(g. 36) and PCB printout (g. 37) can be seen in the appendix. The board includes 4 AD8302phase and gain detectors, each with 2 female SMA connectors for the RF input signals and 4female MCX connectors, 2 for the gain and phase sensor signals, and 2 for gain set and phase set,the feedback signals. The gain and phase set function can be activated with a switch. Additionallythe measurement board hosts a ADP7104ARDZ-R7, a CMOS, low dropout linear regulator, whichprovides from 3.3 to 20V and up to 500mA [16]. The Linear Dropout (LDO) supplies the AD8302with a constant 5V output power.The board has been assembled and tested for the rst time as part of this thesis.

Calibration The newly-designed AD8302 measurement board requires a phase and gain cali-bration at the working frequency of 1GHz.

(a) Gain calibration (b) Phase calibration

Figure 23: AD8302 measurement board calibration

The calibration was executed in a setup with DG645, SG392, and a digital phase shifter. Theresults can be seen in gure 23. For both diagrams the blue graphs shows the detector output for


a gradual input increase in gain or phase. The red graph is a linear regression, whose data willbe used to make the conversion from measured voltage to dBm and degree values in all followingmeasurements. Due to its best linearity, the setup 0 dB/270 was determined as the working point.

Resolution According to the data sheet, the AD8302 has a gain resolution of 30mV/dB and aphase resolution of 10mV/deg. With help of linear regression, the gain and phase resolution forthe linear parts of the calibration curve are determined as:

Measurement board gain resolution: 28.7mV/dB

Measurement board phase resolution: 10mV/deg

The phase output ranges from 0 to 1.8V, and the gain output between ±0.6V. For the gain outputAC-coupling is chosen so 0 dBm input equals 0V output, therefore oscilloscope voltage measure-ments can be performed with a higher precision.

With an MDO3054 oscilloscope [17] with 8 bit resolution for a 40mV measurement range wereach:

Theoretical gain resolution: 0.0054 dB, 0.125% concerning output power (formula 5)

Theoretical phase resolution: 0.0156 deg

The noise level of the measurement will limit the resolution. The requirements on attop mea-surements are 0.1% stability for the amplitude and 0.05 deg for the phase. The standard deviationmeasurements will prot from the statistics of a high number of samples to be able to determinethe pulse stability in this range.

3.6.2 Measurement setup

Figure 24 shows the setup used for phase and amplitude measurements with the AD8302 measure-ment board and the Boonton4500B power meter.

Figure 24: Phase and amplitude stability measurement setup for AD8302 board and Boonton4500Bwith attenuation values


Attenuation In the process of conducting measurements, attenuation, reduction of the RF signalamplitude due to return and missmatch loss, is an important factor. While the stability measure-ments are relative measurements and the internal attenuations hardly matter, they are importantfor absolute measurements.Concerning the AD8302 relative measurement setup, the amplier and reference signal suer underattenuation caused by several components. Directly connected to the signal generator output is aMini Circuits ZAP-D-2 splitter, which splits the output signal in two parts and therefore atten-uates both signals by -3 dBm. The reference signal undergoes further attenuation by the cablesconnecting the splitter to the phase shifter, and the phase shifter to the measurement board plusthe shifter itself, whose attenuation is determined as phase-independent. The amplied signalis attenuated by cables connecting the splitter and the amplier and feeding the signal into theAD8302 board. An additional attenuation is necessary due to the fact that the amplier generatesan output signal with an amplitude of up to 60 dBm, but the AD8302 chip has an input range from-60 to 0 dBm. Therefore two attenuators, 10 and 20 dB, and a 30 dB coupler are used. A coupleris a passive RF device which couples out a dened part of the electromagnetic power to a coupledport for further use, while the rest gets dissipated in a high power RF load. The signal input in theAD8302 is attenuated with -0.3 dB by an RF cable. In opposition the Boonton 57318 peak powersensor is characterized by the manual [14] with a VSWR of 1.15. With the two formulas 10 and14, the mismatch loss MLSensor can be determined as -0.002 dBm. The dierence is considered inthe calculation.

The precise attenuation values and component descriptions can be found in gure 24. The at-tenuation is considered and subtracted out in the following calculations and plots.

3.6.3 Amplitude attop comparison AD8302 measurement board and power meter

To evaluate the measurement abilities and resolution of the AD8302 measurement board, it iscompared with the Boonton4500B power meter. This comparison is only possible for amplitudemeasurements because the power meter is not able to measure phase. With a minimal verticalscale of 0.1 dBm/div, 8 divisions, and an 8-bit resolution, the Boonton4500B has a theoreticalamplitude resolution of 0.0031 dB, which is similar to the AD8302 measurement board.

(a) -7.6 dBm input power (b) -12.2 dBm input power

Figure 25: AD8302 measurement board and Boonton4500B amplitude attop comparison


The amplitude measurement results for the AD8302 measurement board and the Boonton4500Bpower meter for an amplier input power of -7.6 dBm and -12.2 dBm each with the associated am-plitude standard deviation are displayed in gure 25. For both measurement setups and both inputpowers the output pulse shape is analogous, including a attop variation of 0.5 dBm with a localpeak at 45 µs. The oset between the AD8302 measurement and the peak power meter signal iscaused by internal attenuation.

The amplitude standard deviation of the AD8302 measurement board and Boonton4500B mea-surements for -7.6, -12.2 and -15 dBm input power can be seen in gure 26.

Figure 26: AD8302 and Boonton4500B amplitude stability comparison

Both measurement devices show a standard deviation of 0.4% for -15 dBm input power. Whilefor the AD8302 measurement board this is the lowest value achieved, the Boonton4500B reaches itshighest stability for -7.6 and -12.2 dBm input with around 0.1% std. The gain/phase measurementboard's lowest standard deviation is a factor 4 higher than the one of the Boonton4500B, despitethe fact that they have a similar theoretical resolution.

3.6.4 Amplitude attop comparison AD8302 measurement board and AD8302-EVALZ

By reason that the required RF amplitude stability of 0.1% can by far not be achieved with theAD8302 measurement board, the AD8302-EVALZ, an AD8302 evaluation board, designed by Ana-log Devices, is analyzed as an alternative. Thanks to using the same chip, the AD8302-EVALZ hasthe same theoretical resolution as the measurement board: 0.1254% concerning gain and 0.0156deg phase.


(a) -7.6 dBm input power (b) -12.2 dBm input power

Figure 27: AD8302 measurement board and AD8302-EVALZ amplitude attop comparison

In the same approach as in section 3.6.3 the amplitude standard deviation is investigated andcompared for the AD8302-EVALZ and measurement board (g. 27).The RF pulses, although measured in the same setup with the same amplier output level and theidentical calibration data, have an oset of 0.4 - 0.6 dB, which can only be explained by a higherinternal attenuation caused by the connectors or the layout of the AD8302-EVALZ. Internal at-tenuation dierences do not cause major problems in our measurements since the focus is on themeasurement repeatability.

The overview of the amplitude standard deviation over the pulse length for dierent input powersis presented in gure 28.

Figure 28: AD8302 meas and EVALZ amplitude stability comparison

Because the formerly used input power levels could not achieve an amplitude lower than 0.1%,the power level -10.6 dBm, which equal 275W, was added. The evaluation board shows similarattop stability than the power meter. It outperforms the measurement board for every inputpower, reducing the lowest amplitude standard deviation value by a factor of 4. With a standard


deviation under 0.1% for -10.6 dBm input power, it meets the requirements for the system to pro-vide the CLIC drive beam klystron with a stable and reliable pulsed RF signal.

In the interest of further investigation of the cause of the discrepancy of amplitude standarddeviation between the AD8302 evaluation and measurement board, attop measurements withoutan input signal are undertaken. The amplitude standard deviation for both boards for no inputsignals, which means the signal generator and amplier turned o, for open connectors, the RFconnectors on the AD8302 board are left open, and for -10.6 dBm input as reference value, aredisplayed in gure 29. The standard deviations are noted in absolute values. As noticeable in thediagram both boards show each for the two setups `no input' and `open connectors' no signicantdierence in the attop stability. What is perceivable is the fact that the standard deviation ofthe amplitude measurement without input for the AD8302 evaluation board is only about half thevalue of the one of the measurement board. We conclude that the AD8302 evaluation board hasa lower amplitude noise compared to the measurement board designed at CERN.

Thanks to the setup of the CLIC drive beam acceleration structure only a part of the frequencyspectrum is relevant. Noise with a frequency lower than 10 kHz can be counterbalanced by l-ters and a feedback loop, while noise higher than 4MHz is not signicant due to the accelerationstructure, which averages the energy of the particles over a time of 240 ns. Therefore an FFT iscalculated with the `no input' case measurement data in the range of 10 kHz to 4MHz. The signallength is 7471 and the sampling frequency 49.8MHz. The results can be seen in gure 30.

Figure 29: AD8302 measurement board andAD8302-EVALZ amplitude noise std

Figure 30: AD8302 measurement boardand AD8302-EVALZ amplitude noise, 10 kHz-4MHz

The FFT shows a signicantly higher noise for the measurement board. Especially in the rangeof 2-3MHz the measurement board value exceeds the AD8302-EVALZ's one by a factor two.

3.6.5 Phase attop comparison AD8302 measurement board and AD8302-EVALZ

To determine the dierences in phase attop characteristics for the AD8302 measurement andevaluation board, the same approach as for the amplitude comparison (chapter 3.6.4) is chosen.By reason of the Boonton4500B peak power meter lacking a possibility to measure phase, thecomparison is limited to the two AD8302 boards.


The phase level of 270 degree is set manually and chosen because the AD8302 performs at itwith its highest resolution (compare phase calibration, g. 23b). The phase attop measurementsdisplay the same pulse shape for both readout systems with a total attop variation of 1 degree.

Figure 31: AD8302 meas and EVALZ phaseattop comparison for -10.6 dBm input power

Figure 32: AD8302 meas and EVALZ phasestability comparison

For -10.6 dBm input power, the level with the best amplitude attop stability, the evaluationboard shows a signicantly lower standard deviation than the measurement board. The osetoccurring in the phase attop measurements (see g. 31) is caused by phase shift inaccuracies dueto the manual phase setting. For further comparison the standard deviation values for the powerlevels -7.6, -10.6, -12.2 and -15 dBm are displayed in gure 32. The AD8302-EVALZ' phase attopstandard deviation shows a narrow range from 0.03-0.05 degree, fullling the requirement of 0.05deg phase stability for every input power. For the measurement board the lowest stability of 0.15deg is achieved for -15 dBm input power, exceeding the requirement by a factor 3.

Simultaneous to the amplitude attop measurement approach, the phase standard deviation forno RF input and open connectors is measured (see g.33). While the `no input' phase stability isaround 0.3% for both boards, the standard deviation for the case `open connector' of the measure-ment board is 2.5 times the one of the AD8302-EVALZ. The measurement board is very sensitiveto external phase uctuations, a fact which can inuence measurement results signicantly. Itmight be reasonably assumed that damaged or loose connectors, bent or awed cables, and signalgenerator and amplier inaccuracies have a heavier impact on the phase stability for the AD8302measurement board.


Figure 33: AD8302 measurement boardand AD8302-EVALZ phase noise std

Figure 34: AD8302 measurement board andAD8302-EVALZ phase noise, 10 kHz-4MHz

For a closer insight into the noise behavior an FFT, with a signal length of 7471 and a samplingfrequency of 49.8MHz, is calculated for the `no input' measurements (g. 34). The results conrmthe observation that the noise for the measurement board exceeds the evaluation board's.


3.6.6 Bit noise

One further possible limitation inuencing the amplitude attop standard deviation is bit noise.The term bit noise denes the inaccuracy caused by the fact that a digital measurement consistsof discrete values. If the measurement data plotted over the bit values represents only two or threebins, bit noise is existing. In case the distribution resembles a Gaussian shape, it is caused bynormal white noise.

Figure 35: AD8302-EVALZ bit noise over one trace for -10.6 dBm input power

Figure 35 shows the AD8302-EVALZ amplitude output voltage distribution for -10.6 dBm in-put power. The theoretical resolution for the measurement setup with AD8302 and MDO3054oscilloscope is, with 8-bit over 40mV range, 0.156mV. This value ts the distance between twoadjacent bins in the bit noise spectrum. The spectrum has a total width of 26.4mV because the out-put power does not stay at one constant level, but shows a attop variation of 0.5 dB (see g. 27a).

With a theoretical amplitude resolution of 0.1254% and a measurement requirement of 0.1%amplitude stability, a precise measurement should not be possible and bit noise should occur. Asthe attop stability is calculated with the help of standard deviation calculations over 100 measure-ments, a ten times more precise statistical conclusion is possible. Therefore the setup characterizedin this thesis is capable of measurements precise enough to certainly determine the stability in therequired range.


3.6.7 AD8302-EVALZ and measurement board layout comparison

Investigating the reasons for the performance dierences of the AD8302 evaluation and measure-ment board, the layout has to be considered. The schematics (gure 36) and PCB printout (37)of the measurement board and the schematics of the AD8302-EVALZ (gure 38) can be found inthe appendix.

The input stage of the measurement board resembles the one of the by Analog Devices designedevaluation board. In both cases the same dimensions are used for the input AC-coupling capacitors(C3 and C6, naming always refers to the AD8302 measurement board schematics) and capacitorsfor the oset feedback(C4 and C5), which determine the input lter corner frequency.The AD8302 chip requires a voltage supply of 2.7 to 5.5V. It was identied to work best in oursetup with 5V. Therefore the evaluation board is running on 5V, while the measurement board isfed with 15V, which is downscaled to 5V by an ADP7104ARDZ-R7 linear regulator. The maindierence in the power supply setup is that the evaluation board is equipped with a 0Ω decouplingresistor, a so-called zero-ohm link, additionally to the 100 nF decoupling capacitor (C19). Thiszero-ohm link is likely placed due to the fact that the PCB layout makes it easier to mount thisSMD then a wire or a jumper, which requires a dierent automated equipment.In the output stage the dierences between the two boards are more recognizable. An alternativeswitch layout for the gain and phase set function was chosen on the measurement board, althoughthis fact barely inuences noise levels. Rather likely to have an eect is the lack of snubbingresistors on the measurement board. Snubbing resistors are 0Ω resistors that operate as loads inresonant circuits to change the ringing frequency and reduce noise. Often the snubbing resistor iscombined with a small capacity, working as a buer to prevent ringing and slow down voltage riseat the edge of the control signal. The AD8302-EVALZ is equipped with 0Ω snubbing resistors inbetween the AD8302 gain and phase pins and related board connectors. A further investigation ofthis cause, including the adding of snubbing resistors on the AD8302 measurement board wouldbe necessary to identify and analyse the boards' performance.


4 Measurement conclusion and setup recommendation

The thesis outlines the work which was undertaken to generate and pre-amplify the pulsed RFklystron input signal for the CLIC drive beam accelerator. As a result of all the carried out mea-surements and analyses, a clear recommendation can be given concerning device choice and setupas well as power level and phase setting.

The main aspect in the quality evaluation of the pulsed RF signal is repeatability and stabil-ity. Fluctuations in amplitude and phase can be counteracted in the later process as long as theyoccur predictably. The ocially dened requirements, which can be seen in detail in chapter 3.1,are a pulsed RF signal with 1GHz carrier frequency, 150µs pulse length and 50Hz repetition ratewith a power level after the amplier of at least 200W, a pulse amplitude stability of 0.1%, andpulse phase stability of 0.05 degree.

The use of the delay generator DG645, which is serving as the pulse modulation trigger andamplier TTL control signal source, is advisable thanks to its short rise and fall time of 0.7 and2.2 ns (section 3.3.2). The signal generator produces, as dened by the project requirements, apulsed RF signal with a pulse length of 150 µs, a repetition rate of 50Hz, externally pulsed bythe DG645, and a carrier frequency of 1GHz. The Stanford SG392 showed in comparison to theRohde&Schwarz SMB100A a lower amplitude attop standard deviation before (g. 13) as wellas after the amplier (g. 22). Concerning phase noise, the SMB100A had a lower noise levelwhile showing a higher amount and more signicant spurs. Therefore, the SG392 was chosen assignal generator. Gain and output power characterization was executed for the Microwave AmpsL-Band 400W pulsed power amplier AM10 (g. 17), which proved to be able to provide therequired output power and stability. Following a comparison of the AD8302 measurement boardand a Boonton4500B peak power meter, and the observation of the measurement board criticallyunderperforming, the by Analog Devices designed evaluation board AD8302-EVALZ was addedto the comparison. It showed a signicantly better stability concerning attop standard deviationafter the amplier, as well for amplitude (g. 28) as for phase (g. 32). This eect could betracked down to higher noise for the measurement board (gain: g. 30, phase: g. 34). Onepossible cause is the lack of 0Ω snubbing resistors in the AD8302 measurement board output stage(see schematics in g. 36).

To meet the amplitude and phase attop stability requirements the amplier input power is of bigsignicance. The amplier output signal phase attop is barely inuenced by the power level (g.32), as opposed to the amplitude attop standard deviation (g. 28), which reaches a minimumvalue for -10.6 dBm amplier input power. To facilitate an ideal phase resolution, the phase shift isset to either 90 or 270 degree with the help of a phase shifter (compare AD8302 phase calibration,g. 23b). The AD8302 measurement is reliable in a range of ±15 dBm. The described setup hasa theoretical measurement resolution of 0.1254% concerning the amplitude and 0.0156 degree forthe phase.

Measuring an amplitude stability of 0.1% and a phase stability of 0.05 degree imposes high de-mands on the measurement setup. The theoretical amplitude resolution of the system used in thisthesis is slightly below the necessary standard deviation. But statistical behavior of the standarddeviation calculated with 100 measurements allows to gain eectively a factor 10 in resolution andtherewith to measure better than the required stability. For future work it is recommended to use


a more precise measurement setup since the AD8302 board reached its performance limits.

The recommended setup comprises the delay generator Stanford DG645, the signal generatorStanford SG392, the pulsed power amplier Microwave Amps AM10, the AD8302-EVALZ phaseand gain measurement board, and the Tektronix MDO3054 oscilloscope. The system generates,with an amplier input power of -10.6 dBm, a pulsed RF amplier output signal with a power of275W, a mean amplitude attop standard deviation of 0.092%, and a mean phase attop standarddeviation of 0.042 degree. It is recommended to attenuate the 275W output power on the highpower side to the required 200W input power of the klystron.

The setup fullls all the imposed requirements and can be used to generate and pre-amplify theRF klystron input signal of the CLIC drive beam.


List of Figures

1 Artist view of the Compact Linear Collider8 . . . . . . . . . . . . . . . . . . . . . . 12 CLIC layout at 3TeV9 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 CLIC two-beam scheme10 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 CLIC Drive Beam injector11 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 Radio frequency spectrum12 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56 Phase and amplitude stability measurement hardware setup . . . . . . . . . . . . . 97 Delay generator DG645 timing overview . . . . . . . . . . . . . . . . . . . . . . . . 108 Delay generator DG645 edges characterization . . . . . . . . . . . . . . . . . . . . . 119 Trigger and RF signal generator outpu signal modulation timing . . . . . . . . . . . 1210 SG392 output for 1GHz

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1211 SMB100A attop output power comparison . . . . . . . . . . . . . . . . . . . . . . 1312 SG392 attop output power comparison . . . . . . . . . . . . . . . . . . . . . . . . 1413 SG392 and SMB100A attop stability comparison . . . . . . . . . . . . . . . . . . . 1514 Output spectrum of a typical RF oscillator13 . . . . . . . . . . . . . . . . . . . . . . 1615 SG392 and R&S SMB100A phase noise measurement, correlation 10, averaging 8 . . 1716 Pulsed power amplier AM10 bandwidth . . . . . . . . . . . . . . . . . . . . . . . . 1917 Signal generator SG392 output, power amplier AM10 output and AM10 gain . . . 2118 Absolute linear error of signal generator SG392 output over signal generator input . 2219 Signal generator SG392 (blue) and power amplier AM10 output pulse (green) . . . 2320 Power amplier AM10 output signal rise time . . . . . . . . . . . . . . . . . . . . . 2321 Power amplier AM10 output power attop comparison with SMB100A input . . . 2422 Power amplier AM10 output power stability comparison for SG392 and SMB100A

input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2423 AD8302 measurement board calibration . . . . . . . . . . . . . . . . . . . . . . . . . 2524 Phase and amplitude stability measurement setup for AD8302 board and Boon-

ton4500B with attenuation values . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2625 AD8302 measurement board and Boonton4500B amplitude attop comparison . . . 2726 AD8302 and Boonton4500B amplitude stability comparison . . . . . . . . . . . . . . 2827 AD8302 measurement board and AD8302-EVALZ amplitude attop comparison . . 2928 AD8302 meas and EVALZ amplitude stability comparison . . . . . . . . . . . . . . 2929 AD8302 measurement board and AD8302-EVALZ amplitude noise std . . . . . . . 3030 AD8302 measurement board and AD8302-EVALZ amplitude noise, 10 kHz-4MHz . 3031 AD8302 meas and EVALZ phase attop comparison for -10.6 dBm input power . . 3132 AD8302 meas and EVALZ phase stability comparison . . . . . . . . . . . . . . . . . 3133 AD8302 measurement board and AD8302-EVALZ phase noise std . . . . . . . . . . 3234 AD8302 measurement board and AD8302-EVALZ phase noise, 10 kHz-4MHz . . . . 3235 AD8302-EVALZ bit noise over one trace for -10.6 dBm input power . . . . . . . . . 3336 AD8302 measurement board schematics . . . . . . . . . . . . . . . . . . . . . . . . . 4037 AD8302 measurement board PCB printout . . . . . . . . . . . . . . . . . . . . . . . 4138 AD8302 evaluation board PCB schematics; Source: AD8302 Data Sheet [15], page

21, gure 14 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42


References

[1] The history of CERN, accessed: 13.10.2016,URL: http://timeline.web.cern.ch/timelines/The-history-of-CERN

[2] CERN: A Multi-TeV linear collider based on CLIC technology, 2012.

[3] M. Stoehr: RF Basics, accessed:11.11.2016,URL: http://pdfserv.maximintegrated.com/en/an/AN5300.pdf

[4] Electronic Communications Committee: The European table of frequency allocations andapplications in the frequency range 8.3 kHz to 3000 GHz (ECA table), June 2016.

[5] D. Coleman & D. Westcott: Certied Wireless Network Administrator Study Guide, 2006.

[6] GPIB / IEEE 488 Tutorial, accessed: 30.11.2016,URL: http://www.radio-electronics.com/info/t_and_m/gpib/ieee488-basics-tutorial.php

[7] Stanford Research Systems: DG645 Digital Delay Generator, User Manual, 2008.

[8] Stanford Research Systems: SG390 Series RF Signal Generators, User Manual, 2013.

[9] Rohde & Schwarz: R&S SMB100A RF and Microwave Signal Generator Specications, Ver-sion 09.00, July 2016.

[10] D. Pozar: Microwave Engineering, 3rd edition, 2005, ISBN: 978-0470631553, cited on chapter12.3, p.594-595.

[11] Agilent Technologies: Phase Nosie Measurement Methods and Techniques, 2012.

[12] Keysight Technologies: E5052B Signal Source Analyzer, Data sheet, 2016.

[13] Microwave Ampliers Ltd: L-Band 400w Pulsed Power Amplier Model AM10.

[14] Boonton: Boonton power Sensor Manual, April 2011.

[15] Analog Devices: LF-2.7 GHz RF/IF Gain and Phase Detector, Data Sheet, 2002.

[16] Analog Devices: 20 V 500 mA Low Noise CMOS LDO ADP7104 Data Sheet, 2015.

[17] Tektronix: MDO3000 Series Datasheet, 2016.


Appendix


Figure 36: AD8302 measurement board schematics


Figure 37: AD8302 measurement board PCB printout


Figure 38: AD8302 evaluation board PCB schematics; Source: AD8302 Data Sheet [15], page 21,gure 14


b achelor s t hesis - cernorf example, the 100 vk supply voltage of the klystron needs a 10 5...

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