chapter - 4 development of a leo beacon receiver...
TRANSCRIPT
52
Chapter - 4
DEVELOPMENT OF A LEO BEACON RECEIVER SYSTEM
FOR TEC MEASUREMENT AND SIMULATION OF AN
ORTHOGONAL CARRIER SPREAD SPECTRUM SYSTEM
This chapter details the hardware development of a LEO beacon receiver for
TEC measurement completed as part of this work. Simulation studies attempted on
an orthogonal carrier spread spectrum system to address the initial phase ambiguity
of the above hardware developed for TEC measurement is also explained.
4.1 Introduction
The ground-based reception of radio beacon signals transmitted from LEOS is a
historical method to measure the total electron content (TEC) of the ionosphere.
Observations of this kind were initiated right from the beginning of the space era as
reported by Aitchison and Weekes [1959] and Garriott and Little [1960]. The
principle of the experiment is based on the frequency dependence of the refractive
index of radio waves in the ionospheric plasma. The most commonly used
frequencies are 150 and 400 MHz, in a ratio of 3:8, generated from a common
oscillator signal at 50 MHz. The most common beacon satellite constellation is the
polar-orbiting Navy Navigation Satellite System (NNSS), detailed by Newton
[1967], and renamed later as Navy Ionospheric Monitoring System (NIMS) as
mentioned in Leitinger et al [1984]. Beacon receivers developed in the past have
been mostly analog receivers for 150 and 400 MHz signals where the phase
relationship between the two signals is detected by an analog circuit and the
resultant phase values digitized at tens of Hz.
In order to study the ionosphere using existing LEO beacon satellites, an Indian
Coherent Radio Beacon Experiment (CRABEX) project has been conceived by SPL.
The project has two operational phases. The first one dealt with reception of LEO
satellite beacons while the second one involved launching a GSAT beacon
Chapter-4
53
transmitter onboard Indian GSAT-2 mission. The details of the LEOS phase are
presented in this chapter. The major shortcoming in such a system is also mentioned
along with a simulation study using spread spectrum techniques to address this.
4.2 Genesis of LEO beacon satellites
The first beacon satellite system to be used for ionospheric studies is of Low Earth
Orbiting type. These satellites transmit two coherent frequencies, one in the VHF
band and the other in the UHF band. These satellites were initially launched for
navigation purposes. As mentioned above, the first satellite navigation system was
Transit, a system deployed by the US military in the 1960s. The Johns Hopkins APL
Technical Digest [1998] explains that Transit 4A and 4B were the first satellites to
use operational frequencies of 150 MHz and 400 MHz. The operational series of
Transit satellites started with Transit 5A-1 and are more commonly referred to as
“Oscar” series. On Jan. 1, 1997, the remaining Transit system constellation
(consisting of six Oscars in three orbital planes at altitudes (apogees) of about 1100
km) became NIMS with a new application, namely to utilize the Transit system
resources for computerized ionospheric tomography (CIT). In this setup, the NIMS
satellites are being used by ground collection sites as dual-frequency beacons to
determine the free electron profile of the ionosphere as reported by Robert J
Danchik [1998]. Almost along the same time, the Russians also launched their
military navigation satellites, called Parus. Tsikada was a complementary civilian
version of the Parus military naval navigation satellite system for the Soviet
Merchant Marine and Academy of Sciences. These satellites transmitted Doppler-
shifted VHF-UHF transmissions at around 150 and 400 MHz, as reported in
SPACEWARN bulletin.
The OSCAR series was later on followed by several other beacon satellites like
GEOSAT, GEOSAT Follow On (GFO), and Picosat. Later on, during 2006, a series
of six beacon satellites called Formosat, as a Taiwanese-American collaboration,
were launched. The latest in this series is the Indian Radio Beacon for Ionospheric
Tomography (RaBIT) payload developed by SPL and VSSC, launched onboard
Indian YOUTHSAT mission in 2011.
Chapter-4
54
4.2.1 Beacon satellites and their frequencies
The series of NIMS satellites have the VHF and UHF transmitters designed with the
output power of 1.25 W at 150 MHz and 1.75 W at 400 MHz according to
SPACEWARN bulletin. These satellite frequencies generated onboard were
extremely stable due to the very good short-term stability of the caesium oscillators
onboard, starting with a stability of 5 parts in 1010 to about 2 parts in 1015. An
incremental phase shifter moved the frequency any small desired amount with very
great precision in response to a command from the ground, as is described by Robert
J Danchik [1998] so as to avoid frequency conflict between satellites with
consistently overlapping orbits. The Russian satellites transmitted a higher power
and had fixed frequencies. The Table below shows a comprehensive list of the
existing active coherent beacon satellites presently in orbit, along with their
transmitting frequencies.
Table 4.1 List of active LEO beacon satellites
Satellite Satellite
ID Frequencies (MHz)
Launch year
Orbit inclination (°)
Cosmos 2398 27818 149.910, 399.760 1997
83
Cosmos 2378 26818 149.940, 399.840 2001
Cosmos 2414 28521 149.940, 399.840 2005
Cosmos 2336 24677 149.970, 399.840 1996
C/NOFS 27436 150.012, 400.032 2008 13
Oscar 23 19070 149.988, 399.968
1993,1998 90 Oscar 32 19071 149.988, 399.968
Oscar 25 19419 149.988, 399.968
RADCAL 22698 150.012, 400.032 1993 90
Formosat 3A to
3F
29047 to
29051
150.012, 400.032,
1066.752 2006 72
Youthsat 37388 150.000,400.000 2011 99
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Chapter-4
56
4.4 The Coherent Radio Beacon Experiment (CRABEX) national
project - Phase I
Though satellite radio beacons have been used for ionospheric observations for more
than three decades now, there have been very few tomographic experiments from
the equatorial and low-latitude region. The Coherent Radio Beacon Experiment
(CRABEX) project is conceived by Space Physics Laboratory of Vikram Sarabhai
Space Centre, Trivandrum with this in view, involving various national institutes
and Universities spread across India. In the first phase called the LEOS phase, a
suitable receiver system for LEO beacon reception is designed and developed. For
ionospheric tomographic studies, this receiver system is installed at the participating
national institutes and Universities. The data from this chain is being used for ray
tomographic studies as stated by Smitha V Thampi et al [2005].
4.4.1 Scientific objectives of the CRABEX LEOS
The beacon signals from the existing LEO satellites were used to address the
following major scientific problems -
Characterize the low-latitude ionosphere for its continuous variability
and turbulence and evolve tomographic images of the ionosphere using
Computerized Ionospheric Tomography (CIT) techniques.
Better coverage of events like magnetic storms and Equatorial Spread F.
Generate a good database which will be able to represent and model the
unique features of the equatorial ionosphere.
In order to address these objectives, it is necessary that a chain of similar ground
receiving stations be set up from Trivandrum towards the northern latitudes.
4.4.2 Design requirements of the ground receiver
The design of a ground receiver for coherent LEO beacon should follow certain
guidelines to meet the science objectives and satisfy the following requirements for
ionospheric tomography.
Chapter-4
57
The receiver should be able to track any of the “Active” satellites present
in Table 4.1, where it can be seen that all these existing satellites
maintain the same frequency ratio of 3:8. Broadly these can be
categorized to represent the following series of satellites.
Tsikada series, RaBIT : 0 ppm offset (150.000, 400.000 MHz)
NIMS series (Oscars)
Operational : -80 ppm offset (149.988, 399.968 MHz)
Maintenance offset : -145 ppm offset (149.979, 399.944 MHz)
Picosat and Formosat : +80 ppm offset (150.012, 400.032 MHz)
The receiver design should be simple enough to be easily duplicated.
It should be coherent, so that any strong extraneous noise in the bands of
interest also will not make the system unlock, when the system is
tracking the satellite.
The antennae should be located in a preferably RF noise free
environment.
The cable lengths from the antenna to the front end should be phase
matched at both the frequencies.
The receiver should take care of the Doppler shift of the moving satellite.
It should start acquisition automatically as soon as the system locks to
both the frequencies, with preferably an online display of the signals
being acquired.
The DAQ should be able to have simultaneous sampling for both I and Q
outputs of phase differences as well as any other amplitude channels.
The sampling rate of DAQ should be variable up to say 1 KHz.
The resolution should be at least 12 bit for ±10V input.
Chapter-4
58
In order to make use of the data for generation of tomograms, it is required that
multiple receiver stations located along the plane of satellite orbit should observe the
same satellite pass and record the data as told by Sutton and Na [1995]. As most of
the existing satellites are polar orbiting, the stations are chosen along the same
longitude (+/- 2 degrees) so that they can monitor the same satellite simultaneously.
In order to cover India as a whole, the receiver locations are identified at national
institutes situated at a distance of 300 to 500 km; as the Total Electron Content
(TEC) calculated from these different stations give a consolidated picture of the
ionosphere through tomographic constructions.
4.5 Principle of operation of ground receiver
The objective of this system is to measure the ionospheric electron density along the
line of sight integral for calculating the relative total electron content (TEC).
e (4.1)
The measurement of relative TEC is accomplished by measuring the difference in
Doppler shift between the two carrier frequencies (150 and 400 MHz). The Doppler
can be considered to be made up of two components: the motion Doppler and the
contribution due to the ionosphere. So we can write
∆f=f μs . . . t (4.2)
Taking the difference
3∆ 400 8∆ 150 (4.3)
This removes the motion dependent effect of the Doppler shift, leaving only the
contribution due to the ionospheric effects. Thus we need a receiver system which is
able to multiply the instantaneous phase at 400 MHz by 3 and 150 MHz by 8 and
use a phase detector to remove Doppler. The output of this phase detector will
therefore be the instantaneous phase difference between the two frequencies
observed by the receiver and which is attributed as due to ionosphere.
Chapter-4
59
4.6 Link budget calculations
Using the standard equations for satellite-to-ground system propagation of radio
signals, the link budget for the required frequency of 150 and 400 MHz for LEO
system is tabulated as given below. Here, the satellite is assumed to be at a height of
~ 900 km (LEO orbit height), so that the slant range from a receiver location when
the satellite is at 10° elevation from the horizon is ~ 2400 km. It is a known fact that
the satellite is farthest when it is just above the horizon and will be nearer to the
receiver as it approaches overhead. For the present receiver system to sense the
satellite as soon as it rises above the horizon, we assume that the receiver sensitivity
should be designed to track and lock to the signal when it is at an elevation of 10°.
Table 4.2 Link budget for LEOS beacon system
Frequency(MHz) 150 400
Tx Power O/P (mW) minimum 1000 (+30 dBm) 1000 (+30 dBm)
Tx Antenna Gain (dBc) 0 0
EIRP (dBm) +30 +30
Free space loss for 10° elevation -143 -152
Rx Antenna Gain (dB) 0 0
Polarisation Loss (dB) -3 -3
Other losses (dB) -2 -2
Signal Power at Rx I/p (dBm) -118 -127
Rx Noise Temp (°K) 1000 1000
Antenna Noise Temp (°K) 3000 1000
System Noise Temp (dB) 36.02 34.01
Boltzmann Constant (dBm/Hz/°K) -198.6 -198.6
Rx Noise (dBm/Hz) -162.6 -165.6
Rx Bandwidth (dBHz) 30 (1 KHz) 30 (1 KHz)
Rx Noise Power (dBm) -132.6 -135.6
S/N Ratio (dB) 14.6 8.6
Chapter-4
60
i.e. In the present scenario, when 400 MHz is considered as the reference channel as
it suffers lesser change due to ionosphere, the receiver needs a sensitivity of better
than -127dBm and an SNR of >7 dB to lock to the satellite signal, when it is above
10° elevation.
4.7 Specifications of receiver
The next step is to finalise the receiver specifications by satisfying the link
calculations and scientific requirements. Thus the specifications are worked out as
shown in the Table 4.3 below.
Table 4.3 Receiver specifications
Antenna
Type of antenna Crossed dipole/Microstrip patch
Antenna gain ≥ 4 dB for all the frequencies
Antenna Beam width Better than 110°
Receiver system
Type of Receiver Dual band PLL type Doppler tracking
Number of inputs Two (1VHF, 1UHF) (Coherent beacons)
Frequency of operation
Input I (400 MHz + 40 KHz/-60KHz)
Input II (150MHz ± 15 KHz)
Number of frequency bands 4
Signal sensitivity Better than -127dBm for both channels
IF frequencies 10.7MHz and 4.0125MHz
Bandwidth 1 KHz for both frequencies
Outputs Phase compared I and Q outputs
Amplitude of the two signals (0-2.5V max)
Data acquisition system
Type of DAQ SC-2040 simultaneous sampling card and
PCI-6035E data acquisition card
Software Data acquisition program in LabVIEW
Chapter-4
61
4.8 Design and development of receiver
The receiver is designed as a coherent detection system which takes both VHF and
UHF beacon satellite signals simultaneously. It is already known that the ionosphere
affects VHF more than UHF propagation so that in all practical cases, the changes in
UHF channel can be neglected. In this situation, the instantaneous phase difference
between the two received signals is effectively the electron density fluctuations/
changes in the ionosphere detected by the receiver, with the VHF channel as the data
channel and UHF as the reference channel. The antenna is specified to have a broad
beam width to eliminate the need for steering during a satellite pass and also
simplify the system. The receiver system is designed to have an outdoor unit and an
indoor unit, in addition to PC based data acquisition system. As the name implies,
the outdoor unit is kept very near to the antenna and the indoor unit and PC is inside
the lab.
Before conceptualizing the receiver blocks, it is apposite to address the point that
this receiver system is to be duplicated to be kept at different locations across the
country for ionospheric tomography studies as mentioned in the earlier sections.
This also indicates that these stations should be time-synchronized, which calls for
inclusion of a single frequency GPS receiver in the present case.
Thus a simple block diagram of the present receiver system with all the major
subsystems included is shown in figure 4.2. Each subsystem design is also detailed
below.
4.8
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62
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Chapter-4
63
space wavelength. These antennas are also relatively inexpensive to manufacture
and design because of the simple 2-dimensional physical geometry. They are usually
employed at UHF and higher frequencies because the size of the antenna is directly
related to resonance frequency. A single patch antenna provides a maximum
directive gain of around 4-6 dB at UHF.
An advantage inherent to patch antennas is the ability to have polarization diversity.
Patch antennas can easily be designed to have Vertical, Horizontal, Right Hand
Circular (RHCP) or Left Hand Circular (LHCP) Polarizations, using multiple feed
points, or a single feed point with asymmetric patch structures. This unique property
allows patch antennas to be used in many areas types of communications links that
may have varied requirements.
The patch antenna design is based on the transmission line model described by
Gupta and Benalla [1986] and Constantine Balanis [1982]. The use of air as
dielectric tunes the patch dimensions at a slightly lesser wavelength than λ0/2 i.e.,
0.93 λ0/2, where λ0 is the free space wavelength of the resonant frequency. The
height of the dielectric substrate h is chosen as 30 mm for both 150 and 400 MHz. A
typical patch antenna with substrate and the various dimensions marked is shown in
figure 4.3.
Figure 4.3 Microstrip rectangular patch antenna dimensions
The details of the steps in the dimensional calculations are explained below.
Chapter-4
64
Step 1: Calculation of the Width (W ): The width of the Microstrip patch antenna
is given by
(4.4)
where εr represents the relative permittivity of the medium
Step 2: Calculation of Effective dielectric constant (ε reff ):
1 12 (4.5)
Step 3: Calculation of the Effective length (Leff ) (Electrical length):
(4.6)
Step 4: Calculation of the length extension (∆L ):
Δ 0.412. .
. . (4.7)
Step 5: Calculation of actual length of patch ( L ) (Mechanical dimension):
2Δ (4.8)
Step 6: Calculation of the ground plane dimensions (Lg and Wg ):
The transmission line model is applicable to infinite ground planes only. However,
for practical considerations, it is essential to have a finite ground plane. It has been
reported by Punit Shantilal Nakar [2004] that similar results for finite and infinite
ground plane can be obtained if the size of the ground plane is greater than the patch
dimensions by approximately six times the substrate thickness all around the
periphery. Hence, for this design, the ground plane dimensions would be given as:
~6 (4.9)
~6 (4.10)
Step 7: Determination of feed point location (Xf , Yf ):
A coaxial probe type feed is to be used in this design. As shown in figure 4.3, the
centre of the patch is taken as the origin and the feed point location is given by the
Chapter-4
65
co-ordinates (Xf , Yf) from the origin. The feed point must be located at that point on
the patch, where the input impedance is 50 Ω for the resonant frequency. i.e., it
matches the location of this feed point as calculated from
Z Z cos π (4.11)
where Zp is the impedance at feed point P, Ze is the edge impedance of the patch at
the corner, l is the diagonal length and x is the distance of feed point from the
corner. This result is first verified by trial and error method also, by noting the return
loss at each point when the feed point is varied, and the one having the minimum
return loss is fixed as the feed point.
As LHCP type of antenna is preferred in the present case and the feed point falls
along one of the diagonals. The ground plane and patch are made of 15 mm thick
Aluminium sheet. A hollow spacer made of Teflon of height 30 mm and diameter 7
mm is kept at 15 mm from each of the corners of the ground plane and the top plate
is mounted on this. This dielectric material, Teflon with a relative permittivity εr of
2.1 also affects the antenna characteristics. Hence the antenna dimensions are further
trimmed down to include the effect due to the Teflon spacers also.
The table below gives the final dimensions of the patch antenna designed for both
400 MHz and 150 MHz, calculated according to the design equations and
constraints mentioned above.
Table 4.4 Microstrip Patch antenna dimensions
Frequency 400 MHz 150 MHz
Length 332 mm 910 mm
Breadth 316 mm 866 mm
Type of connector N type N type
Location of connector (along the
diagonal, from the centre of intersection
of diagonal)
94.2 mm 262 mm
Chapter-4
66
A variety of naturally occurring and man-made phenomenon exhibiting impulsive
noise behaviour affects the performance of these antennae. Systems performing in
the VHF and UHF range is plagued by man-made noise in metropolitan areas and
generally contaminated by galactic and solar noise as explained by Middleton
[1972]. Hence, the two antennae for 150 and 400 MHz need to be located at a
suitable site having lesser impulsive noise, preferably away from a public road.
Having selected a suitable site, the two antennae are installed with a minimum
distance of > 2λmax i.e. 4 meters between them to minimize their mutual interference.
Hence these are installed on top of the open terrace having a clear view of sky.
The various antenna characteristics like axial ratio and typical beam width are then
measured. These are tabulated in Table 4.5 below.
Table 4.5 Specifications of 400 MHz and 150 MHz antenna
Antenna Type Single feed microstrip air dielectric
Feed 50Ω coaxial
Frequency 400 MHz 150 MHz
Number of ports 1
Polarization LHCP
Axial Ratio 1.2 dB 1.2 dB
VSWR 1.5 1.5
3 dB beam width 60° 60°
3 dB band width 5 MHz 3 MHz
Gain 5.4 dB 5 dB
Efficiency 76% 70%
Ground plane dimensions (mm) 510 x 510 x 15 1100 x 1050 x 15
The signals received from the two antennae are routed to the outdoor unit using
short length cables which produce equal phase delay for the two beacon signals.
This means that the electrical length of both the cables should be same. It is
understood that electrical length is not the same as mechanical length and is
dependent on the dielectric material of the cable. Mathematically, for a standard N-
Chapter-4
67
type double shielded coaxial RG cable, the mechanical (M) and electrical (E)
lengths are related as,
0.87 (4.12)
The cable impedance for both cables are measured for both the frequencies using a
network analyzer, and trimmed accordingly. The cables leading from the antennae
are standard 13mm dia double shielded coaxial low loss cables of type
RG213/RG214, with a typical cable loss of < 0.3 dB for 4 metres. The interface
coaxial connectors used are of type N for ruggedness and ease of handling.
4.8.2 Outdoor unit
The block diagram of the outdoor unit is given in figure 4.4. The principal function
of this unit is to receive the signal at the closest point to the antenna. This helps to
ensure that there is no extra noise getting added to signals and thus deteriorate the
SNR.
Figure 4.4 Block schematic of outdoor unit
Chapter-4
68
The outdoor unit consists of a low noise amplifier, an image filter, a down converter
and an IF amplifier for both the frequencies. All the basic circuits are built as analog
systems. The LNA has a typical noise figure of 2.5dB for 100 to 400 MHz and a
gain of ~30 ±1 dB at both the frequencies. Since the required band width is much
less than 1 MHz, size restriction and affordable loss dictate the filter configuration
while the low IF frequency limits the available image rejection. The filters provide
image rejection of typically 45 dB and an insertion loss of 5 dB.
A buffer amplifier (B) with a typical gain of 15 dB is used to provide additional gain
in the front end. This is followed by a double balanced mixer (M) with a frequency
range of 1 MHz to 500 MHz and a conversion loss of 6 dB which is driven by the
local oscillator signal from the LO multiplier assembly. The output of the mixer is
10.7 MHz for the 400 MHz signal and 4.0125 MHz for the 150 MHz signal and is
amplified by a buffer amplifier (B), filtered and brought out through TNC
connectors. As a filter of high Q is needed to reduce the signal bandwidth without
incurring any signal loss, crystal filters (Xtal) having a 6dB bandwidth of 2.2 KHz is
used.
The LO multiplier assembly consists of a VCXO at 24.3365 MHz tunable over a
frequency range of ~ 100 KHz, with the control voltage varying from 0 to 5 V. The
output of VCXO is passed through a power splitter (PS) for the two channels and
routed through two multiplier chains of X 6 and X 16, followed by band pass filters.
The control voltage of the VCXO is sent from the indoor unit through the 4.0125
MHz IF output cable. The DC power to all the assemblies of the outdoor unit is fed
from the indoor unit via the 10.7 MHz IF cable.
4.8.3 Indoor unit
The indoor unit of the CRABEX receiver is the central processing unit for the
CRABEX Receiver System. The unit is housed in a standard 19’’ rack of 2U height.
It consists of the following principal sub assemblies as shown in figure 4.5.
The 1
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70
Thus the lock frequency of 1.3375MHz is the reference frequency used for phase
comparison. A sample of this locked output is also multiplied by 8 and fed to the
microcontroller assembly for IF frequency monitoring purposes in the front panel
and for generating control voltage for the VCXO in the outdoor unit. The PLL
assembly also puts out a lock detect flag which is fed to a comparator to generate a
flag for microcontroller and for the front panel LED and for monitoring with an
oscilloscope.
Since it is desired to have the In phase and Quadrature phase differences of the 150
MHz signal with respect to the 400 MHz signal, two phase detectors are employed.
The reference frequency for the two phase detectors are derived from a 3dB
quadrature hybrid into which the reference 1.3375 MHz signal is fed as one channel.
Likewise the data signal (1.3375 MHz IF from the 150 MHz) chain is split in phase
and applied to the other signal input port of the phase detector. As the phase
comparison is done here at 1.3375 MHz, the actual phase difference of the 150 MHz
signal with respect to the 400 MHz signal is 3 times the phase difference measured.
The microcontroller module is used to automate the receiver scanning and tracking
functions. It is programmed to scan the preset channels sequentially until phase lock
is detected. Once the system is locked to the satellite as indicated by the lock detect
flag of 10.7 MHz PLL, it enters an auto-tracking loop where the frequency of the
10.7 MHz IF is continuously monitored to change the control voltage of VCXO.
When the frequency changes below a preset value, the channel number is
incremented. The received signal strength from the input voltages is also converted
into dB and displayed as the RSSI value in the front panel.
4.8.4 PC based data acquisition unit
The data acquisition system consists of an 8 channel simultaneous sampling card
SC-2040 which is connected to a 12 bit, 200 KSps PCI multifunction DAQ card
PCI-6035E of M/s. National Instruments Inc. The data sampling card is fitted inside
the indoor unit of receiver. The outputs of the phase detector are connected to this
card along with the signal strength inputs. This has independent sample and hold
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71
circuit with instrumentation amplifier and programmable gain for each channel. The
output of each instrumentation amplifier is routed to the track-and-hold (T/H)
amplifier. In track mode, the outputs of the T/H amplifiers follow their inputs. In
hold mode, the T/H amplifier outputs simultaneously hold the signal levels constant.
The power and sampling pulse to SC-2040 is obtained through the PCI-6035E card
installed in the PC. As the card is to be configured for differential mode operation, a
channel resistance of 470Ω is put between each negative channel and card ground, to
feed the single ended signal according to the guidelines given in the product manual.
The analog outputs are read by the PCI-6035E card and digitized with a negligible
time skew (less than 50 ns) between channels.
4.8.5 Time synchronization with GPS unit
The above receiver system tracks the LEO beacon satellites which are in its tracking
range. As has been mentioned earlier, the main scientific objective of this project is
to obtain ionospheric tomograms with LEO satellite beacons. These beacon satellites
are orbiting at an altitude of 900 to 1000 km and with a speed of approximately 7
kmps. This indicates that a difference of 1 second between the ground receiver
stations can give an error of ~7 km when tomographic reconstruction is done, where
a grid size of 50 metre x 50 metre is assumed normally. Hence this requires a
method which can provide time synchronization of the order of milliseconds.
Now, it is a known fact that the PC time at each station would well differ, unless
they are regularly updated. As automatic operation of the system is planned and so
an automatic method of time synchronization is addressed. As there are more than 2-
3 satellites being tracked by any ground receiver and since their pass occurrence
instances have a day-to-day variability, it is not possible to schedule the time
synchronization automatically at a fixed slot every day.
This calls for a GPS based system wherein the timing data as obtained from a single
frequency GPS receiver is logged in during every satellite pass. This requires that
low cost, compact, rugged and readily available GPS systems can be used. These
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72
units are also easily programmable and interfaced to PC. They are also able to give
better than 10 milliseconds timing resolution during each acquisition.
After a detailed survey of the available GPS receivers in the market which are
technically suitable, Model GPS 25LVS of M/s. Garmin is chosen for this project.
This is a very compact, low cost, single frequency (L1 alone), 12 channel receiver.
The number of channels indicates the maximum number of satellites it can track
simultaneously. It has an active patch antenna with a 2m long cable. This receiver is
interfaced to PC through serial port. During the initial setup and installation, certain
parameters and the software need to be updated to the GPS receiver. The serial port
helps in both transmission and reception using RS 232 protocol. The mode of
communication is in asynchronous format at a baud rate of 9600 or lesser.
This single frequency GPS receiver receives the GPS signals and transmits them to
PC in NMEA (National Marine Electronics Association) standard, which was
defined initially as an electrical interface and data protocol for communications
between marine instrumentation. Under this standard, all characters used are
printable ASCII text (plus carriage return and line feed). NMEA-0183 data is sent at
4800 baud rate in the form of ‘sentences’ as explained in NMEA website.
Each sentence starts with a "$", a two letter "talker ID", a three letter "sentence ID",
followed by a number of data fields separated by commas, and terminated by an
optional checksum, and a carriage return/line feed. A sentence may contain up to 82
characters including the "$" and CR/LF wherein each character is treated as 8 bits or
1 byte. If data for a field is not available, the field is simply omitted, but the commas
that delimit it are still sent, with no space between them. Since some fields are of
variable width or may be omitted, the receiver locates desired data fields by
counting commas, rather than by character position within the sentence. This
property is made use of in extracting the required time and date information through
the in-house developed serial port software in LabVIEW. The optional checksum
field consists of a "*" and two hex digits representing the exclusive OR of all
characters between, but not including, the "$" and "*".
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73
The GPS unit is connected to the COM port 1 of the PC using its interface cable.
The zip files of the unit are copied onto hard disk and unzipped. The GPS unit is
turned on and the interface setup is set to "GARMIN". From the DOS prompt, the
directory is selected as the one in which files were unzipped and the update
program: “updatesw.exe” is run. The upload process takes approximately 3-10
minutes to complete. When the upload process is complete, the unit resets itself and
turns on. Now, the unit is programmed to receive standard NMEA-0183 sentences.
The program “gpscfg.exe” is run and the com port connection is ensured with the
selected baud rate. The approximate co-ordinates of the station are entered and
WGS-84 datum is chosen. A sub-menu indicates the choice of NMEA sentences that
the GPS unit is programmed to receive. The selection of sentence is made from this
and the program is run. In the main menu itself, the programmed sentence gets
immediately updated. The screenshots of the steps is shown below in figure 4.6.
Figure 4.6 Programming menus of GPS
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74
Now, the unit is connected to its antenna with the 2 metre cable. The NMEA
sentence suitable for this work is identified as ‘GPRMC’, whose format details is
listed below.
$GPRMC Sentence (Position and time)
Example (signal not acquired):
$GPRMC,235947.000,V,0000.0000,N,00000.0000,E,,,041299,,*1D
Example (signal acquired):
$GPRMC,092204.999,A,4250.5589,S,14718.5084,E,0.00,89.68,211200,,*25
Table 4.6 $GPRMC Sentence (position and time)
Field Example Comments
Sentence ID $GPRMC
UTC Time 092204.999 hhmmss.sss
Status A A = Valid, V = Invalid
Latitude 4250.5589 ddmm.mmmm
N/S Indicator S N = North, S = South
Longitude 14718.5084 dddmm.mmmm
E/W Indicator E E = East, W = West
Speed over ground 0.00 Knots
Course over ground 0.00 Degrees
UTC Date 211200 DDMMYY
Magnetic variation Degrees
Magnetic variation E = East, W = West
Checksum *25
Terminator CR/LF
This sentence sets the initial latitude/longitude. The position data will be updated
when position fixing begins. This sentence is transmitted every second, and the
duration each time is about 100 msec. This is almost in synchronism with a 1 PPS
output also provided in the receiver module. The in-house developed data
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75
acquisition software gets the time, latitude and longitude from the GPS receiver
through serial port, prior to recording every set of data from the satellite. The
software for both data acquisition and GPS acquisition is written in LabVIEW and is
explained in Section 4.10.
4.9 Testing and characterisation of the receiver system
Any system developed has to undergo a detailed test and characterisation before use
in the field for continuous operation. This involves testing of each subsystem
separately from the PC board level and then integrating into a full system. Before
starting the detailed test, the equipments are first cross calibrated and the cable
losses measured. For the tests mentioned herein, all the cables are ensured to have a
loss of ~0.49 dB at 400 MHz and ~0.33 dB at 150 MHz, and a stability of < 5mV at
400 MHz for the equipments.
The receiver is tested in two configurations. In the first configuration, ie, cable
mode, the receiver unit is connected directly to the signal generator output. Care is
taken to ensure that the output of signal generator is compatible to the input range
specified for the receiver. In the next configuration, ie, radiation mode, the receiver
system is connected to antenna ie, in the configuration needed for satellite reception.
The signal generator output is also connected to a low gain monopole antenna and a
higher signal level than used in the first configuration is transmitted during the tests.
4.9.1 Cable mode of testing for receiver
The test setup for the cable mode of testing is shown in figure 4.7.
Figure 4.7 Cable mode testing of CRABEX receiver with signal generator
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76
The receiver parameters measured are sensitivity, stability and phase correlation of
the receiver. The major details of the tests and its results are mentioned herein.
4.9.1.1 Sensitivity measurement
This test is done to find out the sensitivity of the receiver at 150 MHz and 400 MHz.
The test set up shown in figure 4.7 is used. The signal generator output is kept at a
nominal value of -100 dBm. Test is carried out for 400 MHz first, by first locking
the receiver at 10.7 MHz, and then reducing the signal level so that the system goes
out of lock. Now, the signal level is increased slowly, so that the system locks to the
signl generator. This value is noted as the sensitivity of 400 MHz. Similarly, the test
is done for 150 MHz, ensuring that the 400 MHz channel maintains lock throughout,
since in the actual scenario, data is valid only when 400 MHz is locked to the
satellite, as this is the reference channel and 150 MHz is the data channel. This
sensitivity is displayed as Relative Signal Strength Indicator (RSSI) in the front
panel and in both cases, the difference in the actual measured value and front panel
displayed value of RSSI provides the offset in the displayed levels. In these tests, the
receiver measured a sensitivity of -127 dBm for 400 MHz and -128 dBm for 150
MHz.
4.9.1.2 Stability measurement
This test is done to find the receiver clock stability. This test is done at the minimum
and maximum values of the allowable range of the receiver system. The I and Q
plots with the signal levels set at the maximum level of -93 dBm and the minimum
level of -124 dBm are taken separately each for a duration of 20 minutes. The time
duration of 20 minutes is considered as this is the maximum duration the receiver
would be seeing the satellite during a pass. The percentage error is each channel is
calculated and it is found that in all the cases, this is within the expected error of
10%, since an error of 10% would be giving an error of less than 0.1% in TEC
calculated.
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77
4.9.1.3 Phase difference measurement
This test is done to find how the receiver responds to variation in phase, induced
between the input signals. The dual channel signal generator used for these tests has
the provision to increment the phase of one frequency alone. Thus the output from
both channels of signal generator is first made coherent with an amplitude of -100
dBm so that the receiver maintains lock. Then the phase of 400 MHz is changed in
steps of 5° and data acquired in PC for 15 minutes, for each phase step, at a
sampling rate of 50Hz, using software developed in LabVIEW. Also, in this
receiver, as the phase comparator is working at a frequency of 1.3375 MHz, which
is 1/3 of 4.0125 MHz, ie, down-converted 150 MHz, the phase variation got for the
receiver output is multiplied by 3 to get the actual change in phase. The results are
tabulated and a sample test plot so generated is shown is figure 4.8. The linearity of
the plot indicates that the output phase change measured can be properly correlated
to the input phase change of the incoming signals.
Figure 4.8 Sample test plot of phase test
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78
4.9.2 Radiation mode of testing receiver system
This test is conducted to check the response of the entire ground system consisting
of antennas, cables, outdoor unit, indoor unit and PC based data acquisition system.
The test set up is shown in figure 4.9.
Figure 4.9 Test setup for radiation mode of testing of receiver
In this setup, the actual signal strength reaching the receiver is understood by the
RSSI readings in the front panel of the receiver. As an unmatched single monopole
antenna is used for the transmitter, the signal levels are maintained in the range of -
40 to -50 dBm to make the receiver lock. During the tests, it is found that the
receiver locks onto the signal generator, as soon as RF is switched on. In order to
check if the lock is maintained throughout the sweep, the frequency of 400 MHz is
varied slowly keeping the amplitude constant. It is found that the receiver system
maintains lock throughout the expected range, ensuring that the system is capable of
tracking a LEO satellite.
4.10 Software development
In order to archive and process any data being acquired by the receiver, the output of
the receiver system is connected to PC via a DAQ card. This card requires
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79
acquisition software for acquiring and properly saving the data. The software for
CRABEX LEOS is developed in LabVIEW and has the following major modules.
Time synchronisation
Data Acquisition
Data archival
Automation, with provision for manual mode
Base level signal processing to extract TEC
In order to understand the functions of the above modules, and how to design the
software for automatically acquiring the satellite once it is in view, the operation of
the receiver needs to be understood fully first. This is explained below.
4.10.1 Operation of the CRABEX receiver
The receiver is a standalone system capable of working in Manual and Auto mode.
Manual mode is normally used during testing and characterization of receiver. For
the continuous day to day operation, the receiver works in auto mode. In this mode,
the receiver first works in search mode, wherein it scans continuously through the
initial four channels of each satellite frequency band. This scanning is continuous
until a strong signal is received. A range of voltage has been identified for each
satellite frequency band. This voltage range corresponds to the centre frequency and
satellite Doppler. The receiver starts tracking the satellite at the highest Doppler,
which forms the first channel for that frequency. As the satellite moves, Doppler
gets reduced. This Doppler is effectively filtered out in the tracking loop of PLL by
changing the centre frequency of the filter as the satellite travels. Each of these
frequency bands are denoted by a channel number in the receiver. A filter bandwidth
of ± 500 KHz is chosen for each channel and an overlap of 100 KHz exist between
adjacent channels.
Thus, as soon as a strong enough signal (better than -125 dBm) is received, the
receiver locks onto the signal. The message ‘Lock detected’ appears on the display
window followed immediately by a display of the measured signal strength of the
two frequencies as well as the IF of the 400 MHz viz. 10.7 + 0.0005 MHz. As the
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80
satellite moves, this frequency reduces and as soon as the frequency goes below by
10.6995, the receiver switches over to the next channel, where the IF gets changed
to 10.7005. This is repeated for the entire pass duration. The receiver lock is
indicated by two green LED displays on the front panel. The Doppler shift is
positive when the pass starts as the satellite moves towards the antenna location, and
negative at the end of the pass when the satellite moves away from the antenna
location. The rate of change of Doppler is maximum (of the order of 35 Hz/sec) at
400 MHz, when the satellite is overhead. So the channel number changes faster
during mid pass and slower at both start and end of the pass. During the pass, the
data gets automatically stored onto a file in PC. Once the receiver reaches the last
channel for the satellite being tracked, and the IF comes down to 10.6995 MHz, the
data collection gets terminated.
A sample plot of a portion of the data acquired during a satellite pass of Oscar 32,
elevation 83°, on 04.01.2007 at 12:39:30 is shown in figure 4.10. Here, it can be
seen that the Doppler is high in the beginning, very minimal at the centre and again
increases towards the end.
Figure 4.10 I-Q plots during satellite pass
The date and time of the satellite passes are obtained using a program called
‘TRAKSTAR’. A brief description of the TRAKSTAR program is given below
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81
4.10.2 Tracking the satellites
The Transit satellites are in a circular orbit of 900 km – 1100 km above the Earth.
Two lines of data are injected into the satellites via radio. This data, the Keplerian
two-line elements, predicts the orbit of each satellite. By using the current orbital
elements in the satellite tracking program, the position of the each satellite can be
predicted.
There are several satellite tracking programs available. For this project the
TRAKSTAR program version TrakStar/SGP4 developed by Prof. T. S. Kelso is
used. This program is used for obtaining highly-accurate ephemerides providing
Earth-Centered Inertial (ECI) coordinates, satellite sub-point (latitude, longitude,
and altitude for non-spherical earth), look angles, and right ascension and
declination. It supports determination of visibility conditions for specified
observer(s). It permits the user to calculate any of the following for user-designated
satellites and observing stations.
Earth-Centered Inertial (ECI) Position and Velocity
Latitude, Longitude, and Altitude
Look Angles (Azimuth, Elevation, Range, and Range Rate)
Right Ascension and Declination (Topocentric)
The program begins by allowing the user to select the appropriate option from the
choices displayed as a small window. A window is placed on the screen and the
active option is highlighted. The up and down cursor keys are used to move among
the options; pressing <ENTER> selects the active option. The window remains on
the screen to show which option was selected (only the active window, however, is
highlighted). If Options 3 or 4 are selected, the user is prompted to select whether
only visible passes should be output or all passes. In our case, we have to track all
the satellite passes and so this option is selected. The detail of the first screen is
shown in figure 4.11 below.
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82
Figure 4.11 Details of the first screen of TRAKSTAR
The three types of data files used in this program are *.TLE (two-line element sets)
and *.OB (observer) files and .cfg (configuration) files. For making predictions at
any time it is always best to take the latest TLE. Once updated, the pass predictions
generated with the TLE is valid for 15 days upto one month. The required satellite
TLEs are grouped into a satellite file of filename xxxx.tle, where xxxx denotes a
unique filename chosen by the user. After the input satellite data file is selected,
another window appears which lists the satellites that are grouped in the TLE file. A
sampled TLE file and its details are explained in Chapter 3. The user is then
prompted to select the satellites which are to be tracked ie, the ones to be used for
satellite pass scheduling. This is shown in figure 4.12 below.
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83
Figure 4.12 Program window to tag the satellites to use for pass time calculations
The required satellites are marked in this, and since ‘Look Angles’ option is chosen
in the first menu, a window is also presented to select the observer file (in the same
manner as the satellite data file was selected). The observer file consists of a line for
each observation site with a name (25 characters long), decimal north latitude (in
degrees), decimal east longitude (in degrees), and altitude above mean sea level (in
meters), which corresponds to the ground receiver site. A sample of observer site
file contents is given below.
TVM Trivandrum, IN 8.55 77.0 003
In this, the first three characters of the site name appear in the TRAKSTAR output
and is given to identify the receiver location. Once the input files have been
designated, the start and stop times and the output time interval must be specified.
Once the start time is selected, a window for the stop time appears. After the stop
time is selected, the time interval is presented for input; intervals can start from one
(propagations longer than this will probably not be very accurate). This interval
indicates the time interval needed between two points of satellite path predictions.
While all internal calculations are done using UTC, TRAKSTAR converts input and
output times to local time based on the last two lines of ‘trakstar.cfg’ file. For
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84
example, Indian Standard Time would be represented as: +5.5. Once the data to be
used and the time conditions have been selected, the program begins generating
output.
The screenshot of time selection of TRAKSTAR is shown below as figure 4.13.
Figure 4.13 Screenshot of time selection of TRAKSTAR
The output files generated are text files, with file names having the satellite ID along
with OBS start index which give the look angles for all passes between start time
and stop time. For eg. an output filename ‘OBS19071.txt’ indicates that the data in
this file gives the look angles for satellite ID 19071. A DOS program is made to
convert these “OBS” files to their satellite names like OSCAR32.txt, RADCAL.txt
etc. Each data file consists of date and time followed by azimuth (degrees),
elevation (degrees), range (kilometers), and range rate (km/s) for the entire duration
of prediction, at the time interval specified. Thus if there are five satellites selected
for observation, there would be five text files generated. As these filenames get
rewritten whenever Trakstar is run, these files are copied onto a folder for later use.
These form the input to ASTraS, the Automatic Satellite Tracking Software for the
CRABEX receiver.
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85
4.11 Automation of the receiver operation with ASTraS
For unattended operation of the receiver system for all the required satellite passes,
the data acquisition has been made automatic according to the satellite pass
schedules. A software named ASTraS, (Automatic Satellite Tracking Software) has
been developed in-house with LabVIEW, which automates the data acquisition by
collecting the samples at a prescribed rate, processes the data once the pass is over
and archives both the raw and processed data into unique files. The input for this
software is the output data files of Trakstar as mentioned earlier.
The ASTraS helps to make the system track the LEO beacons automatically. This
software takes care of the serial port data acquisition as well as the analog signal
acquisition. Thus according to the decisions made by a “Passplan” file generated,
the DAQ gets enabled once the satellite is seen over the horizon and tracks it till it
goes out of the horizon. The software also takes care of local data archival once the
pass is over. With a minimum post-processing also incorporated, the front panel
display shows a first-cut TEC plot of the data collected, once a satellite pass is over.
The major features of ASTraS can be summarized as:
Combining different satellite pass files into a single file for one month.
Arranging the above in a chronological order, with relevant details of each
pass like start time, end time, maximum elevation etc given in a single line.
Inclusion of only those passes more than user defined elevation (typ. 30°).
Protection against power failure.
Built-in intelligence to switch to the desired pass in the cases of satellite
pass overlap, with a user defined option.
Automatic data acquisition for required duration with GPS time stamping.
Indication of when the next pass is due.
Online data plots and TEC plot.
Easily upgradable to include other/new similar satellites.
Chapter-4
86
Raw data stored as a text file, with station code, PC date and time as the
filename and the details of the pass, station code and GPS time as the
header.
Processed data with TEC and S4 indices after every pass is also stored
as unique files.
4.11.1 Software design
The ASTraS software is designed using modular programming techniques using
separate, interchangeable components or modules, wherein each module is discrete
scalable, reusable with self-contained functional elements and can easily be changed
to achieve technology transparency. This type of modular design also has the
advantages of cost reduction and flexibility in design, in addition to be easily
possible for augmentation and exclusion. The software has four major modules
which provide the following basic functionalities.
Pass plan generation module: Generates an “all pass” text file giving details
of all the satellites that need to be tracked for the month in chronological
order. In case of a satellite pass overlap, the one with the highest elevation is
chosen, by default.
Data acquisition module: Automatic start and stop of data acquisition of the
serial port for GPS data and DAQ card analog channels for phase data as the
satellite moves across the horizon. Displays online raw data and saves.
Archival module: Archive data with a file name derived from year, month
and date.
Processing module: Post process received data to calculate slant TEC, save
the data and display.
ASTraS is designed to take a text file as input, whose contents describe the location
of the individual satellite pass schedule files generated by Trakstar, as explained in
the last section. By this method, it is always easy to add or delete any new satellite
into the group. In Trakstar the stop time is kept as one month from the start time,
with the time interval for update as one minute. Thus in each of the output files
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In the next step, all the intermediate data files are combined to form an “all pass”
file. This processing can be done for a maximum of any one month at a time so that
there can be upto thirty one intermediate files, which would all be combined
automatically into a single file. A blank line is inserted after the total passes for each
day to identify them easily. Thus the entire “master pass” file is scanned and re-
arranged day by day to generate an “all pass” file, which effectively describes the
pass schedule for the entire month. The sample “all pass” file corresponding to the
“master pass” file mentioned in figure 4.15 is shown below in figure 4.18. Thus the
first module ends with the generation of “all pass” file.
Figure 4.18 Sample “all pass” file contents
The design challenge of two satellite passes occurring simultaneously is managed
with a user defined handle. In the program front panel itself, the user can choose
between maximum elevation or higher power. If maximum elevation is chosen,
which is the default mode, ASTraS compares the elevation column of these passes,
selects the satellite with the maximum elevation and discards the others. The second
choice of higher power is related to the onboard transmitted power of the satellite
and the one with higher power is chosen. As COSMOS series satellites offer higher
power than OSCAR series, this calls for selection of COSMOS satellites, if there is
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an overlap between OSCAR and COSMOS though OSCAR may have a higher
elevation for the observation site than COSMOS. Pass overlapping also addresses
the occurrence of the start time of a pass inside the pass duration of yet another
satellite.
In the second module, the system goes to the data acquisition mode, by reading the
first line of the “all pass” file and comparing the date-time with the PC date-time. In
the case when pass time is behind PC time, the corresponding line is deleted from
the “all pass” file and the next line is read. This is repeated till pass time is ahead of
PC time. This strategy thus takes care of any inadvertent power failure. Once the
start time of the next pass is identified, the system enters into a wait loop, for
duration of the difference of pass start time and PC time. Once these two become the
same, the DAQ subroutine is initiated.
The DAQ subroutine is initialised with the RS232 serial port acquisition. The
program reads the GPRMC command through COM1, extracts date, time, lat, long
and altitude of the location from this and saves this as a header line in the raw data
file. A single program takes care of both serial port as well as parallel port
acquisition. The major highlight of the software is that the use of GPS is transparent
to the user. The user does not have to do any other operation to initialize the GPS in
order to get accurate time. In order to ensure time synchronization, the GPS data is
logged onto the data file, both at the start and stop of acquisition. Knowing the scan
rate of the acquired data, the number of samples that can be collected during any
satellite pass can be found out, and this gives a very accurate time indication. In case
GPS data is not received properly, the software waits for a minute, and then comes
out of the loop, logs in PC date and time and proceeds with analog channel
acquisition. This helps to ensure there is no data loss.
The data acquisition from the four analog channels (ie, I, Q, Amplitude 150,
Amplitude 400) is enabled by initializing the analog inputs and reading it
continuously at a pre-fixed scan rate of 100 Hz. The raw data of all the four channels
get saved onto the raw data file. An online data plot is also displayed with the raw
data on the front panel.
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The file name for this raw data file is extracted from PC date and time, which
ensures each file name is unique. Once data acquisition starts, the stop time of the
pass is compared with the PC time. When the stop time matches with PC time, the
data acquisition stops, saves raw data, and deletes the details of the finished pass
from “all pass” file and comes out of this subroutine.
The archive module then archives the data onto a day folder, which is either created
new, or exists already. The archival format is year, month, and date folders. In the
Processing module, TEC is calculated from the raw data obtained above. Once the I
and Q channel outputs are acquired in the output file, off-line processing is done to
compute the phase data. The relative phase is given by
tan (4.13)
This relative phase obtained between I and Q channels is proportional to the relative
slant TEC (STEC) along the propagation path of the signal as
(4.14)
where φ is measured in radians, STEC is in m-2 and CD = 1.6132x 10-15 for NNSS
satellites as given by Leitinger[1994]. The values of the phase thus obtained lies
between 0 and 2π and when plotted against time, they have the appearance of ramps,
as shown in figure below.
Figure 4.19 Phase plot of raw data from receiver for a typical satellite pass
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By cumulating these phase values, it is possible to obtain continuous phase records
in complete cycles with the minimum value of the curve arbitrarily set to zero. This
method is adopted in the processing module of ASTraS. Here initial data quality
checks are done first, which deals with the initial noise removal also. Since the DAQ
starts from very low elevation itself, there can be noisy data at the start when the
receiver system has not fully locked to the satellite. This can be identified by
calculating the phase difference between the I and Q channels and finding very
sharp transitions between adjacent samples. These values are then removed and only
valid data is saved onto a temporary file. In the new file, a 10 point mean is taken
before finding the cumulative differential phase. This is then multiplied with a
constant to give the values in TEC units where in one TEC corresponds to ~912o
phase difference. This TEC file is saved onto a “TEC” folder with file name as
“original name_TEC.txt”. A data plot of the same is also provided in the front panel.
In order to reduce the initial noisy data values, a control called ‘pass time’
calibration is provided in the program front panel. This represents a delay in seconds
which can be applied at the start of the DAQ program, so that the ‘start time’
mentioned in the “all pass” file is delayed by this value. This helps in the post
processing by reducing the initial noisy data values. But it has the disadvantage that
there could be some initial loss of good data as the satellite rise time varies with
respect to its elevation and azimuth at the observing site and can be different for
different orbits.
The modular level architecture of ASTraS is depicted in the flowchart in figure
4.20.
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Figure 4.20 Software architecture of ASTraS
4.11.2 Software implementation
The front panel of the ASTraS is shown in figure 4.21. The software is developed
and tested separately for each module, before integrating it with the receiver system.
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Figure 4.21 ASTraS front panel before acquisition
The input to the software is given on the top-most left corner, which is the location
of the files. This can be manually entered or browsed using a drop-down menu. The
‘Pass time calibration’ and the ‘elevation/satellite’ power switch are also selected as
per the requirement. The output from the first module forms an indicator ‘Next Pass
date and time’, which is the first line from the “all pass” file. The PC time is
continuously updated, as part of the second module, and this also appears as
‘Current System Date and Time’. The red indicator ‘Waiting for pass’ gets changed
to ‘DAQ in progress’ during satellite tracking. The top display is an output of the
second module, which plots the raw data as it is being acquired. Once the data is
acquired, the raw data is archived and the fourth module of TEC post-processing
takes place, whose plot gets displayed in the second waveform chart. ‘Offset of good
plot’ gives the initial noise data values. Another front panel control called ‘TEC
duration’ helps to remove the noise at the end of data file, since it is difficult to find
out sharp transients at the end unlike in the start position. The TEC processing will
be done for a maximum of this duration. This TEC plot is thus an indication of the
system performance. Since raw data is also available, the TEC values can be
recalculated during later off line processing and scientific evaluation as per
requirements.
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The software has been integrated with the coherent beacon receiver system. The
TLE files for the satellites that need to be tracked are downloaded from NORAD site
every month. Trakstar program is then run and the output generated, ASTraS is then
run with location of this folder as its input.
The inclusion/ exclusion of any new/inactive satellite is also easy as it involves
adding/deleting the particular satellite TLE to the Trakstar program and
including/removing the corresponding file name in the text folder. This mode has
also been tried out with the RaBIT satellite included in the existing list and the
currently non-operational Oscar series removed. This software has been successfully
used for CRABEX data collection and archival since 2007 at the receiver station at
Trivandrum. The output screen during a satellite pass is shown in figure 4.22 with
the TEC plot of the previous pass. The top plot in the grey coloured background
shows the I and Q channels (white and red curves) of the phase data, which is used
for TEC calculation. The sine wave denotes the instantaneous Doppler frequency of
the satellite. The other two curves (green and blue) in this denote the amplitude
channels of the raw data. This is used to study ionospheric scintillation. The bottom
plate shows the vertical TEC calculated for the pass that has happened earlier, which
has 282 initial noise values and TEC is calculated for duration of 10 minutes after
noise removal.
Figure 4.22 Front panel during a satellite pass
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As the receiver gives amplitude channels also which can be used for scintillation
measurement at the location, a separate module is developed offline to give the
scintillation indices for the entire pass duration. It is understood that a study of
scintillation indices gives a direct indication of the changes in the ionosphere and
hence this data can also be used for scientific analysis.
Similar receivers located at different stations along the satellite orbital plane also
record TEC simultaneously for a satellite pass. For this project, the stations are
located currently at Space Science Office, ISRO HQ, Bangalore; National Balloon
Facility, TIFR, Hyderabad; Master Control Facility, Bhopal and NPL, New Delhi,
apart from the focal centre at SPL, Trivandrum, as depicted by yellow circles in the
map shown in figure 4.23.
Figure 4.23 Receivers located at different stations in CRABEX project
A typical STEC plot obtained for simultaneous reception and recorded by all these
stations for a satellite pass starting at 11:10 IST (first lock time at Bangalore) is
shown in figure 4.24. It is to be noted that for tomographic studies and further
scientific analysis, this STEC would be first converted to VTEC as explained earlier.
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Figure 4.24 Simultaneous TEC plots from 5 stations
4.12 Overview of results of present system and use of spread
spectrum modulation
Once the electron density NT at a particular point during the satellite pass is
established, the variation of NT for all ray paths during the period for which data was
acquired can be deduced as shown by Ramarao [2004]. Thus with the data received
from a single station, the variation of the TEC over a particular location can be
computed and examined, with a 2nπ ambiguity on the initial phase. Now, if there
exists more than one receiving station located along the satellite path, all these
would be able to provide such TEC variations for the same satellite pass, as
described earlier. It can be seen from figure 4.24 that the initial phase in even any
two of the cases is not exactly matching. Thus in the technique of radio tomography
developed with this simultaneous TEC data from the chain of ground receiving
stations, proper care is taken to overcome this initial phase ambiguity by various
methods , one of which is the popular 2-station method of Leitinger [1975].
In this work, a new method of addressing this initial phase ambiguity is studied. This
explores the use of a spread spectrum modulated signal to be transmitted from the
satellite using either binary amplitude or phase modulation techniques, so that for a
single receiver location the group delay at these frequencies is measurable, which
varies approximately as the inverse square of the carrier frequency. The received
-10 0 10 20 30 40 500
5
10
15
20
25
30
35
40
TEC
(TEC
U)
Geographic Latitude(Deg)
Trivandrum Bangalore HyderabadBhopal Delhi
27/03/2008 11:10 IST
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signal may then be decoded by correlating it in known fashion, with a selected
portion of the transmitted pseudo-random noise coded signal as explained by David
Farmer et al [1990].
One of the simplest methods of modulating the signal in order to measure phase
difference at the receiver is to have a single carrier signal, having an in-phase and
quadrature component, modulated by the code so that the final signal is of the form
. . (4.15)
where the amplitude of signal is represented by the constants Ai. Assuming the
clocks for the code and the carrier frequencies are coherently derived from the same
frequency source at the transmitter and receiver, it is possible to derive the phase
change undergone by the carrier because of the transmission medium.
For the simulation study, Maximal Length Sequences or m-sequences having good
auto-correlation and balance property with minimum autocorrelation side lobe peak
to minimize the false-lock probability during code acquisition is used. In order to
assess the feasibility of the use of spread spectrum modulation for derivation of
phase between two coherent signals, various levels of mathematical simulations are
done. In the LabVIEW environment, the basic building blocks of signal and noise
generators are available and for PN code generation and modulation-demodulation
techniques, modules available in the modulation toolkit have been used.
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4.13 Design methodology of Orthogonal Carrier Spread Spectrum
system
The design involves generation of a coherent I-Q signal modulated separately by the
same PN code. The design flow block diagram is represented in figure 4.25.
Figure 4.25 (a) Design flow block diagram: Transmitter and channel
Figure 4.25 (b) Design flow block diagram: Receiver
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Here, the data signal is generated with the help of a ‘Sine Wave Generator (SWG)’
module, which has frequency, amplitude, initial offset, sampling rate and number of
samples, as its control inputs. In order to maintain a coherent mode of signal
generation, all these parameters are derived from the same set of ‘controls’ and
given to both the SWGs, with an added offset of 900 for the latter one i.e., the signals
generated are A sinωt & A sin(ωt+ 900 ). The PRN code is generated with a ‘MLS
Signal Generator (MSG)’ module. In order to maintain uniformity in the simulation,
the sampling rate and number of samples for SWG and MSG are kept same. It is
possible to change the properties of the MSG by varying its polynomial order. The
output of the two SWGs are digitized and multiplied with MSG separately. This
forms the transmitter.
The communication channel distorts the signal passing through it by changing its
amplitude and phase (attenuation and polarization changes). These effects are
simulated by using a ‘Noise Generator ‘NG’ VI, which adds noise to the signal and a
Delay Block ‘DB’ VI, which provides a time shift in the signal, which gets
translated effectively to a frequency or phase shift. As explained in Chapter 2, the
ionospheric model for simulation can be approximated to a parabolic phase
functional filter, which in digital domain can be considered as equivalent to a linear
time-delay filter, as described by Michel C Jeruchim et al [2000]. The maximum
time delay that occurs is called the delay spread of the signal in that environment.
This delay spread can be short so that it is less than symbol time or larger. Both
cases cause different types of degradation to the signal. In general when the delay
spread is less than one symbol, we get what is called flat fading. When delay spread
is much larger than one symbol that is called frequency- selective fading. Rayleigh
fading is a term used when there is no direct component and all signals reaching the
receiver are reflected.
At the receiver, the two signals are received separately and a locally generated PN
code is made to match with the incoming signals separately. In order to retrieve the
data, this PN code should be in synchronization with the transmitted code. The
synchronization problem can be split into three major parts: code-acquisition, code-
tracking and carrier-tracking as detailed by Glas and Skolnik [1994]. Acquisition is
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the process in which the local PN code is shifted along the received signal to find
synchronization.
A novel method of code tracking and acquisition is attempted for the simulation, by
utilizing the cross correlation property of the PN code. It is already shown that the
selected PN code has a very good auto-correlation, implying that when the incoming
signal and locally generated signals are in phase, they will have maximum
correlation. In order to get this synchronization, the cross-correlation of the locally
generated signal and the incoming signal are calculated, and it is found that when
both are in phase, they produce a maximum at the centre, which happens to be equal
to the number of samples. This avoids all instances where in the maximum value of
the cross correlation is slightly more than the ideal or predicted value. With the
matching achieved, the baseband signal is retrieved by multiplying the received
signals separately with the shifted PN code, followed by a narrow band Butterworth
BPF of fifth order or more in each of the receiver channels.
The design of the filter plays a major role in final detection of the signal as this
provides a handle on the received signal properties. The primary applications of
filtering in communication systems, as defined by Michel C Jeruchim et al [2000],
is to select desired signals, minimize the effects of noise and interference, modify
the spectra of signals and shape the time domain properties of digital waveforms. All
digital communication systems include a filter in the receiver which performs the
task of matched filtering, with the input matched to the received pulse. Here, the
bandwidth of the filter is chosen to be < 10% of the centre frequency, so as to
eliminate all higher order frequencies. The two filtered signals are then fed to a
phase detector which gives the instantaneous phase difference between the two
received signals. As is possible with simulations, this measured value is then
compared with the instantaneous phase difference from the input end.
The entire program is implemented as a three-frame block diagram for sequential
execution of the various frames, since LabVIEW is based on dataflow programming
by default. The first frame is the transmitter, the second denotes channel and third is
the receiver. The program is executed frame by frame only. This helps in
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eliminating any difference in path changes occurring due to the program processing
time. The phase difference between the baseband inputs are measured in the first
frame and passed on directly to the third frame for final comparison with the
received phase difference.
4.14 Results of simulation studies
During the simulation, all the variable inputs like frequency, amplitude, sampling
rate, number of samples and polynomial order were varied one by one at the
transmitter end. Variation of the frequency and number of samples called for tuning
of the receiver by varying the filter parameters. Keeping the sampling rate constant
at 1 KHz and number of samples at 1000, the baseband signal was varied from 1 to
100 Hz in steps and the polynomial order varied from 1 to 64. With the number of
samples as constant, the auto correlation property of various MLS sequences are
studied to finalize the order suitable for this simulation. A 3-D plot of the same with
the change in the peak value is shown in figure 4.26. A set of eight MLS sequences
were identified based on the peak value of their autocorrelation. As explained above,
it is also noted that the autocorrelation curves has a total of 2 x number of samples
data points, with the maximum always appearing at the centre.
Figure 4.26 Autocorrelation of various MLS sequences
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In the communication channel, the DB VI provides varying levels of time shift to
control the phase delay of the transmitted signals. The NG VI introduces random
white noise depicting the channel noise. These are added to the transmitter signals.
At the receiver end the locally generated PN Code is multiplied with the phase
shifted incoming signal and the cross-correlation of the product is examined.
Through several levels of iteration, it has been found that the cross-correlation is
optimum when the peak of the product coincides with the centre of the correlated
values, which happens when the peak is equal to the number of the product samples.
It is also found that at this maximization at the centre point, the tracking delay is the
same as the delay introduced in the communication channel. Thus this technique
effectively removes the uncertainty introduced and helps in effective demodulation
by tracking the code properly.
In the final filtering process it is found that a Butterworth band pass filter of order 5-
7 with fH = f + 0.5 and fL = f - 0.5 works best in the lower range of ‘f’. As the signals
generated are quadrature in nature, the same procedure is followed in the second
channel also. The filtered signals from both the channels are passed through a phase
detector to find the output phase variation. As it is possible to have a comparison of
the initial phase of the signals at the transmitter and the final phase derived at the
receiver, this is also incorporated in the simulation program. The results show that
since we have a coherent quadrature type signals transmitted from the input, a
similar type of recovery at the receiver would help in finding out the phase
difference between the received signals. A comparison of the received phase
difference with the ideal one would eventually lead to the actual phase change that
has occurred due to the communication channel. The simulation screenshots are
shown here for a very low frequency. The front panel screenshots with the
transmitter, channel and receiver portion of a typical case of f = 2 Hz, number of
samples = 1000, sampling rate = 1000, Polynomial Order of MLS = 25 and delay =
100, is shown in figure 4.27. In order to optimize the received signals, the receiver
BPF was designed for order 6, with cut-off at 1.97 Hz and 2.06 Hz. It is found that
the receiver attains phase lock with the incoming signal at delay + 1 point always.
This is again verified with the incoming signal from the other data channel also.
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Figure 4.27 Detailed front panel of the simulation software
Varying delays were given for DB and these delays were automatically detected in
the tracking loop and the signals were detected without phase deviation. This
remains true until up to 40% of the sampling frequency, after which retuning of the
receiver filter is needed to get the exact signal back. Varying the polynomial order
of MLS also introduces changes in the entire system. The phase differences are also
found in all cases and are found to be within 1% of expected phase.
The tests are done by adding noise signals also at the channel level. It has been
found that this does not vary the property of the received signal considerably as SS
signal itself is like a noisy signal to a casual receiver. The phase shift between input
signals was also varied in steps from 0 to 3600 and it has been found that the
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reception is optimized for 900 phase difference between the signals i.e., quadrature
signals. Thus a pair of quadrature signals, modulated by a suitable MLS code can be
used to exactly derive the phase
4.15 Summary
The chapter explains in detail the design and development of a coherent beacon
receiver system for tracking LEO beacon satellites for ionospheric studies. A
receiver chain is established across the Indian continent with similar receivers for
ionospheric tomography studies. The problem of 2nπ ambiguity with the initial
phase measurement for TEC calculation is brought out. To address this, a spread
spectrum beacon system is proposed using simulation studies.
A preliminary result of simulation of coherent spread spectrum signals with
LabVIEW software has been detailed here. A new method of code tracking is
attempted by making use of the location of the peak of the cross-correlated data. For
a pair of coherent quadrature signals transmitted, the receiver is able to accurately
derive the phase difference between the incoming signals. This technique can thus
be proposed for measurement of phase for ionospheric studies in future beacon
systems, with appropriate transmitting frequencies and PN codes.