d4.4 - activity report on cnt and graphene based … · canseliet 1 ex national research and...

CARBON BASED SMART SYSTEM FOR WIRELESS APPLICATION

Start Date : 01/09/12 Project n° 318352 Duration : 48 months Topic addressed : Very advanced nanoelectronic comp onents: design, engineering, technology and manufacturability

WORK PACKAGE 4

DELIVERABLE D4.4

Activity report on CNT and graphene based antenna t ests

Due date : T0+40 Submission date : T0+48

Lead contractor for this deliverable : UNIVPM

Dissemination level : PP – Restricted to other pro gramme participants

D4.4 : Activity report on CNT and graphene based antenna test

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WORK PACKAGE 4 : Test activities

PARTNERS ORGANISATION APPROVAL

Name Function Date Signature

Prepared by: M.Dragoman R&D Engineer 04/10/2016

Prepared by: D.Mencarelli R&D Engineer 04/10/2016

Prepared by: S.Xavier R&D Engineer 04/10/2016

Approved by: Afshin Ziaei Research Program Manager 05/11/2016

DISTRIBUTION LIST

QUANTITY ORGANIZATION NAMES

1 ex Thales Research and Technology TRT Afshin ZIAEI

1 ex Chalmers University of Technology CHALMERS Johan LIU

1 ex Foundation for Research & Technology - Hellas FORTH George KONSTANDINIS

1 ex Laboratoire d’Architecture et d’Analyse des Systèmes

CNRS-LAAS George DELIGEORGIS

1 ex Université Pierre et Marie Curie UPMC Charlotte TRIPON-CANSELIET

1 ex National Research and Development Institute for Microtechnologies

IMT Mircea DRAGOMAN

1 ex Graphene Industries GI Peter BLAKE

1 ex Thales Systèmes Aéroportés TSA Yves MANCUSO

1 ex SHT Smart High-Tech AB SHT Yifeng FU

1 ex Universita politecnica delle Marche UNIVPM Luca PIERANTONI

1 ex Linköping University LiU Rositsa YAKIMOVA

1 ex Fundacio Privada Institute Catala de Nanotecnologia

ICN Clivia SOTOMAYOR

1 ex Tyndall-UCC Tyndall Mircea MODREANU


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CHANGE RECORD SHEET

REVISION LETTER DATE PAGE NUMBER DESCRIPTION

Template

V1 15/09/2016 23 IMT & UNIVPM and UPMC Contribution

V2 04/10/2016 26 Final Version


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CONTENTS 1 EXECUTIVE SUMMARY ............................................................................................................... 5

1.1 GRAPHENE ANTENNA (G-ANTENNA) ........................................................................................... 5 1.2 CNT ANTENNA .......................................................................................................................... 6

2 EXPERIMENTAL CHARACTERIZATION OF THE G-ANTENNA .... ............................................. 7

2.1 RETURN LOSS CHARACTERIZATION (LAAS) ................................................................................ 7

2.2 RETURN LOSS CHARACTERIZATION (IMT) ................................................................................. 11

2.3 RADIATION PATTERN CHARACTERIZATION ................................................................................. 13

2.4 GRAPHENE PARAMETERS EXTRACTION AND EQUIVALENT CIRCUIT .............................................. 15

2.5 X BAND SLOT ANTENNA LOADED WITH A GRAPHENE PATCH: E.M. MODEL AND SIMULATION RESULTS .......................................................................................................................................... 16

3 EXPERIMENTAL CHARACTERIZATION OF THE CNT ANTENNA .. ......................................... 20

3.1 CNT BUNDLE: PARAMETER EXTRACTION ................................................................................... 20 3.2 INSERTION LOSS CHARACTERIZATION ....................................................................................... 21

3.3 MODELLING OF A 10 GHZ CNT ANTENNA WITH INDUCTIVE COUPLING ......................................... 24

4 CONCLUSION ............................................................................................................................ 25

4.1 GRAPHENE ANTENNA .............................................................................................................. 25 4.2 CNT ANTENNA ............................................................................................................................ 26


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1 EXECUTIVE SUMMARY This deliverable covers WP4 testing activities on CNT and Graphene antennas fabricated in WP3 (IMT, TRT, UNIVPM, UPMC). Graphene and CNT antennas have been tested by on probe measurements in order to provide characterization of their e.m. parameters, namely input matching and radiation pattern. The antenna parameters (radiation efficiency, gain, and cross-polarization ratio) can be derived from the above measurements. The available equipment includes i) Agilent PNA-X: 10 MHz - 20 GHz, 801 points, IF BWD 10kHz, 1-port SOL calibration, ii) Probe station Suss PA200, iii) GSG Probe 150µm pitch (Z-Probe cascade).

1.1 GRAPHENE ANTENNA (G-ANTENNA)

Fig. 1: (a) CST MWS design (with main dimensions) of the graphene CPA; (b) detail of the feeding point; (c) optical image of the fabricated graphene CPA; (d) different experimental configurations, as indicated in Table I.

(a) (b)

(c)

(d)


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The graphene coplanar patch antenna (CPA) is a new type of antenna based on a graphene monolayer. Its design was presented in the WP2. The antennas are fabricated on a 4 in. high-resistivity Si wafer, with a ~300 nm SiO2 layer grown through thermal oxidation. A CVD grown graphene layer is transferred on the SiO2, while the CPW structure is made of 400-nm thick gold. The graphene antenna was fabricated on the CVD-grown graphene wafer having a Si/SiO2 substrate with a thickness of 500 µm/300 nm, respectively. Figs. 1a-1b show the CST MWS design of the graphene CPA with the main dimensions, whereas Fig. 1c displays the optical image of one of the fabricated prototypes. The antenna was also tested with a metal patch placed on top of the die in order to check effect on radiation: two positions are assumed, either very close to the probes (setup 4) or only on the antenna patch (setup 3). The situation is depicted in Fig. 1d. Experimental test was made by considering its return loss and the radiation diagrams at different frequencies. Finally, an equivalent circuit model by means of AWR will be proposed to further characterize and validate the fabricated device.

Table I

The DUT is tested under different conditions, both in terms of bias (between Signal-lateral GND and Signal-Back GND) and boundary conditions (Back GND and absorbing/insulating foam): Table I summarizes the different setups.

1.2 CNT ANTENNA The CNT antenna under test is shown in Fig. 2 : basically, it is given by a coplanar waveguide terminated with a vertically grown CNT bundle.

Aspect ratio: >7


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Fig. 2. CNT-based antenna and experimental setups.

In Fig. 2 above, we also report the experimental setup used to measure the electromagnetic response of the CNT antenna, both for ESPCI/UPMS and LAAS laboratories. The main features of the experimental approach of ESPCI/UPMC are given by i) coplanar probes RF excitation, ii) SOLT (TUOSM) calibration with Al2O3 calibration kit, iii) Darkness environment (to avoid substrate sensitivity to light, iv) performances repetability on several identical samples. Similarly, for LAAS measurements, we have : i) automatic platform, ii) coplanar probes RF excitation, iii) Far-Field calibration, iv) performances repetability on identical samples.

2 EXPERIMENTAL CHARACTERIZATION OF THE G-ANTENNA

2.1 RETURN LOSS CHARACTERIZATION (LAAS) The following Fig. 3 shows amplitude and phase of the reflection coefficient of the graphene antenna after calibration by standard de-embedding procedure. The effect of CPW signal-ground biasing (in case of antenna suspended on GND chuck) are highlighted above. In particular, the reported curves show, as an important result, that the reflection parameter of the antenna can be tuned by a DC voltage. Electromagnetic resonances in the in the X band (8–12 GHz), are clearly observable in Fig. 3: two main resonances occur @ 9.95 and 10.75GHz (among many other less pronounced). Bias does have an effect but only minor: ~50MHz reduction for 0-90V on both resonances. From the scattering parameters, a clear loss behaviour is observed:

SETUP ESPCI/UPMC

SETUP LAAS


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abs(S11)< -10dB from 4GHz up] Since no losses (<1dB/cm at most ) are expected from HRS substrate they are likely due to graphene (resistive load).

Fig. 3. Reflection coefficient (amplitude and phase) of the graphene antenna for different voltage bias.

2 4 6 8 10 12 14 16 180 20

-30

-20

-10

-40

0

dB(S

(1,1

))dB

(S(2

,2))

dB(S

(3,3

))

9.5 10.0 10.5 11.09.0 11.5

-30

-20

-10

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0

freq, GHz

dB(S

(1,1

))dB

(S(2

,2))

dB(S

(3,3

))

2 4 6 8 10 12 14 16 180 20

-100

0

100

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200

phas

e(S

(1,1

))ph

ase(

S(2

,2))

phas

e(S

(3,3

))

9.5 10.0 10.5 11.09.0 11.5

-100

0

100

-200

200

freq, GHz

phas

e(S

(1,1

))ph

ase(

S(2

,2))

phas

e(S

(3,3

))

Bias (Signal -Lateral GND):

0 V

50 V

90 V


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2 4 6 8 10 12 14 16 180 20

-30

-20

-10

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0

dB

(S(4

,4))

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freq, GHz

dB(S

(4,4

))dB

(S(5

,5))

2 4 6 8 10 12 14 16 180 20

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0

100

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200

phas

e(S

(4,4

))ph

ase(

S(5

,5))

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0

100

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200

freq, GHz

phas

e(S

(4,4

))ph

ase

(S(5

,5))

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-10

0

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freq, GHz

imag

(Z_4

)im

ag(Z

_5)

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60

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90

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freq, GHz

real

(Z_4

)re

al(Z

_5)

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10

20

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50

60

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90

0

100

real

(Z_4

)re

al(Z

_5)

9.5 10.0 10.5 11.09.0 11.5

-40

-30

-20

-10

0

10

20

30

40

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50

freq, GHz

imag

(Z_4

)

0 V 50 V 90 V

(no-bias)

(no-bias)

Setup 2 (Bias Signal – Chuck):

Setup 3 (met. patch cover. CPW-TL+G anten):

Setup 4 (met. patch cover. ONLY G anten):


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Fig. 4. Amplitude and phase of the reflection coefficient of the graphene antenna, and corresponding input impedance, for different voltage bias, without metal cover, and with metal cover, over graphene only, and over both graphene and feeding line.

With reference to setups 2,3, and 4 of Table I, we investigate, in Fig. 4, the effect of back GND bias and top-metal cover (antenna suspended on GND chuck). In Fig. 5, we consider the effect of removing back GND plane, i.e. antenna suspended on absorbing/insulating foam. In this case, the resonances disappear, which means that the slot radiation is affected by the contribution of patch

radiation modes.

Fig. 5. S11 (amplitude and phase) and Z11 (real and imaginary parts) of the graphene antenna for different voltage bias, with insulating and absorbent foam.

2 4 6 8 10 12 14 16 180 20

-30

-20

-10

-40

0

dB(S

(9,9

))dB

(S(1

0,10

))

2 4 6 8 10 12 14 16 180 20

-100

0

100

-200

200

freq, GHz

phas

e(S

(9,9

))ph

ase(

S(1

0,10

))

2 4 6 8 10 12 14 16 180 20

10

20

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50

60

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100

real

(Z_9

)re

al(Z

_10)

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-30

-20

-10

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10

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50

freq, GHz

imag

(Z_9

)im

ag(Z

_10)

0 V 50 V (Bias G-S)

0 V

Setup 5 (Absorbent foam) :

Setup 6 (Insulating foam): No Bias


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Fig. 6. S11 (amplitude and phase) of the graphene antenna for different voltage bias, for the setups of Table I. For clearness and readability, in the Fig. 6 above, we report a summary of all the above results concernig the input reflection. In general, from figures 1 to 5, we can observe that the voltage bias does have an effect, even if not so large: ~50MHz reduction for 0-90V on both resonances.

2.2 RETURN LOSS CHARACTERIZATION (IMT)

The experimental setup in IMT laboratories, for testing return loss and radiation pattern, is shown in Fig. 7.

2 4 6 8 10 12 14 16 180 20

-40

-30

-20

-10

-50

0

dB

(S(1

,1))

dB

(S(4

,4))

9.5 10.0 10.5 11.09.0 11.5

-40

-30

-20

-10

-50

0

freq, GHzdB

(S(1

,1))

dB(S

(4,4

))


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To detector

Generator Agilent E8257D

graphene antenna

15 cm

Fig. 7. Graphene antenna measurement in IMT labs. We measured the reflection parameter |S11| as a function of frequency (f) using an on-wafer probe station linked to an Anritsu-37397D Vector Network Analyzer (VNA). The SOLT (Short, Open, Load and Thru) calibration was performed to calibrate the VNA before measurements. A CS-5 standard CPW calibration kit was used. The graphene-based antenna is DC biased using the probe station ground-signal-ground (G-S-G) probe tips. So, the DC voltage is applied between the central conductor and the grounds of the CPW using the internal bias tee of the VNA coupled to a Keithley 4200-SCS (Semiconductor Characterization System) able to provide controllable voltage/current sources. The antenna is fixed on a metallic plate playing the role of a backside metallization. The results are displayed in Fig. 8 for the X band (8-12 GHz). We see an overall reflection loss of about -9 dB and two resonances located at 8.8 GHz and 11.4 GHz, corresponding to |S11| minimum values of -12.2 dB and -13.4 dB, respectively, at 0 V applied DC bias. We see that the entire S11 dependence on frequency is shifted up and down depending on the applied voltages, while the resonances are shifted left or right with about 24 MHz. This is due to the fact that the surface resistance of the antenna is decreasing at 50 V and, respectively, increasing at -200 V. We see that, at 11.4 GHz, when the surface resistance is decreasing, the matching of antenna is improving with 1.5 dB while when the surface resistance is increasing we lose about 0.4 dB. The large values of the DC applied voltage are due to the fact that the slot widths are rather large: 350 µm on x-axis and 600 µm on y-axis (see Fig. 1a), and so the field applied between the graphene patch and the coplanar ground is rather small.


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Fig. 8. Input matching of the antenna, for different values of the voltage Bias. The antenna is measured in two operative conditions: on ground plane (PEC), and on adsorbent material (resistive sheet). As mentioned, the above measurements confirm tunability of the graphene antenna, which is an achievement well beyond state-of-the-art. The input matching of the antenna is changed with an applied DC field without using ferromagnetic or ferroelectric materials. As a future development of NANORF, the DC voltage will be reduced by optimization of the biasing configuration of the antenna.

2.3 RADIATION PATTERN CHARACTERIZATION The radiation pattern of the graphene antenna was measured using the setup from Fig. 7. The antenna was contacted on wafer with a G-S-G probe and used as emitter. It was fed by a PSG Analog Signal Generator (Agilent E8257C) with a 6-15 GHz microwave signal, modulated in amplitude (10 kHz square AM). A X-band waveguide flange was placed at a distance of 150 mm (satisfying far-field conditions in the X-band) and connected to a 10 MHz-40 GHz detector (Anritsu). The detected signal is amplified by a SR560 LNA and plotted on an oscilloscope (Tektronix DPO2024). Since the antenna has a thickness of 500 µm, much smaller than the microwave radiation wavelengths, which are of the order of few cm at least, the antenna will not only radiate above the radiation plane represented by slots but also below, thus reducing the radiation efficiency. So, we have measured the detected signal at the position (0,0) – perpendicular to the metallization – in two conditions: (i) placing a microwave absorbent on the backside of the antenna and (ii) placing a metallic surface on the backside of the antenna. The 2D radiation patterns in the H-plane (orthogonal to the feed line of the antenna) were recorded at 7 GHz and 12 GHz in the case of the graphene-based antenna with a backside metallized surface (see Figs. 9a and 9b), and at 8 GHz, 10 GHz, and 12 GHz in the case of the graphene-based antenna with a backside absorbent (see Fig. 9c for the X-band central frequency, i.e. 10 GHz). The 3 dB beamwidth (when the received power is reduced by half compared to the maximum) is between 40° and 80 ° for the radiation patterns presented in Figs. 9a-c. In the case of the graphene-based antenna having a backside metallization, there are no side lobes at 7 GHz with one side lobe appearing at 12 GHz, meaning that part of the radiated microwave


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signal travels through the substrate towards the backside metallization and is then reflected back, being added with the frontside radiated signal. For the backside absorbent case, although the effects of the backside radiation are strongly diminished, multiple side lobes appear at 8 GHz and 12 GHz. The simulated gain of the antenna is -8 dB without backside metallization and -6 dB with backside metallization at 10 GHz. No calibrated gain measurements could be performed with the available equipment. The radiation efficiency and gain are weak but can be improved by reducing the sheet resistance of graphene. This can be achieved by optimizing the technological process and the antenna topology, including the graphene DC bias configuration. In our case, we have biased graphene with a DC electric field via coplanar configuration so the electric field is tangential to the graphene surface.

(a)

(b)


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measures FEM solver,

decreasing

surface

impedance:

measure 650, 600, 550

Fig. 9: a) radiation pattern of the graphene-based antenna with a backside metallization normalized at the maximum value at 7 GHz. The simulated 3 dB beamwidth is 69.4°, b) radiation pattern of the graphene-ba sed antenna with a backside metallization normalized at the maximum value at 12 GHz. The simulated 3 dB beamwidth is 76.5°, c) radiation pattern of the gra phene-based antenna with a backside microwave absorbent normalized at the maximum value at 10 GHz. The simulated 3 dB beamwidth is 77.2°. A further improvement of the proposed graphene antenna is currently on going and is based upon a thinner HR Si layer (down to 100 µm) under the graphene resonator. This way, it is possible to deposit a backside metal (gold) electrode – using backside lithography – in order to apply the DC bias perpendicularly, since this is the most effective way to change graphene’s surface conductivity.

2.4 GRAPHENE PARAMETERS EXTRACTION AND EQUIVALENT C IRCUIT A simple comparison between experiments and simulation (full wave FEM electromagnetic solver) of the graphene antenna allows for an extraction of the graphene surface impedance. This comparison and the trend of graphene impedance values are reported in Fig. 10.

Fig. 10. Comparison between experiments (bold lines) and measures (light lines). In the above curves, graphene is assumed as a simple resistance. By making use of an equivalent circuit RLC circuit model of the antenna, as recently published by us [M. Dragoman et al. APL, vol. 106, Issue 15, 2015, 153101], we obtain a more accurate model: the equivalent circuit is a R-L-C series circuit, where R is the grapheen’s surface resistance and L is the graphene surface kinetic

(c)


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inductance. In the unbiased case, L =0.3 nH, corresponding to a reactance at 10 GHz equal to X=ωL=19 Ω. Further, R is 120 Ω, for unbiased case, and C=1.104 pF is the total calculated capacitance between graphene and the underlying metallic surface. Situation is depicted in Fig. 11. Fig. 11. Equivalent circuit of the graphene antenna.

2.5 X BAND SLOT ANTENNA LOADED WITH A GRAPHENE PATC H: E.M. MODEL AND SIMULATION RESULTS

Slot antennas, loaded with graphene patches, placed between the metallic strip and the surrounding ground plane, in different configurations, were described in D3.6. In the current reporting period the configuration with one graphene patch placed opposite the coplanar waveguide (CPW) feed line was optimized and test structures were fabricated and characterized. The results were presented at the European Microwave Conference in 2015 [1]. In this configuration, the bias is no longer applied using a back gate approach. Instead, the bias is applied directly through the CPW feed. By varying the conductivity of the graphene patch the electric field distribution inside the antenna slot can be controlled. A fully parametric 3D electromagnetic model was developed using CST MWS (Fig. 12). A 0.2 x 0.2 mm2 graphene patch is placed in the slot, between the metal strip and the surrounding ground plane, opposite the CPW feed line. The bias is applied at the same time as the RF signal, through the ground-signal-ground (G-S-G) probes of the Vector Network Analyzer (VNA), with the aid of a bias tee. The graphene was modeled as a tabulated surface impedance, with the sheet resistance and reactance defined as parameters which could then be varied during the simulations. The substrate is high resistivity (ρ = 5 kΩcm; εr = 11.9) silicon of 525 µm thickness and the chosen metallization is gold (σ = 4.37·107 S/m). The structure is excited with a multipin rectangular waveguide port.

Fig. 12. Electromagnetic model and general layout dimensions of the X band slot antenna with a graphene patch (all

dimensions are given in millimeters).


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Fig. 13a shows the simulated S11 parameter as a function of frequency for various sheet resistance (Rsq

parameter) values. The reactance (Xsq) was considered negligible, since the operating frequency is only

10 GHz [2]. Fig. 13b shows a detail of the resonance frequency, demonstrating the tuning effect of the

graphene patch.

(a) (b)

Fig. 13. Simulated S11 parameter for various sheet resistances (Rsq): (a) wideband frequency response; (b) detail of the tuning effect.

Other important antenna parameters are the radiation efficiency (Fig. 14a) and the antenna gain (Fig. 14b). Even though the efficiency deteriorates when Rsq = 250 Ω/square it is still better than 30% at 10 GHz. Fig. 14c shows the |Z11| parameter simulated for various sheet resistance values of the graphene patch.

(a) (b) (c)

Fig. 14. Simulation results for different sheet resistance values of the graphene patch: (a) antenna efficiency – linear

scale; (b) maximum gain; (c) |Z11| parameter.

Since the slot is the main radiating element and the graphene patch is only 0.2 x 0.2 mm2 the radiation is not severely impeded by it, with gains as high as 3 dBi for a high sheet resistance (1250 Ω/sq). This is also proven by the high antenna efficiency (>60% at 10 GHz for Rsq = 1250 Ω/sq). The simulated 3dB gain bandwidth is between 7.6 – 12.2 GHz. Fig. 15 shows the E-Plane (defined along the feed line) and the H plane (orthogonal to the E plane) for two values of the sheet resistance. Since the gain also takes into account the mismatch losses, the antenna is not only tuned with respect to the input matching, but also with respect to its radiation capability.


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Fig. 15. E and H planes at 10 GHz for the sheet resistance (Rsq) of the graphene patch equal to 1000 Ω/sq (red trace)

and 250 Ω/sq (blue trace).

There are two symmetrical main lobes, with no sidelobes. This is due to the fact that the high resistivity silicon substrate has a thickness of only 525 µm, much smaller than the wavelength in the dielectric (~8.7 mm). Test structures were fabricated on a 525 µm thick high resistivity (ρ = 5 kΩcm) silicon wafer. A silicon oxide layer of ~0.3 µm was grown through thermal oxidation, and graphene was fabricated by Graphenea (http://www.graphenea.com/) through chemical vapor deposition (CVD) and transferred over the whole Silicon wafer. Next, the graphene is patterned through reactive ion etching, followed by metal deposition and patterning. A thin (50 nm) Palladium layer is deposited first as adhesion layer, followed by a 0.3 µm Gold layer. A photo of the fabricated structure is shown in Fig. 16, with a detail of the graphene patch. The graphene patch was contacted through a metal-graphene overlap 0.2 x 0.02 mm2 in size. The antenna structure was characterized on-wafer using a Anritsu 37397D VNA with a PM5 Suss Microtec on wafer probing station. A standard on wafer short-open-load-thru calibration (1601 measurement points, four measurements averaged per point) was performed for the following frequency ranges: 0.04 – 20 GHz and 9 – 11 GHz. The antenna was placed on absorbent material in order to reduce the effects of the back-side lobe. The bias voltage was applied using the internal bias tee of the VNA and a Keithley 4200-SCS (Semiconductor Characterization System). From DC current-voltage measurements the following differential resistances were measured: 700 Ω/sq at 0 V; 757 Ω/sq at -10 V and 833 Ω/sq at -20 V.

Fig. 16. Photo of the fabricated structure and a detail of the graphene patch.

Fig. 17 shows the measurement results for the three bias voltages, with a detail of the resonance

frequency. The tunability effect is demonstrated by the experiments, with a 20 dB difference in amplitude

of S11 and a 7.5 MHz frequency shift, as detailed in Table II.

Table II. Resonance Frequencies as a Function of Applied Voltage

Graphene

patch


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Resonance

frequency

[GHz]

|S11| [dB] Applied voltage

9.85 GHz –42.78 dB 0 V

9.845 GHz –62.3 dB –10 V

9.8425 GHz –46.05 dB –20 V

Fig. 17. Measured |S11| parameter for different biasing voltages.

In order to estimate the radiating properties of the antenna, a transmission measurement was also performed. The second probe of the VNA was suspended over the center of the antenna and the transmission was recorded. The reference trace is the recorded noise level of the VNA, for the two probes in the same position, but without contact to the antenna. The results are shown in Fig. 18a. Since the difference in sheet resistance is quite low (from 700 Ω/sq to 833 Ω/sq) and the value still large, the transmission is only slightly affected by the bias voltage. Fig. 18b shows a detail of the transmission characteristic, showing a difference of ~1 dB between the 0 V and –20 V bias (in good agreement with the simulations from Fig. 14b). Lower sheet resistances would lead to suppression of the radiation of the antenna in the X band. This could lead to a control of the maximum transmission range.

(a) (b)

Fig. 18. Measurement transmission characteristic for different biasing voltages: (a) compared to reference transmission; (b) detail of transmission characteristic at different bias voltages.

References [1] A-C. Bunea, D. Neculoiu, M. Dragoman, G. Konstantinidis, G. Deligeorgis, “X Band Tunable Slot Antenna With Graphene Patch,” Proceedings of the 45th European Microwave Week, EuMW2015, Paris, France, 6 – 11 September 2015, pp. 614-617. [2] Perruisseau-Carrier, J.; Tamagnone, M.; Gomez-Diaz, J.S.; Carrasco, E., "Graphene antennas: Can integration and reconfigurability compensate for the loss?," Proceedings of the 43th European Microwave Week, EuMW2013, Nuernberg, Germany, 6 – 10 Oct.

9.8 9.82 9.84 9.86 9.88 9.9Frequency (GHz)

-65

-55

-45

-35

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|S1

1| [

dB

]

0V

-10V

-20V

9 9.5 10 10.5 11Frequency (GHz)

-100

-90

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-60

|S21

| [dB

]

0V

-10V

-20V

Ref

9.5 9.7 9.9 10.1 10.3Frequency (GHz)

-70

-68

-66

-64

-62

-60

|S21

| [dB

]

9.9 GHz-65.04 dB

9.8 GHz-64.19 dB

9.9 GHz-63.91 dB

0V

-10V

-20V


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3 EXPERIMENTAL CHARACTERIZATION OF THE CNT ANTENNA

3.1 CNT BUNDLE: PARAMETER EXTRACTION In this section, we report on MWCNT bundles as layers of electromagnetic material, for properties identification: their effective permittvity is extracted for different layer thicknesses, from S-parameters measurements. In Fig. 19, the three configurations of the standard calibration approach, adopted here, is reported schematically [G. Vincenzi et. al., “Open-Thru de-embedding for Graphene RF devices” in IEEE-IMS Tampa, 2014]. In Fig. 20, the calibrated results for real and imaginary parts of the dielectric constant of a MWCNT bundle layer of 20 µm are reported: the frequency behavior has been analyzed up to 100 GHz. Fig. 19. Independent measurements needed to perform the e.m. calibration of a MW-bundle layer.

Fig. 20. CNT bundle experimental material electromagnetic properties (real and imaginary part of the effective dielectric constant).

DUT


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3.2 INSERTION LOSS CHARACTERIZATION In the following figures we report about experimental characterization of the CNT-based antenna (global behaviour analysis in frequency). In Fig. 21, we show the input matching of the antenna, and correponding input impedance, in order to compare the device behavior with and without the CNTbundle: in this case the latter is directly fed by a tapered coplanar waveguide.

Fig. 21. Measure of CNT antenna respose with and without a CNT (ΦCNT = 10 µm – LCNT = 50 µm), at the end of a

tapered CWG feeding.

In addition, CPW RF feeding is designed with optional series or shunt capacitive coupling, like we show in Fig. 22.

Fig. 22. Different taper terminations of the feeding CWG.

Direct coupling

Shunt capacitive coupling

Series capacitive coupling


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Fig. 23. Measure of CNT antenna respose with and without a CNT (ΦCNT = 10 µm – LCNT = 50 µm,) at the end of a CWG taper with serial capacitive coupling.

It is clear that the presence of the CNT changes significantly the matching properties of the antenna. In Fig. 23 and Fig. 24, we show the same results as in Fig. 21, but in this case the CNT bundle is fed by a tepered coplanar waveguide with, respectively, serial and parallel capacitive coupling addition. We observe, by comparison of figures 21, 23 and 24, that the largest effect by the CNT, on the input matching of the antenna, is achieved in case of serial capacitive coupling (Fig. 23).

Fig. 24. Measure of CNT antenna respose with and without a CNT (ΦCNT = 10 µm – LCNT = 50 µm), at the end of a CWG taper with parallel capacitive coupling. To conclude this section, a comparison of different results, obtained in different experimental runs by using different CNT bundles, is reported in Table III. Results at 10 GHz are “expected” as they are still


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under characterization. Here λ0/4 is the radiation length of a metal antenna of the same size of the CNT bundle (see section 3.4). Table III Aspect Ratio Diameter [µm] Length [µm] Radiation length

@ 60 GHz

Radiation length

@ 10 GHz

[expected]

Run1 10 50 λ0/100 λ0/600

Run2 20 120 λ0/40 λ0/250

Run3 70 490 ~ λ0/10 ~ λ0/60

3.3 Radiation pattern characterization In this section the radiation pattern of the CNT antenna, in serial capacitive-coupling configuration, is reported. Measurements are made in far-field conditions (see Fig. 25) for a working frequency of 60 GHz (2*d²/λ). The analysis includes: i) angular transmission between CNT-based antenna and calibrated receiver (horn antenna - WR12), ii) transmission calibration with two reference horns (WR12), iii) cross-polarization validation. Details are reported in Fig. 26. Fig. 25. Measurement of the radiation pattern of the CNT antenna: experimental setup.


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Fig. 26. Measurement of the radiation pattern of the CNT antenna.

3.3 MODELLING OF A 10 GHZ CNT ANTENNA WITH INDUCTIV E COUPLING

The imaginary part of the CNT bundle impedance, after de-embedding, seems to suggest negative values for the equivalent inductance, in the range 1 − 5 nH at 10 GHz. Including an inductive stage to compensate for the above inductance leads to the simulation design shown in Fig. 27. Numerical results are also reported.

a)


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(b) Fig. 27. a) Inductive termination of the CWG taper, b) reflection coefficient of the antenna with and without CNT.

4 CONCLUSION

4.1 GRAPHENE ANTENNA In Table IV, we summarize the achievements of the radiating elements based on graphene: the NANORF goals, i.e. central frequency within 2-80 GHz, -20 dB return loss, beamwidth 70°, have been obtained by our implemented device. Table IV

Although the radiation efficiency is generally lower than that of a metallic antenna, due to ohmic losses, the graphene antenna is a wideband antenna in comparison to a metal antenna with the same geometry and working at the same frequencies. Even more importantly, the possibility of tuning of the input matching and the radiation by an external DC voltage has been experimentally verified, especially for the case of the X band slot antenna loaded with a graphene patch.

Specification State-of-the-art NANORF goal

Frequency: 10 GHz

None ACHIEVED

|S11| Maximum: -10 dB Minimum: -20 dB

None ACHIEVED

Beamwidth <70° None ACHIEVED


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4.2 CNT ANTENNA The following Table V summarize the main achievements of the CNT antenna using, as an example, the case of CNT bundle diameter = 10 µm and length = 50 µm. In this case, the radiation at 60 GHz may suggest that the slow-wave effect decreases the resonant wavelength λ of a CNT antenna, in comparison to the one of a metal antenna, λ0, by a factor of 25: λ0=c/f=3*10^8/(60*10^9)=25*(4*L)=25* λ. Table V – Frequency = 60 GHz.

For frequency around 10 GHz, the experimental data are now being derived: on going work aims at final radiative device, implementing a complementary inductive part (with TE mode excitation).

Specification State-of-the-art

NANORF achievement

Frequency : 2-80 GHz Not reported Achieved

|S11|: -30 dB Not reported -25 dB

Beamwidth <70° Not reported Achieved

Bandwidth 1 -15 GHz Not reported 5.5 GHz

d4.4 - activity report on cnt and graphene based … · canseliet 1 ex national research and...

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