dc voltage control of inverter interfaced dual active

DC Voltage Control of Inverter Interfaced Dual

Active Bridge Converter for V2L applications

Yunting Liu

CURENT, Department of EECS

The University of Tennessee

Knoxville, TN, USA

[email protected]

Xiaorui Wang

Department of ECE

Michigan State University

East Lansing, MI, USA

Wei Qian

Department of ECE



Ameer Janabi

Department of ECE



Bingsen Wang

Department of ECE



Xi Lu

Research and Innovation Center

Ford Motor Company

Dearborn, MI, USA

Ke Zou


Ford Motor Company

Dearborn, MI, USA

Chingchi Chen


Ford Motor Company

Dearborn, MI, USA

Abstract—This paper proposed a dc voltage control for

inverter interfaced dual active bridge converter. This converter is

used in SiC based on-board charger for electric vehicle. The

inverter interfaced dual active bridge converter has a dc link as a

middle stage of AC/DC conversion. This dc-link voltage is unstable

without control. Especially in applications like vehicle-to-load

(V2L) applications, the converter is required to operate under

both 120-volt ac load and 480-volt ac load so the vehicle can export

power to external loads. The large variation of dc link voltage

requires a robust general dc voltage control to guarantee a safe

operation of load change. This paper analytically derives the

small-signal model of the inverter interfaced dual active bridge

converter at wide range of operating point. Based on the small-

signal model, a robust dc voltage control is proposed to adapt to

the wide range of dc link voltage. The proposed dc voltage control

is verified by a 25-kW SiC based inverter interfaced dual active

bridge converter prototype.

Keywords—inverter interfaced Dual Active Bridge (DAB), dc

voltage control, small-signal model, vehicle to load (V2L)

I. INTRODUCTION

The inverter interfaced dual active bridge (DAB) converter has been widely used in electric vehicle on-board charger applications[1]. The inverter interfaced DAB converter consists of a three-phase front-end and two H-bridges as shown in Fig.1. The front-end three-phase inverter interacts with ac grid. The feature of this topology is that it has a wide range of operation and has a chance to adapt to multi-functions. This inverter interfaced DAB converter has been proposed with multi-functions in [2]. The converter discussed in [2] performed various functionalities such as battery charging (G2V), vehicle to grid (V2G) regenerative capability, and vehicle-to-load (V2L) feature. With the V2L function, the converter can output different types of voltage, such as 120-volt single-phase and 480-volt three-phase.

However, the multi-function concept increases the complexity of DAB control. In order to enhance the DAB with multi-function, we need to consider control algorithms for many operating points. To switch control algorithms from different operating points may consume valuable computation time of DSP. This becomes severe for wide-bandgap (WBG) based converters since normally the control algorithms are required to complete in one switching cycle. The switching frequency of WBG based converters normally falls in the range of 100 kHz to 500 kHz, which is 10 times faster than Si based converters. The fast switching requires the general control for multiple operating points, and even multiple functions. A faster and simpler controller is needed for WBG based inverter interfaced DAB system so that the converter can benefit most from the fast switching of WBG devices.

In scenarios like V2L applications, the converter is required to operate under both 120-volt ac load and 480-volt ac load. The wide range of dc-link voltage (VC1) challenges the control system to adapt to wide range of operating points. However, few literatures discussed the dc voltage control especially in V2L applications. Many existing literatures discussed the small-signal model of DAB converters[3]–[5]. A few other papers discussed the small-signal model of inverter-interfaced DAB converters[6], [7]. The focus of existing literatures is to develop a general model of DAB to understand the dynamics of DAB. Some control algorithms are established on the small-signal model for DAB[8]–[11]. The control objectives are normally transformer current or output voltage. The dc bus voltage control

Fang Z. Peng

Center for Advanced Power Systems

Florida State University

Tallahassee, FL, USA

HV

Batt

P1

P2

P3

P4

Q1

Q2

Q3

Q4

S1

S2

S3

S4

C1 C2

Ls

Lleak

1:1

Lbat

Vbat

Q3

Q4

Fig. 1. The inverter interfaced DAB converter.

978-1-7281-3761-2/19/$31.00 ©2019 IEEE 319

Authorized licensed use limited to: Michigan State University. Downloaded on December 04,2020 at 18:51:40 UTC from IEEE Xplore. Restrictions apply.

for inverter interfaced DAB has been discussed in [12], [13]. The voltage control in these papers are too complicated to adapt to multi-function converters.

This paper proposed a general dc voltage control for inverter interfaced DAB converter that is suitable for a wide range of operating points. This paper is organized as follows. The small-signal model of inverter interfaced DAB converter that considers the primary side bus voltage as an independent state variable is derived in Section II. Based on the small-signal model of the converter, the wide range operating point for multi-functions are assessed in Section III. A robust dc voltage control is proposed in Section IV to cover all the operating points for the converter. Some experiment results to verify the effectiveness of the proposed method are presented in Section V.

II. SMALL-SIGNAL MODEL OF INVERTER INTERFACED DAB DC

VOLTAGE DYNAMICS

To simplify the small-signal modeling of the converter, the topology is reduced to single phase as shown in Fig.2. To further simplify the analysis, we assume that Lbat is small and the C2 voltage is the same as the battery voltage. Therefore, VC2 is no long a state variable. The high-frequency transformer model is replaced by the power transfer function,

𝑃 =𝑛𝑉C1𝑉C2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝐷(1 − 𝐷) ()

Where 𝑛 = 1 is the turns ratio of the transformer for this paper. 𝑓𝑠 is the switching frequency. 𝑉C1 and 𝑉C2 are the capacitor voltages. 𝐷 is the phase shift between primary voltage and secondary side voltage. The single-phase shift control is adopted in this paper. the DAB is modeled with the power transfer function in this paper because the high frequency transformer is externally presenting a power transfer function in lower frequency domain. If considering the internal dynamic of the high frequency transformer, the conventional averaging model is not suitable since the averaging window (one switching cycle) will eliminate all the high frequency components. For example, the leakage inductance current becomes zero after averaging. This makes the averaging model unsuitable for internal dynamics of the DAB. C1 voltage 𝑣𝐶1 and Ls current 𝑖𝑠 are selected as state variables in this paper. The state equations are,

{

𝑝 =

𝑣𝐶1𝑣𝐶2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝐷(1 − 𝐷)

𝑝 = 𝑝0 + 𝑝𝐶1

𝑝𝐶1 = 𝐶1𝑣𝐶1𝑑𝑣𝐶1

𝑑𝑡

𝑝0 = 𝑖𝑠𝑣𝑠

𝐿𝑠𝑑𝑖𝑠

𝑑𝑡= 𝑞𝑣𝐶1 − 𝑣𝑠

(2)

Where 𝑞 is the equivalent duty cycle of H-bridge Q1-Q4. Eq.(2) can be simplified as

{𝐶1𝑣𝐶1

𝑑𝑣𝐶1

𝑑𝑡=

𝑣𝐶1𝑣𝐶2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝐷(1 − 𝐷) − 𝑖𝑠𝑣𝑠

𝐿𝑠𝑑𝑖𝑠

𝑑𝑡= 𝑞𝑣𝐶1 − 𝑣𝑠

(3)

Replace 𝑥 = �̃� + 𝑥𝑒 where 𝑥 = 𝐷, 𝑞 , 𝑣𝐶1 or 𝑖𝑠 . 𝑥𝑒 is the state variable values at steady state.

{

𝐶1(�̃�𝐶1 + 𝑉𝐶1)

𝑑(�̃�𝐶1+𝑉𝐶1)

𝑑𝑡

=(�̃�𝐶1+𝑉𝐶1)𝑣𝐶2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘(�̃� + 𝐷) (1 − (�̃� + 𝐷)) − (𝑖�̃� + 𝐼𝑠)𝑣𝑠

𝐿𝑠𝑑(�̃�𝑠+𝐼𝑠)

𝑑𝑡= 𝑞(�̃�𝐶1 + 𝑉𝐶1) − 𝑣𝑠

(4)

Eq. (4) can be simplified as

{

𝐶1𝑉𝐶1

𝑑(�̃�𝐶1)

𝑑𝑡

=(�̃�𝐶1+𝑉𝐶1)𝑣𝐶2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘(�̃� + 𝐷 − 𝐷2 − 2𝐷�̃�) − (𝑖�̃� + 𝐼𝑠)𝑣𝑠

𝐿𝑠𝑑(�̃�𝑠)

𝑑𝑡= 𝑞�̃�𝐶1 + �̃�𝑉𝐶1 + 𝑞𝑉𝐶1 − 𝑣𝑠

(5)

Since the steady state has

{0 =

𝑣𝐶1𝑣𝐶2

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝐷(1 − 𝐷) − 𝑖𝑠𝑣𝑠

0 = 𝑣𝑠 − 𝑞𝑣𝐶1 (6)

Therefore,

{𝐶1𝑉𝐶1

𝑑�̃�𝐶1

𝑑𝑡=

𝑉𝐶1𝑣𝐶2(1−2𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃� +

𝑣𝐶2𝐷(1−𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃�𝐶1 − 𝑖̃𝑠𝑣𝑠

𝐿𝑠𝑑�̃�𝑠

𝑑𝑡= 𝑞�̃�𝐶1 + �̃�𝑉𝐶1

(7)

Take Laplace transform to (7),

{

𝑠𝐶1𝑉𝐶1�̃�𝐶1(𝑠) =𝑉𝐶1𝑉𝐶2(1−2𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃�(𝑠) +

𝑉𝐶2𝐷(1−𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃�𝐶1(𝑠) − 𝐼𝑠(𝑠)𝑣𝑠

𝑠𝐿𝑠𝐼𝑠(𝑠) = 𝑞�̃�𝐶1(𝑠) + �̃�(𝑠)𝑉𝐶1

(8)

Therefore,

(𝑠𝐶1𝑉𝐶1 −𝑉𝐶2𝐷(1−𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘+

𝑞𝑣𝑠

𝑠𝐿𝑠)𝑉𝐶1(𝑠)

�̃�(𝑠)=

𝑉𝐶1𝑉𝐶2(1−2𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘 (9)

(𝑠𝐶1𝑉𝐶1 −𝑉𝐶2𝐷(1−𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘+

𝑞𝑣𝑠

𝑠𝐿𝑠)𝑉𝐶1(𝑠)

�̃�(𝑠)=

𝑉𝐶1𝑣𝑠

𝑠𝐿𝑠 (10)

The transfer functions are

𝐻1(𝑠) =𝑉𝐶1(𝑠)

�̃�(𝑠)=

𝑠𝐿𝑠𝑉𝐶1𝑉𝐶2(1−2𝐷)

2𝑠2𝐿𝑠𝐿𝑙𝑒𝑎𝑘𝐶1𝑉𝐶1𝑓𝑠−𝑠𝐿𝑠𝑉𝐶2𝐷(1−𝐷)+2𝑣𝑠2

𝑉𝐶1𝑓𝑠𝐿𝑙𝑒𝑎𝑘

(11)

𝐻2(𝑠) =𝑉𝐶1(𝑠)

�̃�(𝑠)=

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝑉𝐶1𝑣𝑠

2𝑠2𝐿𝑠𝐿𝑙𝑒𝑎𝑘𝐶1𝑉𝐶1𝑓𝑠−𝑠𝐿𝑠𝑉𝐶2𝐷(1−𝐷)+2𝑣𝑠2

𝑉𝐶1𝑓𝑠𝐿𝑙𝑒𝑎𝑘

()

HV

Batt

P1

P2

P3

P4

Q1

Q2

Q3

Q4

S1

S2

S3

S4

C1 C2

Vs Ls Lleak

1:1

Lbat

Vbat

is

Fig. 2. The single-phase inverter interfaced DAB converter.

320


III. OPERATING POINT ASSESSMENT

We can see from (11) that the transfer function 𝐻1(𝑠) is a function of VC1 and D. D may vary from 0 to 0.5. In V2L applications, VC1 may vary from 170 V to 700 V to adapt to 120-V load and 480-V load. The key parameters of the converter that under analysis in this paper is summarized in Table I.

The bode plot of 120-V load transfer function 𝐻1(𝑠) and 480-V load transfer function 𝐻1(𝑠) are shown in Fig.3. The bode plot of 120-V load transfer function 𝐻2(𝑠) and 480-V load transfer function 𝐻2(𝑠) are shown in Fig.4. The phase shift D is discretized in every 15 degree. We can see from the figures that the transfer function does not deviate from each other too much at two operating points. Therefore, we could rely on a controller with identical parameters to regulate the state variables.

IV. ROBUST DC VOLTAGE CONTROL FOR WIDE RANGE

OPERATION

As seen in (8), the state variable �̃�𝐶1(𝑠) is coupled with

𝐼𝑠(𝑠). In order to decompose the state variables, the controller is

designed to assign a fast loop to control 𝐼𝑠(𝑠) and a slow loop to

control �̃�𝐶1(𝑠) . The control objective of the fast loop is ac voltage is real system since the output of converter should be controlled as a voltage source. The control objective remains to

𝐼𝑠(𝑠) in this paper since the test is conducted under resistive load. The output ac voltage is proportional to output current in

this specific scenario. 𝐼𝑠(𝑠)𝑣𝑠 term in (8) could be deemed as zero since Is has reached to steady state in the fast control loop.

Also, 𝑞�̃�𝐶1(𝑠) term in (8) could be deemed as zero since VC1

changes very slow in the fast loop. Therefore, �̃�𝐶1(𝑠) can be

controlled by �̃�(𝑠) independently and 𝐼𝑠(𝑠) can be controlled

by �̃�(𝑠) independently. Eq. (8) can be simplified as

{𝑠𝐶1𝑉𝐶1�̃�𝐶1(𝑠) =


2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃�(𝑠) +

𝑉𝐶2𝐷(1−𝐷)

2𝑓𝑠𝐿𝑙𝑒𝑎𝑘�̃�𝐶1(𝑠)

𝑠𝐿𝑠𝐼𝑠(𝑠) = �̃�(𝑠)𝑉𝐶1

()

The transfer functions are

𝐻(𝑠) =𝑉𝐶1(𝑠)

�̃�(𝑠)=


2𝑠𝑓𝑠𝐿𝑙𝑒𝑎𝑘𝐶1𝑉𝐶1−𝑉𝐶2𝐷(1−2𝐷) ()

𝐺(𝑠) =𝐼𝑠(𝑠)

�̃�(𝑠)=

𝑉𝐶1

𝑠𝐿𝑠 ()

The H-bridge Q1-Q4 are used to control Is. The dual active bridge is used to control VC1. The bode plot of 120-V load transfer function 𝐻(𝑠) and 480-V load transfer function 𝐻(𝑠) are shown in Fig.5. The bode plot of 120-V load transfer function 𝐺(𝑠) and 480-V load transfer function 𝐺(𝑠) are shown in Fig.6.

A PI controller is designed for the VC1 regulation and another PI controller is designed for Is regulation. The control strategy is shown in Fig.7. The parameters of PI controller are summarized in Table II. The open-loop bode plots after implementing the PI controls are shown in Fig. 8-9. We can see from the bode plot that the phase of zero cross point is less than 180 degree for all operating points. This guarantees that the general PI controller is suitable for both 120-V load and 480-V load.

V. EXPERIMENT

To verify the effectiveness of the proposed dc voltage control, the scale-down experiments are conducted. The experiment circuit is similar to Fig.1. The ac terminals are

TABLE I. KEY PARAMETERS

Filtering

inductor,

Ls

10

μH

DC

Capacitor,

C1

3300

μF

Battery

voltage,

VC2

350

V

Leakage

inductance,

Lleak

22

μH

Switching

frequency,

fs

100

kHz

Load

voltage, Vs

120,

480

V

Capacitor

voltage,

VC1

170,

700

V

Phase

shift, D

0 ~

0.5

-100

0

100

200

1 10 102 103 104 105

-270

-180

-90

0

90

Bode Diagram

Frequency (rad/s)

Pha

se (

deg)

Mag

nitu

de (

dB)

H1 for 120-V load

H1 for 480-V load

Fig. 3. The transfer function H1 bode plot for 120-V load and 480-V load.

-100

0

100

200

300

-360

-270

-180

-90

0

Bode Diagram

Frequency (rad/s)

Phase

(deg)

Mag

nit

ude (

dB

)

1 10 102 103 104 105

H2 for 480-V loadH2 for 120-V load

Fig. 4. The transfer function H2 bode plot for 120-V load and 480-V load.

TABLE II. PARAMETERS FOR PI CONTROLLERS

HPI = KP + KI/s

GPI = KP + KI/s

KP KI KP KI

0.01 1 0.001 1

321


connected to resistive loads in wye-connection with resistant of

12 Ω each phase. The battery voltage is 130 V. The load voltage is set to 80 and 140 V. The PI controller parameters for these two operating points are the same. In order to demonstrate that the PI controller parameters are the same, the experiment sets a hot dc link voltage change from 80 V to 140 V and backward. The key parameters of the experiment setup are summarized in Table III. The control is implemented in TI-DSP TMS320F28379D. The CPU clock is 200 MHz. The sampling frequency is 100 kHz. The equivalent PI controller parameters in continuous time domain is defined by,

𝐾𝑃𝐷𝑆𝑃 = 𝐾𝑃

𝐶𝑜𝑛𝑡. ()

𝐾𝐼𝐷𝑆𝑃 = 𝐾𝐼

𝐶𝑜𝑛𝑡. 𝑓𝑠𝑎𝑚⁄ ()

The PI controller equivalent parameters are the same as Table II.

-50

0

50

100

150

-180

-135

-90

Bode Diagram

Frequency (rad/s)

Pha

se (

deg)

Mag

nitu

de (

dB)

1 10 102 103 1040.1

H for 120-V load

H for 480-V load

Fig. 5. The transfer function H(s) bode plot for 120-V load and 480-V load.

-100

-50

0

50

100

150

-180

-135

-90

Bode Diagram

Frequency (rad/s)

G×GPI for 120-V load

G×GPI for 480-V load

10 102 103 104 105 106

Pha

se (

deg)

Mag

nit

ude (

dB)

Fig. 9. The open-loop bode plots after implementing the PI control for Is

under 120-V load and 480-V load.

PI+

–

VC1*

VC1 Vdc1

ia

ib

ic

+

+

+

ia*

ib*

ic*

PI

–

–

–

ia,b,c

Slow loop

Fast loop

( )H s

Q

D

( )G s

Fig. 7. Complete control strategy.

-100

-50

0

50

100

150

-270

-180

-90

Bode Diagram

Frequency (rad/s)

Phase

(deg)

Mag

nit

ude (

dB)

1 10 102 103 1040.1

H×HPI for 120-V load

H×HPI for 480-V load

Fig. 8. The open-loop bode plots after implementing the PI control for VC1

under 120-V load and 480-V load.

-50

0

50

100

150

-91

-90.5

-90

-89.5

-89

Bode Diagram

Frequency (rad/s)

G for 120-V load

G for 480-V load

102 103 104 105 106 107

Phase

(deg)

Mag

nit

ude (

dB

)

Fig. 6. The transfer function G(s) bode plot for 120-V load and 480-V load.

TABLE III. KEY PARAMETERS OF EXPERIMENT SETUP

Filtering inductor, Ls 10 μH

DC Capacitor, C1 3300 μF

Battery voltage, VC2 130 V

Leakage inductance, Lleak 22 μH

Switching frequency, fs 100 kHz

Load voltage, Vs 50, 90 V

Capacitor voltage, VC1 80, 140 V

Load resistance, Rabc 12 Ω

322


The corresponding experiment results are shown in Fig. 10-13. VPrim is the primary side voltage of transformer which is close to C1 bus. VSec is the secondary side voltage of transformer which is close to C2 bus. Ileak is the transformer current which is measured on primary side. Fig.10 shows the dc voltage jumps from 140-volt to 80-volt under the proposed controller. Fig.11 is the zoom-in details of the jump-down dynamics with the leakage inductance current waveform. Fig.12 shows the dc voltage jumps from 80-volt to 140-volt. Fig.13 is the zoom-in details of the jump-up dynamics with the leakage inductance current waveform. The experiments indicate that the controller parameter is validated to both operating points of the dc-link voltage.

VI. CONCLUSION

This paper proposed a robust dc voltage control for inverter interleaved DAB. The small-signal model of the inverter interfaced DAB converter is derived at a wide range of operating point. Based on the small-signal model, a robust dc voltage control is proposed to adapt to the wide range of dc link voltage. The proposed dc voltage control is verified by a SiC based inverter interfaced dual active bridge converter prototype. The experiment results showed that the proposed control could cover a wide range of dc link voltage.

ACKNOWLEDGMENT

The authors would like to thank Ford Motor Company for the technical and fellowship support.

REFERENCES

[1] D. Sal, D. Frey, J. Schanen, and J. Ferrieux, “Isolated Single Stage Bidirectional AC-DC converter with power decoupling and reactive

power control to interface battery with the single phase grid,” 2018 IEEE Appl. Power Electron. Conf. Expo., pp. 631–636, 2018.

[2] X. Wang et al., “A 25kW SiC Universal Power Converter Building Block

for G2V, V2G, and V2L Applications,” 2018 IEEE Int. Power Electron. Appl. Conf. Expo., pp. 1–6, 2018.

[3] H. Qin and J. W. Kimball, “Generalized average modeling of dual active bridge DC-DC converter,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 2078–2084, 2012.

[4] W. Han and L. Corradini, “Analytical Small-Signal Transfer Functions

for Phase Shift Modulated Dual Active Bridge Converters Using Phasor Transformation,” 2018 IEEE Energy Convers. Congr. Expo., vol. 1, pp. 1442–1448, 2018.

[5] S. S. Shah and S. Bhattacharya, “A simple unified model for generic

operation of dual active bridge converter,” IEEE Trans. Ind. Electron., vol. 66, no. 5, pp. 3486–3495, 2019.

[6] H. K. Krishnamurthy and R. Ayyanar, “Building block converter module

for universal (AC-DC, DC-AC, DC-DC) fully modular power conversion architecture,” PESC Rec. - IEEE Annu. Power Electron. Spec. Conf., pp. 483–489, 2007.

[7] B. Singh, “Modelling of Inverter Interfaced Dual Active Bridge

Converter,” 2018 Int. Conf. Comput. Power Commun. Technol., pp. 680–685, 2018.

VC1 Load change

Vab,bc

Ia

Fig. 10. DC voltage jumps from 140-volt to 80-volt under the proposed

controller.

Load changeVC1

VSec

VPrim

Ileak

Fig. 11. The zoom-in details of the jump-down dynamics with the leakage

inductance current waveform.

Load changeVC1

Vab,bc

Ia

Fig. 12. DC voltage jumps from 80-volt to 140-volt under the proposed

controller.

Load changeVC1

VPrim

VSec

Ileak

Fig. 13. The zoom-in details of the jump-up dynamics with the leakage

inductance current waveform.

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[8] D. D. Nguyen, G. Fujita, Q. Bui-Dang, and M. C. Ta, “Reduced-Order Observer-Based Control System for Dual-Active-Bridge DC/DC Converter,” IEEE Trans. Ind. Appl., vol. 54, no. 4, pp. 3426–3439, 2018.

[9] Z. Yu, J. Zeng, J. Liu, and F. Luo, “Terminal sliding mode control for

dual active bridge DC-DC converter with structure of voltage and current

double closed loop,” ANZCC 2018 - 2018 Aust. New Zeal. Control Conf., pp. 11–15, 2019.

[10] F. Xiong, J. Wu, Z. Liu, and L. Hao, “Current Sensorless Control for Dual Active Bridge DC-DC Converter with Estimated Load-Current

Feedforward,” IEEE Trans. Power Electron., vol. 33, no. 4, pp. 3552–3566, 2018.

[11] W. Song, N. Hou, and M. Wu, “Virtual Direct Power Control Scheme of Dual Active Bridge DC-DC Converters for Fast Dynamic Response,” IEEE Trans. Power Electron., vol. 33, no. 2, pp. 1750–1759, 2018.

[12] M. Ali, M. Yaqoob, L. Cao, and K. H. Loo, “Disturbance-Observer-Based

DC-Bus Voltage Control for Ripple Mitigation and Improved Dynamic

Response in Two-Stage Single-Phase Inverter System,” IEEE Trans. Ind. Electron., vol. 66, no. 9, pp. 6836–6845, 2019.

[13] Y. Guo, W. Ma, J. Meng, and Y. Wang, “A Virtual Inertia Control Strategy for Dual Active Bridge DC-DC Converter,” 2nd IEEE Conf. Energy Internet Energy Syst. Integr. EI2 2018 - Proc., pp. 1–5, 2018.

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dc voltage control of inverter interfaced dual active

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