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Millimeter-Wave Antennas and Arrays

Wonbin Hong*Samsung Electronics, Suwon, Republic of Korea

Abstract

The growing interest associated with exploiting the millimeter-wave spectrum for future wirelesscommunication devices and networks has rendered the ensuing need for in-depth research and develop-ment of advanced millimeter-wave antennas. This chapter provides an overview of the background andmotivation of ultrafast, low-latencymillimeter-wave wireless applications as well as key characteristics ofthe spectrum. Major antenna design considerations and techniques are introduced and discussed in detail.The discussion is followed with state-of-the-art millimeter-wave antenna design techniques applicable tonext-generation wireless devices. The chapter is concluded with discussions of future directions.

Keywords

Millimeter waves; Planar antennas; Beam steering; Phased array; 60 GHz; 5G communications

Introduction

Millimeter waves (mmWaves) are collectively defined as frequency spectrums in which the free-spacewavelength can be described using millimeter units. Formally denoted as an extremely high frequency(EHF) by the International Telecommunications Unit (ITU), the spectrum ranges from 30 to 300 GHz.The very concept of mmWave radio technology commands over a century-long history with earlyapplications in radio astronomy and military applications half a century ago. However, to the majorityof the demographic, mmWave remained a distant, obscure terminology until recent years.

It was not until the relatively recent remarkable technological advancements in silicon and compoundsemiconductors, coupled with the relentless demand for sophisticated wireless data infrastructure, thatmmWave technologies gained traction among diverse technology circles. As of present, potentiallydisruptive technologies, such as extreme high-data-rate low-latency wireless communication (Chen andNiknejad 2011; Emami et al. 2011; Ghosh et al. 2014; Roh et al. 2014; Talwar et al. 2014), advancedresolution radar imaging (Valdes-Garcia et al. 2013), and inter- and intra-chip communication networks(Baek and Hong 2013), are primary examples of how mmWave radio and radars are reshaping theparadigm of wireless technologies. For each of these applications, increasing the operating frequency byseveral orders of magnitude is a viable and an effective solution to the current worldwide frequencyspectrum overcrowding issues below 6 GHz. Moreover, the channel capacity denoted in bits per secondsestablished in the classical work by Shannon and Hartley (Shannon 1948) is directly proportional to thechannel bandwidth. For instance, the short-range, high-data-rate IEEE 802.11ad WiGig standard canachieve PHY (physical layer) rates that are ten times or more higher than conventional Wi-Fi by utilizingmore than 2 GHz of channel bandwidth centered around the unlicensed 60 GHz IMS (industrial, medical,scientific) band. The latest 4G Long-Term Evolution (LTE) cellular technology continues to become

*Email: [email protected]

Handbook of Antenna TechnologiesDOI 10.1007/978-981-4560-75-7_64-1# Springer Science+Business Media Singapore 2015

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increasingly sophisticated and efficient using innovative technologies such as carrier aggregation (CA),single-user and multiuser multiple-input multiple-output (SU-MIMO/MU-MIMO), and heterogeneousnetworks (HetNets) using macro and small cells. Encouraged by notable advancements in cost-efficientmmWave RFIC technologies (Natarajan et al. 2009; Cohen et al. 2013; Tsukizawa et al. 2013; Kong et al.2013; Bereka et al. 2014; Okada et al. 2014), ultra-wideband mmWave wireless communication technol-ogies that can augment LTE networks are being actively researched and experimented by numerousresearch institutions and private sectors (Sun et al. 2014; Taori and Sridharan 2014; Chin et al. 2014; Honget al. 2014a). By exploiting the large available bandwidth at mmWave, low real-time latency access andbackhaul networks with more than 10Gbps peak data rates are projected to be the backbone of future fifth-generation wireless communications.

The benefits of mmWave technologies do not come without challenges. The inherent difference atmmWave compared to frequencies below 10 GHz demands unique design considerations in hardwarecircuitry. The antenna is one of the most critical elements that affect the overall performance of mmWaveradios. In this chapter, section “Theory” will first discuss and understand the unique nature of mmWavespectrum and major factors that must be taken into consideration during the antenna design. Afterward,key mmWave design techniques and schematics will be presented in detail in section “Key mmWaveAntenna Design Techniques”. Section “Notable mmWave Antenna Applications” will exemplify notablemmWave antenna topologies currently employed in high-volume applications such as consumer elec-tronics and will discuss the top-down design process. Section “Conclusion” presents a number ofimportant future trends in mmWave antenna technology, and the chapter is concluded.

Theory

Key Properties of mmWavesAt mmWave spectrum, there are several notable loss factors that will significantly affect the behavior ofradio transmission and reception. A number of key propagation loss factors that affect radio signaltransmission and reception in the following will be first identified and discussed.

Free-Space Propagation LossFree-space propagation loss is defined as the propagation loss that is dependent on the distance andoperating frequency of two isotropic transmit and receive antennas. Equation 1 describes the free-spacepropagation loss in closed-form expression,

Free� Space Propagation Loss ¼ 4pRl

� �2

(1)

where R is the distance between transmit and receive antennas and lambda is the free-space wavelength atthe operating frequency. Equation 1 can be rewritten in dB scale and frequency for better understanding asshown in Eq. 2:

Free� Space Propagation Loss dBð Þ ¼ 20logF þ 20logRþ 92:4 (2)

where frequency F is expressed in GHz and distance R is expressed in kilometers (km). It can be intuitivelyinterpreted that the free-space path loss will increase in logarithmic scale of the operating frequency. Theeffect of free-space propagation loss on the radio link budget can be visualized using the following Eq. 3,where PRX is defined to be the receiving power (dB) and PTX is defined to be the transmitting power (dB):


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PRX ¼ PTXl

4pR

� �2

(3)

Precipitation LossAt mmWave frequencies, the sizes of the raindrops are typically within several multitudes of the free-space wavelength. This causes noticeable scattering during radio signal transmission in the presence ofrain. Figure 1 presents the radio signal attenuation per kilometer (km) as a function of rain rate and rainintensity. As it can be observed, radio signals below 10 GHz generally experience limited levels ofattenuation as a function of rain rate. In contrast for mmWave radio transmission, the specific region andthe yearly rain rate must be taken into account from the beginning of the signal link budgeting and theantenna design as this is one of the major factors that account for signal fading.

Atmospheric Absorption and Gaseous LossAtmospheric constituents tend to absorb radio signals at much higher rates at mmWave frequenciescompared to lower microwave frequencies. This tendency is loosely proportional to signal frequency, anda number of notable attenuation peaks can be observed as presented in Fig. 2. This can be explained by thenatural mechanical resonance behaviors of oxygen (O2) molecules and water (H2O) molecules at certainmmWave frequencies. For example, the oxygen absorption rates at 30 and 60GHz are 0.02 and 15 dB/km,respectively. The O2 molecules reach maximum absorptions around 60.3 GHz. The inherent nature ofsevere gaseous losses at certain mmWave frequencies are the primary factors behind majority of thedevised close-range communication scenarios. Otherwise, spectral regions within the mmWave fre-quency bands with limited gaseous losses commonly denoted as “atmospheric windows” are utilizedfor transmissions requiring midrange and above scenarios.

100

150 mm/h100 mm/h

50 mm/h

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Frequency (GHz)

Spe

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50 100 200 500 1,000

Fig. 1 Precipitation loss as a function of rain rate (Marcus and Pattan 2005)


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Foliage LossLosses incurred by foliage can significantly affect long-term fading of the mmWave channel. Foliage lossillustrated in Fig. 3 is dependent on the density of the foliage, the tree species, humidity, etc., andnon-negligible (Marcus and Pattan 2005). The empirical foliage loss can be expressed as Eq. 4 as followswhere f is defined as the operating frequency inMHz andD is denoted as the depth of foliage transverse inmeters when D < 400 m:

Lfoliage ¼ 0:2f0:3D0:6 dB (4)

Material LossIt is imperative to carefully consider signal attenuation loss incurred by various materials during thedesign stage. With few exceptions, radio signals beyond 5 GHz are subject to an exponential increase inattenuation as the signal waves propagate through the stated material. This tendency is presented in Fig. 4.Consequently, proper material selection becomes one of the most critical design aspects for mmWaveantennas as the material composition of the antenna substrate, the radome, and the connectors is closelylinked to the radiation efficiency. Furthermore, the importance of low-loss interconnects and antennafeeding and packaging technologies is emphasized to minimize attenuations of hard-fought mmWave

402010

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Fig. 2 Atmospheric absorption (Marcus and Pattan 2005)

Fig. 3 Illustration of foliage loss


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signals that were received and generated by the antenna and the RFIC, respectively. It is imperative thatantenna and mmWave radio link designers must be accurately aware of the loss tangent properties ofmaterials that the radio waves are likely to encounter during propagation scenarios. As it is not commonfor majority of the material vendors and manufacturers to possess material properties beyond 10 GHz,permittivity and loss tangent extraction and calculation techniques (El Sabbagh et al. 2004; Fratticcioliet al. 2004; Fang et al. 2004) must be considered and selectively applied for each design scenario.

Antenna GainAs it can be observed from the previous subsection, notable levels of signal attenuation from numerousfactors are one of the most compelling properties at the mmWave frequency spectrum. In order to establishsuccessful radio transmission, the collective signal attenuation must be less than the radio link budget. Theradio link budget can be expressed as described in Eq. 5 below:

Radio Link Budget dBð Þ ¼ PTXþ GTXþ GRX� Rsensitivity dBð Þ (5)

Rsensitivity is defined as the minimum received power that is detectable for the specified bit error ratio(BER) or packet error ratio (PER). Rsensitivity incorporates the various signal attenuation loss factorsdescribed earlier. The term should not be confused with the receiver noise figure or the signal-to-noiseratio (SNR) as these terms are also inclusive of the Rsensitivity calculation. The designing of the mmWaveRX antenna is closely related to Rsensitivity as the signal attenuation loss in the antenna feedlines degradesthe noise figure of the receiver. PTX is defined as the transmit power and can be understood as either thetotal power transmitted at the TX antenna port or the total power generated by the power amplifier. Ingeneral, the amplitudes of the two tend to be different as internal signal losses such as routing lossesbetween the power amplifier and the TX antenna or switch and duplexer losses tend to be non-negligibleat mmWave frequencies. Conclusively speaking, maximizing the radio link budget (dB) is the principalpriority in mmWave radio system design.

Fig. 4 One-way signal attenuation as a function of frequency


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Based on Eq. 5, increasing PTX is an intuitive method in achieving the desired radio link budget andmmWave radio signal coverage. However, it remains an extremely difficult technical challenge to achievean ideal balance between permissible levels of power consumption and power-added efficiency (PAE)using present-day semiconductor technologies. The scarcity of available power consumption capacity isevidently rampant particularly in the consumer electronics arena. For example, the recent design trendfeaturing continuously improved hardware capacity for slimmer smartphones results in significant strainon realizing improved battery life for mobile and wearable devices. Utilizing III–V compound semicon-ductor technologies in conjunction with CMOS processing for better RF output power and efficiency atmmWave can potentially alleviate this constraint. However, identifying cost-effective fabrication andpackaging schematics remains a pertinent issue at present.

The significant power constraints related to PTX exemplifies the importance of antenna gain atmmWave frequencies. For better understanding of the effect of the antenna gain, Eq. 3 can be reorganizedto be function of the antenna aperture as presented in Eq. 6:

PRX ¼ PTX1

4pR2

� �l2

4p

� �(6)

For resonant antennas, the apertures of mmWave antennas are within the sub-mm and mm range, whichare typically a fraction of low-microwave-frequency antenna apertures. This decrease in antenna apertureat mmWave naturally results in smaller captured energy at the receiver side which explains the inverserelationship between PRX and antenna operating frequency. Employing omnidirectional antennas similarto those that are typically used for Wi-Fi and cellular communications, the relatively small antenna gaincombined with signal attenuation loss at mmWave can potentially reduce received signal power levels inthe orders of 20logF where F is operating frequency in GHz. For example, for identical PTX, the PRX at30 GHz will be 20 dB less than that at 3 GHz. Consequently, these design parameters become significantlimiting factors in transmitting and receiving radio signals beyond very close range, line-of-sight (LoS)distance. This phenomenon justifies many of the mmWave applications designated for reactive andFresnel zones for which the antenna design aspects will be discussed later in this chapter. Evidently, themmWave antenna aperture becomes an ever-important piece in the puzzle. As antenna aperture is directlylinked to its directivity and gain, Eq. 6 can be rewritten in the form of Friis transmission formula as shownin Eq. 7:

PRX ¼ PTXGTXGRXl

4pR

� �2

(7)

GTX and GRX are denoted as the respective antenna gain for transmit and receive antennas. Furthermore,Eq. 7 can be reorganized to illustrate how transmit and receive antenna apertures affect PRX as shown inEq. 8 where Ae, Tx and Ae, Rx are defined as the effective antenna aperture for transmit and receive antennas,respectively:

PRX ¼ PTX4pAe,Tx

l2

� �4pAe,Rx

l2

� �1

4pR2

� �l2

4p

� �

¼ PTXAe,TxAe,Rx1

l2R2

� � (8)


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From Eq. 8, PRX can be interpreted as the product of the effective apertures of the two antennas and isinversely proportional to the square of the operating frequency and communication distance. Figure 5illustrates a simple demonstration of the effectiveness of increasing the antenna aperture at mmWave. Asan example, a rudimentary patch antenna at 3 GHz is designed. When applying an identical patch antennatopology at 30 GHz, the antenna aperture would decrease in accordance to the square of the antennascaling factor, eventually resulting in a 20 dB difference in PRX. However, by designing an antennatopology at 30 GHz which features an antenna aperture size that is identical to that at 3 GHz, theconsequent antenna gain enhancement compensates for the inherent path loss difference associated withinRsensitivity as described earlier. This demonstration underlines the critical importance of exploitingnumerous antenna gain improvement design methodologies for mmWave communications. It alsoexplains why majority of mmWave antenna designs employ various aperture and phased-array antennadesign schematics. At mmWave frequencies, losses within the antenna substrate material interconnectinsertion losses and polarization mismatch can be mostly attributed to the reduction in antenna gain.

Antenna Radiation CharacteristicsThe antenna gain is inherently dependent on the directivity of the antenna. The antenna directivity isdefined to be the ratio of the radiation intensity in a given direction (y,’) in the far-field region to theradiation intensity value which is averaged over a surface area of sphere. Resonant-type or electricallysmall antennas tend to radiate equally in all directions and effectively have very low directivity which isclose to 1 in amplitude or 0 dBi. For applications where the antennas mostly operate within the near-fieldor very close range, the directivity of mmWave antennas is less of a design priority. In contrast forscenarios where the mmWave transmitter and receiver are separated by more than a few meters indistance, the directivity of the antenna plays a substantial role in establishing a radio link. It is possibleto categorize how the antenna radiation characteristics affect the overall quality of mmWave radios basedon the deployments of the transmitter and receiver.

Fig. 5 Demonstrating the effectiveness of increasing the antenna aperture


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Fixed-Beam Applications (Point to Point)The RX and TX antennas are respectively configured based on careful radio link estimations using SNR,effective isotropic radiated power (EIRP), and received signal strength indication (RSSI). The deploy-ments of the mmWave transmitter and receiver are carefully devised often using iterative ray tracing andchannel propagation modeling and eventually set at stationary positions. Such deployment schematics areoften used for close- and midrange network architectures such as mmWave backhaul/fronthaul as well asvery close-range applications such as chip-to-chip (C2C) or board-to-board communication scenarios. Anexemplary user scenario is illustrated in Fig. 6.

The beamwidth and directivity of the stationary mmWave antennas must be carefully designed tosuppress unexpected misalignment between the transmitter and receiver. The sensitivity of the antennamisalignment for LOS communications becomes a greater issue as the communication range increasesand the antennas become more directive. This becomes an important design consideration in particulardue to the fact that majority of point-to-point mmWave communications are designed for outdoorenvironments where anything as subtle as wind, snow, birds, and any other expected environmentaldistractions can offset the antenna alignments by a few millimeters. The beamwidths of the mmWaveantennas must be designed to withstand such degradations to preserve the quality of service (QoS).Some of the recent mmWave antenna designs for point-to-point communication circumnavigate this issueby employing precise mechanical beam steering. Electrical beam steering methods such as phased arrayare also being researched in recent years. However, interconnect losses between the antenna elements andthe radio-frequency front end become increasingly problematic as the number of antenna elementsincreases.

Antenna Beamforming Applications Using Beam SteeringFor situations where LOS environment propagation environments or point-to-point communicationconditions cannot be guaranteed, it is essential to able to redirect the mmWave signals in order to searchand reestablish a connection link within a very short time. Widely referred as to antenna beamformingtechnology, mmWave signals with narrow beamwidths generated by directive antennas are steered acrossthe elevation and azimuth angle to transmit and receive radio signals using the most efficient propagationpath. Antenna beamforming is highly effective for indoor environments where there tends to be limitedwall penetrations, high absorptions, and various multipaths due to signal reflecting off furniture, ceiling,humans, etc., at mmWave spectrum. Compared to frequencies commonly used for present-day cellularand Wi-Fi applications (below 6 GHz), radio waves and mmWave frequencies rely on reflection whenestablishing links for NLOS propagation conditions as illustrated in Fig. 7. Antenna beamforming is also a

Fig. 6 Example of a mmWave fixed-beam application


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critical and useful methodology for outdoor and indoor-to-outdoor applications. Obstacles such aspedestrians, vehicles, lightning posts, and foliage can instigate double knife-edge diffractions (Vogler1982) causing “point of failures” in fronthaul, backhaul, and access networks. By spatially reusing theatmosphere, mmWave outdoor communication systems can not only circumvent signal blockage but alsosignificantly enhance the network capacity and data rates.

Encouraged by the advancements of silicon technology and antenna packaging techniques, electricalbeam steering has been increasingly favored over conventional mechanical beam steering at mmWavespectrum due to advantage in form factor, economic cost, and accuracy. Electrical beam steering can bebroadly categorized into two techniques as follows:

Switched Beam Steering Switched beam steering techniques combine the radio wave signals in thearray using a fixed set of weight vectors and time or phase delays. This is widely used for access points andcellular base stations where the direction of arrival (DoA) is well understood. Each antenna element mustbe designed to the radiation within a predefined “sector of coverage” and oftentimes utilize active RFcomponents, such as switches, MEMS, combiners, etc., for real-time beam steering. Due to the smallantenna aperture at mmWave frequencies, the overall antenna dimension employing switched beamsteering can be advantageous for maximized far-field radiation coverage as illustrated in an example inFig. 8. mmWave sub-array schematics can be employed to enhance the directivity targeted for each sector.Similar to point-to-point communication applications, the directivity of the mmWave antenna must becarefully designed to avoid unexpected nulls in the combined far-field radiation patterns. This isespecially the case for military and space applications where design tolerances are usually very tight.However, for commercial mmWave applications, such as mmWave access points and sensors, the antennapattern requirements are typically lower. Low cost, design complexity, ease of manufacturing, andform factor must be carefully considered. An antenna array with simplified corporate feeding networks(Pozar 1992), sub-array overlapping, or interleaved array arrangements (Abbaspour-Tamijani

Fig. 7 Indoor mmWave beam steering scenario


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and Sarabandi 2003) can be very instrumental in reducing the number of active RF components.For interleaved mmWave array, the mutual loading between adjacent antenna elements is dependent onthe state of the antenna beam and methods to mitigate the mutual coupling effects must be taken intoconsideration. Furthermore, fixed beam steering techniques can be applied in conjunction with adaptivebeam steering for mmWave applications requiring beyond long-distance coverage and wide beamsteering range.

Adaptive Beam Steering Initially developed in the 1960s for military applications such as sonar andradar, adaptive beam steering was initially devised to mitigate the effect of signal jamming. The concepthas continued to evolve over the decades and is now commonly defined as a technique to automaticallyadopt in real time based on predefined criterion to constructively combine the signals in a directionexhibiting the least amount of propagation attenuation.

A line array consisting of N number of identical point sources that are uniformly spaced as a function ofd is represented in Fig. 9. Assuming that the array is receiving an incoming signal Sn(t) at the strongestdirection y, the slanted angle y introduces a delay in the time of arrival of the signal between the referencesignal point source P0(t) and Pm(t) denoted as t where tn =nt. t is defined as

t ¼ d sin yc

(9)

The received signals can be superimposed in a constructive manner in the direction of y by subsequentlyadding true-time delays so that the signals arrive in a coherent manner. Then Sn(t) can be expressed asfollows in Eq. 10:

Sn tð Þ ¼ S t � tn � tshift, n� �

cos oRF t � tshift, n� ��

(10)

Majority of mmWave communication applications fit the model of narrowband conditions in which thefrequency bandwidth is a fraction of the center frequency oRF. In such cases, the true-time delay can beapproximated with a constant phase shift instead. Denoted as phased array (Natarajan 2007), Eq. 10 can

Fig. 8 Sectorized switched beam steering


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then be reorganized in the frequency domain and can replace the true-time delay with phase shifts ’n ateach point source. Then Sn(t) can be expressed as follows in Eq. 11:

Sn tð Þ ¼ S t � tnð Þ cos oRF t � tnð Þ � fnð Þ (11)

where j isfn ¼ N � nð ÞfThe total received signal of the phased array in the direction of y can be expressed as the summation of

Sn(t) as follows in Eq. 12:

Sarray tð Þ ¼XN�1

n¼0S t � tnð Þ cos oRF t � tnð Þ � fnð Þ

¼XN�1

n¼0S t � ntð Þ cos oRFt � noRFt� N � nð Þfð Þ

(12)

Assuming narrowband conditions,

S tð Þ � S t � ntð Þ � S t � tð Þ ¼ S0 tð Þ

It is possible to rewrite Sarray (t) as the following:

Sarray tð Þ�� ¼ sinN oRFt� fð Þ

2

sinoRFt� f

2

S0 tð Þ

��

��(13)

Equation 13 infers that for planar antenna arrays where d > l/2, the far-field radiation pattern of thephased array will be subject to grating lobes. Phased-array antennas where d << l/2 can potentiallyresult in excessive mutual coupling and introduce nonlinear terms in Eq. 13. Hence, it is generally the ruleof thumb to approximately configure d = l/2. The 3 dB beamwidth is dependent on N and y and beapproximated as in Eq. 14. Based on the rules of antenna reciprocity, the beam steering behavior remainsidentical during transmission mode:

Fig. 9 An illustration of a uniformly spaced lined array


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BW 3dB ¼ 0:87lNd cos yð Þ (14)

The derivation of the phased-array antenna pattern demonstrates that each phase shifter located atrespective point sources must be variable from 0 to 2p to scan the main beam of the antenna array acrossthe hemisphere (i.e., �p/2 � y � p/2). For practical use cases, the scanning angle of the main beam isreduced due to the rise in side lobes and can be detrimental in distinguishing the most coherentpropagation path for beamforming algorithms using the maximum power criterion. The effect of antennabeam steering range and the beamwidth on mmWave beamforming protocols can be understood in aqualitative manner.

Figure 10 exemplifies a set of beamforming protocols for the 60 GHz-based IEEE 802.11ad standardwhich is devised for uncompressed high-definition (HD) video streaming and high-speed data transfer.Understanding the specifics of the protocol is beyond the scope of this handbook, and the readers arereferred to Jo et al. (2014) for details. The basic principle of concept can be described as below:

• To define a set of antenna steering directions y for a spherical sector coverage• To cover the given spherical sector with minimum antenna gain• To minimize the number of antenna positions used

• Then, the generated set of the antenna positions is applied at both TX and RX sides, and an iterationover all mutual TX and RX antenna positions from this set is used to define a combination giving themaximum transmission power.

The radiation patterns during the quasi-omnidirectional antenna configuration and directional TX/RXantenna configuration are highly correlated to the efficiency of the adaptive beam steering algorithmillustrated in Fig. 10. Oftentimes, it is necessary to introduce variations in the antenna beamwidth betweenthe sector level sweep (SLS) and the beam refinement protocol (BRP) frame. A main beam of an N � Mphased-array antenna can be characterized by its respective beamwidth BWev and BWaz in the elevationand azimuth plan as shown in Fig. 11. During the adaptive beamforming optimization process, the SLSframe initiates link detection by selectively training the TX and RX antenna. Antenna array configurationswith predefined BWev and BWaz are used in search of the best sector. The BRP frame uses an iterativeprocedure to “train” the RX and TX antenna in identifying the optimal propagation path in that certainsector and oftentimes employ identical BWev and BWaz used during the SLS . In situations where thereceived power levels fall beyond the dynamic range of the wireless link budget, the protocol reverts backto the SLS frame using a refined main beam exhibiting different BWev and BWaz. The dynamic scalabilityof BWev and BWaz is correlated to the efficiencies of the SLS and BRP frames.

The steering angle y must be designed to cover the maximum physically allowable rangewhile minimizing the beam steering iterations for a timely transmit and receive training field(TRN-T/TRN-R). The resolution of the antenna array beam steering is a function of antenna spacing dand the angular resolution (or the number of maximum allowable bits) of the corresponding phase shifters.Pencil-beam and fan-beam antenna patterns are commonly preferred for mmWave-based adaptive beamsteering for this matter. The final decision will be made based on the specific application.

Key mmWave Antenna Design Techniques

This section discusses a number of antenna design parameters and ensuing techniques that are particularlycritical at mmWave frequencies. For wireless communication applications, low-profile antenna topologies


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are preferred due to its ease of fabrication, assembly, and integration with the rest of the wireless system.Oftentimes, the reduction of the antenna topology is realized at the cost of important antenna character-istics such as directivity, gain, and bandwidth. Therefore it becomes imperative to clarify the advantagesand disadvantages of each antenna designs when faced with specific allowable antenna real estate.Fabrication and assembly cost is another important design consideration in consumer electronics andautomobile industry and this is closely correlated to the selection and lamination of the antenna materialand type of the antenna. To assist in designing the appropriate type of antenna for a specific application,several mmWave antenna types that are frequently used are illustrated and summarized in Table 1.

The aforementioned antenna types can be used independently or in a combination in the form ofmultipurpose antenna arrays. The design procedure can be collectively characterized as listed below:

1. Identify the target frequency of operation and required antenna bandwidth.2. Determine the antenna gain utilizing the wireless link budget for a target communication distance and

coverage.3. Estimate the optimal directivity, radiation pattern, and polarization based on the environment

encompassing the antenna.4. Assess the maximum allowable antenna real estate and determine the best placement within the

wireless device.5. Assess the antenna feeding mechanism and calculate the feedline loss.6. Select the type of antenna to be used for the specific application.7. Choose the antenna substrate material, methods of packaging, fabrication, and integration with the

mmWave RFIC.8. Devise methods of antenna performance evaluation and identify design parameters for occasional fine-

tuning.

Table 1 Summarized comparison of mmWave antenna types

Antenna type Ease of fabrication Real estate Gain Bandwidth

Planar and printed antenna Feasible Small Moderate Narrow

Dielectric resonator antenna Moderate Moderate Moderate Low

Substrate-integrated waveguide antenna Moderate Moderate Moderate Large

Reflector antenna Difficult Large High Moderate

Slot antenna Feasible Moderate Low Moderate

Lens antenna Difficult Large High Narrow

Fig. 11 Definition of the antenna beamwidth


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In-depth discussions on the design procedure will be presented in the following subsections.

mmWave Antenna Packaging and Integration

Antenna PackagingMethods of integrating the antenna with the rest of the wireless module are one of the most distinguishabledifferences between mmWave RF front ends and those designed at microwave frequencies (typicallybelow 10 GHz). Antennas for preexisting wireless communications such as 3G/4G or legacy Wi-Fi areregarded as an independent design block within the entire wireless architecture. The antenna, RFIC,digital IC, and the subsequent interconnects are separately designed based on predefined and oftentimesstandardized hardware interfaces. The antenna and RFIC are typically connected via industry standardinterconnects such as coaxial RF connectors, surface-mount (SMT) contact clips, printed circuit board(PCB) lines, and RF switches that are frequently used in conjunction with external microwave matchingnetworks. The major contributors to the insertion loss between the antenna and RFIC are input reflectioncoefficient properties and the dielectric loss of interconnect and the matching networks. However, thiscomposition becomes problematic when the target operating frequency is located at mmWave spectrum asthe insertion loss is increased dramatically compared to the microwave spectrum. Emphasis on containingthe insertion loss of the interconnects between the antenna port and the RFIC pad becomes one of theprimary concerns for mmWave antenna design. Hence, the basic rule of thumb for mmWave antennas is tominimize the physical distance between the antenna port and the RFIC pad.

In general, there are two major philosophies for mmWave antenna packaging – the system on chip(SoC) and the system in package (SiP). SoC antenna packaging is realized in the form of on-chip antennaswhich are directly incorporated within multilayer metals using silicon technologies such as CMOS orIII–V semiconductors. An on-chip antenna configuration using 130 mmCMOS technology is exemplifiedin Fig. 12. CMOS implementations are advantageous over III–V technologies for its scalability, higherlevels of integration with the rest of the microwave integrated circuits (MMIC), and lower cost. On theother hand, III–V technologies are superior in terms of higher electron mobility, power handling capacity,and lower substrate loss. The low resistivity of the substrate used for on-chip antennas is the primaryattribution to its poor antenna gain and high insertion loss across the antenna feeding networks. Theresistivity is in the orders of 10 Ω-cm for high-volume CMOS technologies and approximately in theorders of 107–109 Ω-cm for III–V technologies. mmWave antennas designed using low-resistivity

550 μm 550 μm

SiO2

Si Feed

Two parallel groundplanes each 0.6 μm thickseparated by 0.6 μm

Fig. 12 Design example of an on-chip slot antenna (Behdad et al. 2007)


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substrates are subject to severe signal attenuation and this is exacerbated by the thin metallic layersincorporated using standard foundry processes. The thickness of the metallic layers is fractions of the skindepth at mmWave frequencies and the antenna topology is further penalized with substantial substratelosses. On-chip antennas employing high-resistivity silicon substrates are effective methods in reducingthe signal attenuation constant. However, the high fabrication cost remains an obstacle for mass marketapplications such as consumer electronics. In recent years, periodic electromagnetic bandgaps in the formof metamaterial surface or artificial magnetic conductor (AMC) surfaces have been postulated for on-chipantennas as presented in Fig. 13 (Lin and Ooi 2009; Chu et al. 2009).

The main motivation behind SoC antenna packaging is to realize the entire RF front-end architecture ina completely monolithic fashion. As previously explained, the reduced physical distance between themmWave antenna element and the rest of the MMIC does not always warrant improved antenna feedlinelosses. For applications where multiple number of antenna elements are required, SoC antenna may not bejustifiable in the sense of engineering and economics. For example, in 60 GHz phased-array applications,the increased area incurred by the inclusion of multiple on-chip antenna elements within a silicon RFICdie will dramatically drive the unit price of the RFIC die upward. Attributed to these trade-off relations,SoC antenna packaging schematics are better suited for mmWave spectrum beyond 90 GHz or for veryclose-range mmWave wireless communication scenarios.

Another efficient method is to directly integrate the antenna with the rest of the wireless module.Defined as system in package (SiP), the packaging methodology is illustrated in Fig. 14 using multilayerlamination topologies.

The lateral integration schematic presented in Fig. 14a is suitable for applications where an end-fireradiation mode is advantageous. The mmWave antenna element is designed and placed in very closedistance to the RFIC using microstrip, stripline, or coplanar waveguide (CPW)-based antenna feedingnetworks. This topology is especially compatible with dielectric resonators antennas (DRA), Yagi-Udaantennas, and horizontal dipole antenna types. Oftentimes, an additional transition such as a balun or amatching subnetwork is required to fine-tune the characteristic input impedance of the antenna port. High-permittivity substrates such as ceramic, high-end organic materials, and polymers are selected for thechosen antenna designs such as DRA and subsequently integrated with the surface of the mmWave carrierboard using surface-mount technology. For planar antenna types such as Yagi-Uda antennas and hori-zontal dipole antennas, utilizing the top surface of the PCB carrier board is more advantageous in terms ofdesign simplicity, cost, and overall aperture of the mmWave antenna module. Figure 15 shows a uniqueantenna packaging method – for mmWave frequencies where the profile of the resonant antenna is small

Fig. 13 SoC antenna implemented on an artificial magnetic surface (AMC)


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enough to be fully embedded within the antenna carrier board. Using multilayer lamination technology,the antenna element is constructed using vertical interconnects such as stacked vias, through vias, andburied vias. The antenna element topology is equivalent to a vertically oriented monopole antenna andwill radiate in the end-fire direction.

The effectiveness of the lateral integration schematic tends to degrade as the number of antennaelements increases. This is primarily attributed to the increased complexity and physical length of theantenna feeding networks as the antenna elements must be mutually separated at a predefined minimumpitch. Consequently, the lateral integration schematics are commonly used for mmWave wireless com-munication scenarios where the total number of RF paths and the antenna element is relatively small.

Figure 14b illustrates a vertical integration schematic. This type of antenna packaging configurationoffers a wider range of antenna type selections compared to the lateral integration schematic. The antennaelements and the RFIC are typically located on opposite faces of the antenna module substrate for twodistinctive reasons: (1) minimized interconnection insertion loss due to the employment of vias and(2) minimized antenna array footprint and antenna module dimension. Assembling the RFIC with theantenna module in an inverted manner is denoted as flip-chip bonding, also known as controlled collapsechip connection, or C4. The solder bumps are deposited on the top side of the RFIC during the final waferprocessing process constructing RF ball pads. The RFIC is eventually inverted or “flipped” over so thatthe solder bumps come in direct contact with the signals pads of the antenna module. Subsequently solderis reflowed to construct the interconnect. At mmWave spectrum, flip-chip bonding tends to be preferredover conventional wire bonding attributed to superior rigidity and parasitic effects. The parasitic induc-tance of bond wires at mmWave frequencies is approximately 1 nH per 1 mm bond wire length. Thisamount of parasitic is sufficient to cause detrimental detuning of the antenna impedance matching.The limited controllability on the exact wire bonding procedure is another challenge when the RFICdie requires large number of bonding connections with the carrier board.

Fig. 15 The end-fire monopole antenna realized using stacked vias

Fig. 14 SiP mmWave antenna integration schematic. (a) Lateral integration. (b) Vertical integration


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Planar antennas with broadside pattern properties such as patch antennas and slot antennas areparticularly advantageous with the vertical integration schematic. The die size of mmWave RFICs iswithin the range of a few mm2, while the mmWave antenna array topology tends to be much larger.Therefore, in addition to the vertical interconnects, lateral antenna feedlines such as microstrip, striplines,or CPW transmission lines are used to route the RF signals and align the ball pads of the RFIC to thecorresponding antenna element ports. Likewise, the insertion loss attributed by the antenna feedingnetwork is proportional to the number of RF paths and antenna elements.

Antenna MaterialThe substrate material for the mmWave antenna modules is a critical design parameter in terms of cost-effectiveness, design flexibility, and antenna performance. Advanced antenna materials such as low/high-temperature co-fired ceramics (LTCC/HTCC) and organic printed circuit boards represented by liquidcrystal polymers (LCP) feature scalable permittivity and low loss tangent values at high RF frequencies.For such reasons, they are preferred over high-volume PCBs such as FR-4 for high-performancemmWave antenna designs. In addition, the fabrication and lamination processes using advanced antennamaterials oftentimes exhibit more precise and sophisticated design rules such as smaller producible linewidths and finer via pad diameters compared to FR-4. These advantages collectively grant mmWaveantenna engineers enhanced degrees of freedom during the design stage. Ease of integration with variouslumped components or ICs (die) is another important criterion in selecting among available antennamaterials. The designed antenna aperture must feature minimum warpage to avoid damaging surface-mount lump components. Warpage can be quantitatively characterized using parameters such as theYoung’s modulus (Wang et al. 2003). Thermal stability and rigidity is another important design consid-eration. The excessive heat created by the MMIC circuit is typically harnessed or funneled using thermalvias or external heat sinks. In cases where the mmWave antenna modules are extremely small indimension, the antenna material is exposed to higher heat temperatures and can result in unexpectedmechanical deformation. Such thermal expansions of the antenna material can lead to damages or cracksin the solder balls between the antenna pads and the IC. Antenna materials with coefficient of thermalexpansion (CTE) values that are closer to that of the IC lessen the probability of such incidents fromoccurring.

In terms of scales of economy, the material and fabrication cost of advanced antenna materials canfrequently be several times higher than that of FR-4. The lamination can be simplified by reducing thenumber of layers or the total antenna module dimension for further cost savings.

mmWave Wireless Module IntegrationmmWave wireless modules can be in large classified into three major components – the antenna module,RFIC, and the digital IC. The three components are ultimately integrated together on a carrier board whichcontains external peripheral components such as peripheral component interconnect express (PCIe) oruniversal serial bus (USB) interfaces. Extensive interconnections transporting numerous signals andreference in the form of RF, IF (intermediate frequency), reference clocks, IC power supply voltage,and reference signal ground coexist within wireless communication modules. The complexity of theinterconnections and RF architecture must be taken into account in devising an appropriate method ofintegration. Three distinctive methods of mmWave wireless module integration are illustrated in Fig. 16.

The key differentiating features between the wireless module integration technique presented inFig. 16a compared with Fig. 16b and c are the composition of the wireless module and the ensuingassembly procedure. This type of wireless module topology is preferred when the different materialsubstrates are used for the antenna module and the carrier board, respectively. The antenna module entailsa vertical integration schematic oftentimes using advanced multilayer laminations such as LTCC, HTCC,


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or organic PCBs. The three major components identified as the antenna module, RFIC, and the carrierboards are subsequently assembled to realize an integrated wireless module using the following three-stepassembly procedure:

1. Flip-chip attach the RFIC to the antenna module.2. Surface mount the antenna module containing the RFIC with the carrier board using grid-array

assembly technologies such as ball grid array (BGA) or land grid array (LGA).3. Surface mount external peripherals such as connectors and lumped components on to the carrier board.

The finalized wireless module is connected with the digital IC either via a microwave transmission lineor coaxial-type transmission line depending on the physical separation distance between the two. Withinthe wireless module, the IF signals, reference clock, power supply voltage, and the signal groundreference are interconnected from the digital IC via the antenna module and the carrier board. The flip-chip solder balls must be meticulously designed to minimize RF reflections and impedance mismatches.The electrical properties of the vertical interconnect represented by the flip-chip solder ball can bemodeled using numerical electromagnetic computations based on computer-aided design (CAD) render-ings. Figure 17 illustrates a flip-chip solder ball transition between the antenna module and the RFIC pad.The reactance of the transition tends to be inductive (several nH) and this can be compensated usingpassive matching networks such as shunt stubs which are attached to the antenna feedlines. The solder balllayout constructed on the bottom of the antenna module must be identically mirrored on the top surface ofthe RFIC die. For mmWave antenna modules, optimizing the solder ball layout is one of the most criticaldesign stages out of the entire process. The following key design parameters are affected by the solder balllayout:

– The RFIC die size and front-end architecture– Antenna material, manufacturer, and fabrication method– Characteristic impedance of the vertical interconnect at the target operating frequency

Fig. 16 mmWave wireless module integration techniques


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Accurate characterization of the solder ball transition is extremely critical as the cascaded LNA, PA,and switches which will most likely not feature purely resistive 50 Ω input impedance. It is imperative toimplement reactive matching networks on the solder ball layout. The topology of the solder ball layoutmust be devised so that each signal pad remains balanced and operate in the quasi-TEMmode. Each signalpad must be encompassed by a minimum of four reference ground pads as illustrated in Fig. 18a. Closelyresembling a coaxial topology, the pitch P between the signal pad and the reference ground paddetermines the characteristic impedance of the vertical interconnect as well as the permittivity of thesolder balls, encapsulation, underfill, and antenna material. The minimum allowable pitch P is frequentlylimited by the fabrication accuracy and resolution of the PCB lamination technology involved. Insituations where permissible real estate of the RFIC or the antenna module cannot afford four referenceground pads, the solder ball layout can be further simplified to a CPW configuration as illustrated inFig. 18b. This rule of thumb can be further expanded for solder ball layouts containing multiple signalpads as shown in Fig. 18c. Failure to properly shield the signal pads with sufficient number of referenceground pads will pick up stray electric fields and induce unwanted noise in the antenna feeding network.

As discussed previously, the mechanical rigidity of the flip-chip solder balls are affected by the specificsubstrate material of the antenna module and heat generated by the RFIC. The flip-chip solder balls can besignificantly enhanced mechanically, owing to recent advances in flip-chip underfill materials andprocesses.

mmWave antenna modules using the three-step configuration generally require several hundreds ofgrid-array assemblies to construct an electrical, mechanical, and thermal passage between the RFIC andthe carrier board via the antenna module. Needless to say, any minor damaging of the grid arrays canpotentially be related to a direct failure of the entire mmWave module. The importance and the sensitivityof the grid array result frequently require extensive levels of quality assurance inspections involving time-consuming and costly testing such as X-ray scanning. Compounded with the usage of advance antennamaterials, the fabrication cost incurred by the ball grid arrays may exceed target price ranges for selectedhigh-volume applications.

Fig. 18 Topologies of the solder ball layout. (a) Quasi-TEM mode. (b) CPW mode. (c) Multiple signal bump layout

Fig. 17 Layer view of the flip-chip solder ball transition


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Figure 16b presents a wireless module integration technique which can be instrumental in resolvingcost-related issues discussed in the three-step configuration. As opposed to a separate antenna module andcarrier board configuration, the stated topology fully embeds and integrates the antenna module inside thecarrier board. This approach eliminates the need for a discrete antenna module and the entire laminationcan be realized monolithically. The overall assembly process of the wireless module is simplifiedcompared to the three-step configuration as described in the following:

1. Flip-chip attach the RFIC to the monolithic antenna module/carrier board.2. Surface mount external peripherals such as connectors and lumped components.

It should be noted that the specific sequence between the two assembly procedures can be interchangeddepending the feasibility of applying the stencils for the flip-chip assembly process with or withoutsurface-mount components. As it can be observed, the two-step wireless module configuration entirelyeliminates the need for any grid-array assemblies. Consequently, the overall time and cost of the wirelessmodule integration is noticeably reduced, translating to a more competitive unit cost for high-volumeapplications. Bypassing the grid-array assembly enables integration flexibility with the rest of the wirelesscommunication hardware. This integration technique enables a multilayer mmWave antenna design to berealized within a wireless device hardware without any additional discrete material cost and assemblyprocess. The whole motherboard and the wireless module can be integrated as one, simultaneouslydesigned and fabricated monolithically. Figure 16b is extremely advantageous for scenarios where thecharacteristics of the selected mmWave antenna design, the antenna material, its specific user scenario,and the antenna placement are well established. One notable limitation comes with the need to implementthe digital IC in close proximity to the antenna for super heterodyne wireless architectures. In addition, insituations where a faulty mmWave RFIC die or an error within the antenna module is identified, the entirecarrier board must be replaced as opposed to just replacing the antenna module in Fig. 16a. Figure 16cmitigates this issue with a stand-alone antenna module.

The two-step wireless module configuration is valid under the precondition that the antenna module,carrier board, and occasionally the motherboard use identical or very similar substrate materials, lamina-tions, and design rules. This can be rephrased to state that the antenna material is dependent on thematerial composition of the carrier board and the motherboard. Few exceptions withstanding, conven-tional high-volume PCB materials such as FR-4 are likely to be used in consumer electronics, telecom-munications, and automobile industries. As the material loss tangent for conventional PCBs can be in theorders of 0.005 and higher, it is imperative to devise methods in reducing the insertion loss incurred by theantenna feeding networks within the antenna module/carrier board. Utilizing the CPW topologies asopposed to microstrip topologies and efficient antenna feedline routing and layout are effective methods.

Hybrid mmWave Antenna Packaging TechnologiesIt becomes possible to apply hybrid packaging and integration approach for mmWave frequencies that arehigh enough for resonant-type antenna elements to be deduced to extremely small form factors.

Figure 19 illustrates a mmWave wire bonding antenna topology. The inductive reactance and thecapacitive reactance respectively derived from the wire bond and the signal pad on the PCB carrier boardare utilized at a target resonant frequency. As the antenna geometry is quite fragile, additional encapsu-lation such as a polymer-based casing is required to protect the mmWave wire bond antenna fromunexpected deformation. The mmWave wire bond antenna is instrumental especially when dealingwith a limited number of end-fire radiation mode antenna elements typically at frequencies beyond80GHz. The design flexibility is relatively limited as the wire bonding process is subject to few variations.


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Another notable hybrid packaging and integration technique is described as the embedded wafer-levelball grid array (eWLB). This approach is derived by introducing fan-out regions inside a conventionalwafer-level package (WLP) as illustrated in Fig. 20. This is especially beneficial for multi-chip packagingapplications such as high-power communications in the automobile and telecommunication arena. Theconcept of the eWLB antenna is to devise a planar antenna topology inside the fan-out region of thepackage by utilizing the redistribution layer (RDL) (Al Henawy and Schneider 2011). This type ofpackaging technique is best fit for broadside antenna modes. The degree of freedom becomes somewhatlimited when designing antenna array geometries as the topology must be devised with the antennaelements encompassing the IC situated at the center. Nonetheless, the eWLB antennas are one of the mosteffective methodologies for close- and midrange mmWave communications beyond V-band frequencies.

Wideband mmWave Antenna DesignThe acute disparity between the available spectrum below 10 GHz and the required frequency bandwidthfor multi-Gbps wireless communication underlines the need of facilitating mmWave frequencies.In accordance to the classical Shannon-Hartley information theory, one of the most notable methods of

Fig. 19 Diagram of the mmWave wire bond antenna and its fabricated prototype (Willmot et al. 2009)

Fig. 20 Illustration of the mmWave eWLB antenna


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significantly enhancing the wireless capacity at mmWave frequencies lies in the expansion of thefrequency bandwidth. Bandwidth enhancement techniques that are relatively well understood andfrequently applied at microwave frequencies can also be incorporated in mmWave frequencies. Addi-tional design considerations which are unique to mmWave applications such as the aforementionedpackaging and integration design parameters must be further taken into account. The laminations andmultilayer package topologies can be capitalized in devising a robust, efficient, and broadband antennadesign topology. Single-layer mmWave antennas such as dipole antennas, Yagi-Uda antennas, andtraveling wave antenna types can also be designed to be compatible with the rest of the antenna modulepackage at the cost of increased lateral dimension.

Generally speaking, there are two ways of achieving broadband multilayer antennas at mmWavefrequencies. The first approach involves the employment of low-permittivity antenna substrate materials.Antenna designs based on low-permittivity dielectrics feature lower Q which translates to a more usableantenna bandwidth. The second approach involves increasing the thickness of the dielectric (Katehi andAlexopoulos 1983). However, this tends to induce surface-wave excitations which are especially detri-mental for planar antenna geometries. For TE and TMmode surface waves, the cutoff frequencies can beexpressed as follows in Eq. 15 (Harrington 1961):

f c ¼nc

4hffiffiffiffiffiffiffiffiffiffiffiffier � 1

p� � (15)

where c is the speed of light and n is an integer number for the corresponding TE and TM surface modes.Parameter h is defined as the height of the dielectric substrate of the antenna. More surface modes can existas n increases. In addition, the increase in height of the dielectric substrate enables lower-order surfacemodes to couple easier. In majority of situations, the dimension of the mmWave antenna module can beregarded as a finite-size dielectric substrate. In such cases, the excitation of surface-wave modes causesundesired diffractions which are difficult to characterize. The edge diffractions eventually result inradiation nulls in the main lobe, rise of side lobes, and deteriorated gain and polarization. The effect ofsurface modes on the far-field behavior for mmWave antennas is analyzed in Liu et al. (2009). The trade-off related to increasing the thickness of the dielectric substrate for bandwidth enhancement can becircumvented to a certain extent by slowly varying the excitation across the designed mmWave antennaelement. As a rule of thumb, antenna substrate profiles with h < 0.01l are recommended for planarantenna type topologies.

Selection of the optimum permittivity of the antenna material substrate and its height is succeeded bythe designing of the mmWave antenna geometry. For rudimentary planar antennas such as patch antennas,achieving more than 10 % bandwidth may be technically challenging. Generating multiple numbers ofclosely spaced antenna mode resonances by carefully controlling the parasitic capacitance is an effectivetechnique for multilayer laminations. Figure 21 illustrates variations of mmWave planar antenna sche-matics based on parasitic capacitance coupling methods. Figure 21a is described as the vertically stackedantenna geometry and is among the most effective methods for broadband applications requiring morethan 15 % bandwidth. The ensuing design consists of multiple parallel RLC circuits, respectivelyrepresented by patch antennas. The combined RLC circuit is excited by series capacitance couplingwhich is predominantly dependent on the separation gap between the patch antennas. In reality, thisseparation gap is highly correlated to the specific antenna substrate material, the available substrate height,and the lamination and fabrication method. Therefore the fine-tuning range of the separation gap istypically discrete. Figure 21b illustrates a horizontally coupled antenna configuration. As the nameindicates, the electric field generated by the parasitic capacitive coupling is contained in the horizontalplane. As a result, this type of antenna configuration is especially suitable for linearly polarized antennas


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having the E-plane located in the direction that is transverse to the horizontal separation gap. This type ofbandwidth enhancement is useful for mmWave antennas requiring thin antenna substrates or limitedlayers of lamination. The drawback compared to the vertically stacked antenna geometry is the increaseddimension in the horizontal plane. Additional antenna miniaturization techniques may be required forantenna array applications. The vertically stacked antenna and the horizontally coupled antenna techniquecan be used in conjunction as described in Fig. 21c. This configuration further enhances the antennaaperture efficiency and can be very effective for planar antenna designs at mmWave spectrum that requiremore than 20 % bandwidth. The trade-off is the increased complexity in the design which can lead topossible degradation of the fabrication yield.When applying conventional, high-volume PCB technology,the fabrication tolerance is oftentimes in the range of tens of microns, and this can result in detuning theparasitic capacitive coupling by more than 30 %. One way to stabilize the fabrication process and avoidcostly fine-tuning the antenna design is to first design and fully characterize the vertically stacked antennaand the horizontally coupled antenna geometry respectively and separately. Consequently the twomethods can be used in conjunction once the dominant design parameters that affect the antennabandwidth are well analyzed.

The antenna feeding is another critical consideration for broadband mmWave antennas. It can belargely classified into uniplanar and multilayer antenna feeding techniques as categorized in Fig. 22.Figure 22a represents a microstrip edge feed method which is compatible for patch antennas and planarmonopole antennas. The microstrip feed is characterized as a series inductive feed and this method isfavored for its simplicity as both the antenna and feedline geometry can be fabricated in simultaneously.The drawback is the design inflexibility incurred by the limited number of available design parameters andthe increasing far-field radiation contribution of the microstrip feed as the footprint of the mmWaveantenna becomes smaller. Therefore it is important to limit the length of the microstrip feed at mmWavespectrum. Figure 22b complements the restrictions and the limitations of the microstrip feed by utilizing a

Fig. 21 Bandwidth enhancement techniques for mmWave multilayer antennas

Fig. 22 Illustrations of planar antenna feeding techniques. (a) Microstrip edge feed. (b) CPW feed. (c) Probe feed. (d)Capacitive feed


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CPW feed technique. Similar to the Fig. 22a, the antenna feedline is situated on the same plane whichcontains the antenna. Compared to the microstrip feeding method, the CPW method offers more designflexibility and reduced erratic far-field radiation by the antenna feedlines. The uniplanar antenna tech-niques illustrated collectively suffer from drawbacks related to increase in the lateral real estate. FormmWave phased-array applications, the complexity of the antenna feeding networks is alleviated bydevising sub-array topologies using corporate and serial feedings. In addition, the inductive nature of theantenna feedline typically limits the antenna bandwidth to less than 10 %. Figure 22c represents a probefeed method. For mmWave multilayer laminations, the probe is realized using signal through-hole vias.Quasi-TEM mode can be realized by encircling the signal via with multiple numbers of ground vias toclosely emulate a coaxial vertical transition. This method benefits from negligible antenna feed radiationand extremely small profile. The input impedance matching of the antenna element to the probe iscontrolled by adjusting the contact location of the probe in the lateral direction. However, the bandwidthremains relatively narrowband. A capacitive antenna feeding method is illustrated in Fig. 22d. Theantenna element is excited via electromagnetic coupling and feature enhanced design flexibility. Shieldedmicrostrip or slot lines can be designed for the antenna feed transmission line. Using this method, it ispossible to excite dual antenna resonance by adjusting the antenna feed geometry and is one useful way ofenhancing the bandwidth. Figure 23 is denoted as the aperture-coupled feed (Pozar 1985). As one of themost favored and widely used mmWave antenna feeding techniques, the feeding structure consists of atransmission line and an aperture. The antenna element located at the top of the multilayer lamination isexcited due to the electromagnetic coupling between the apertures. The dimension of the aperture can becarefully designed so that it contributes constructively in the main direction of the mmWave antennaelement. Naturally this method is preferred for broadside antenna modes and is best fit for patch antennatypes. The input impedance of the antenna can be matched by adjusting the location where the

Fig. 23 Diagram of the aperture-coupled feed


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transmission line and the aperture intersect on the horizontal plane. As multiple numbers of antennaresonances can be realized through the aperture and the antenna element, the aperture-coupled feed can bea very powerful technique in realizing broadband characteristics.

In order to better understand how the antenna design topologies and the feeding methods are used inpractical scenarios, one of the most notable state-of-the-art broadband antennas designed for mmWavecommunications (Liu et al 2011) will be investigated. Figure 24 illustrates a cross-sectional view of anintegrated mmWave antenna based on a five-layer PCB lamination. The antenna substrate material for thethird layer denoted as Sub3 is chosen to feature dielectric permittivity which is approximately 12 % lowerthan the rest of the antenna substrate materials. This enables the antenna feedline designed on the 3rd layer(M3) of featuring an adequate width-to-height ratio for improved impedance matching. To furtherimprove the antenna feedline efficiency, low-loss adhesives are applied on Sub3, whereas relativelylossy adhesives are applied elsewhere to reduce the overall cost and enhance manufacturability. As it canbe observed, portions of Sub4 and Sub5 are removed to form an open cavity. The form factor of the opencavity is designed so that the RFIC can be completely inserted inside the mmWave antenna module. Thisconfiguration eliminates the need for vertical signal transitions using via holes and also eliminatesrequirement of associated matching networks. Signal pads and reference ground pads are implementedon the bottom layer M5 for ball-grid-array assembly to the carrier board.

The presented mmWave antenna renders an aperture-coupled feed technique to achieve broadbandperformance. The mmWave patch antenna element is designed and implemented on the M1 layer.A sealed air cavity is created by removing portions of Sub2. This essentially lowers the effectivepermittivity of the patch antenna resulting in lower Q. The ground plane located in M2 functions as thereference ground for the patch antenna and also is an effective way of shielding erratic radiation and signalcross talk within the antenna module. The aperture located on the M2 layer is designed based on a set ofresonant, half-wavelength slots as shown in Fig. 25. By merging the two antenna resonance modes, morethan 15 % 2:1 VSWR bandwidth can be achieved. The slots are designed to have slightly differentresonant frequencies in comparison to the patch antenna. The exact position of the two slots aredetermined so that the surface waves generated by the slots and the patch antenna are destructivelyadded. As a result, the antenna gain can be further improved. The slots are fed using a set of transmissionlines situated in M3. The metallic pattern located in M4 functions as the reflector for the slots since a slotantenna is inherently bidirectional.

GND plane

antipad

Sub1

Sub2

M1

M2Sub3

M3

Sub4M4

Sub5M5

signal pad via

chip

reflector GND padGND via

antenna feed line

patch slotGND via

Fig. 24 Cross-sectional view of the integrated mmWave antenna using aperture-coupled patch antenna (Liu et al 2011)


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mmWave Antennas Design Technologies for Directional and Reconfigurable RadiationPatternsThe applicability of classical omnidirectional antennas is greatly limited to very close-range mmWavecommunication scenarios due to numerous propagation loss factors discussed in detail in section“Theory.” In addition mmWave omnidirectional antennas are prone to receiving undesired incomingstray signals across a diverse range of angle of arrivals (AoA), which can significantly affect the noisefigure of the receiver circuitry. This phenomenon, denoted as desense, can degrade the sensitivity of thewireless architecture by up to 10 dB or more under real-life propagation environments and is one of theprimary concerns during the design phase. Spatial diversity techniques in the form of antenna beamsteering are instrumental in circumventing the influence of interferers especially for multipath propagationenvironments. Given ideal antenna design conditions, a beam steering antenna such as in the form of aphased array featuring a complete spherical radiation pattern coverage and fine angular resolution wouldbe the most effective solution. Unfortunately in most realistic antenna design conditions, such topologiescannot be realized due to constraints associated with dimension, economical, and technical costs. Onepractical approach involves enabling mmWave antenna elements featuring sectorized radiation patterns.This technique is particularly useful in propagation environments where the angles of arrival anddeparture can be estimated based on statistical spatial channel models. Directional mmWave antennaelements can be largely classified into broadside and end-fire topologies and will be discussed in detail asfollows.

mmWave Broadside AntennaFor planar mmWave antennas exhibiting broadside antenna mode, the main radiation lobe is situatedalong the axis that is approximately perpendicular to the plane which contains the antenna geometry.Broadside antenna mode is instrumental when the transmitting and the receiving planar antenna topol-ogies are configured to be facing one another. In another scenario, which is confined to indoor propagationenvironments, the two broadside antennas can be implemented adjacent to one another and establish aNLOS wireless link through utilizing the ceiling as an effective reflector. When multiple broadsideantenna elements are employed to devise antenna arrays, the accuracy of the mmWave NLO antennabeam steering is affected by the relative surface roughness of the ceiling at the operating carrier frequency.

Conventional planar patch antennas are one of most widely used broadside antennas at mmWave. Theantenna gain is typically in the range of 3–5 dBi for each patch antenna element under the precondition

slot patch

reflector

1000 line

Fig. 25 Feeding structure used for the mmWave aperture-coupled patch antenna in (Liu et al 2011)


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that relatively low-loss antenna substrate materials are used. mmWave patch antennas can be furtherexpanded into a two-dimensional array in order to attain greater antenna directivity and gain in thebroadside direction. In accordance to the well-understood phased-array theory, the array factor isproportional to the logarithmic value of the total number of antenna elements. This relation infers thatit is possible to continue enhancing the antenna gain and directivity in the broadside direction through theaddition of antenna elements. However, in reality, the effectiveness of this method is eroded beyond acertain number of antenna elements. This is due to the growing disparity between the footprint of themmWave RFIC die and the broadside antenna array. As illustrated in Fig. 26, the physical lengths of theantenna feedlines which route the mmWave signals between the RFIC signal pads and the respectiveantenna element ports are proportional to the total number of antenna elements comprising the broadsideantenna array. The cumulative insertion loss incurred in the antenna feedline network continues to riseuntil a certain threshold where it equals the array factor as presented in Fig. 27. Beyond this threshold, thegain of the broadside antenna array starts to decrease as the insertion loss of the antenna array becomesgreater than the antenna array factor. Increasing the mmWave RFIC die introduces complicated problemsassociated with extremely expensive unit costs and unexpected warpage as is deemed unrealistic. Due tothis limitation, it becomes difficult to attain an antenna gain beyond 30 dBi using mmWave patch antennaarrays.

In situations where the antenna gain and the directivity need to be enhanced further, an additionaldielectric lens can be implemented with the original antenna geometry. This design consists of a mmWavebroadside patch antenna situated on the upper layers of the PCB lamination. Illustrated in Fig. 28, adielectric lens featuring an elliptical or quasi-elliptical shape having a radius R is integrated with acylindrical-shaped dielectric with an extension height denoted as h The curvature of the dielectricfunctions as a plano-convex lens, introducing true-time delays (TTD) to form a highly directional antennaradiation pattern. The variations of extension height h modify the elliptical shape of the lens, leading todifferent broadside directivity. Optimization of the integrated lens antenna is performed by modifyingextension height h to control the antenna directivity and antenna array beam steering range. The losstangent of the dielectric lens and the internal signal reflections in the dielectric lens body are the twoprimary factors that degrade the antenna gain. The broadside antenna element which is feeding theintegrated lens is typically placed at the integrated lens focus to mitigate internal reflections. A fullyfabricated mmWave integrated lens prototype and variations of dielectric lens are respectively presentedin Fig. 29.

For mmWave access points, backhauls, and compact mobile terminals, the volumetric convex inte-grated lens solution can be potentially problematic. Designs based on spatial true-time-delay unitsemploying bandpass filter technologies can be effective flat lens design techniques. Also denoted as

Fig. 26 Layout of a mmWave broadside antenna array and the ensuing antenna feedlines


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mmWave transmit array, the topology comprises of single or multilayer quasiperiodic resonant elementsarranged to locally compensate the phase delay associated with different path lengths. An illustrativeconcept depicting the transmit array is presented in Fig. 30. The resonant elements constituting thetransmit array layers rephrase the incoming spherical waves and retransmit the mmWave signals in theform of plane waves. The required phase adjustments are dependent on the distance the signal has traveledfrom the broadside antenna to each of the resonant elements. Ri denoted in Fig. 30 is defined as the vectorof ith element from the phase center of the broadside antenna. ri is denoted as the position vector to the ith

Fig. 27 Gain limitation of large-scaled mmWave broadside antenna arrays due to antenna feedline loss

Fig. 28 Integrated lens antenna concept

Fig. 29 (a) Fully integrated lens antenna (Artemenko et al. 2011). (b) Dielectric lens variation (Designed by Radio Gigabit Inc#)


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resonant element from the center of the transmit array lens and uo is the direction of the transmitted mainbeam of the antenna. k is denoted as the propagation constant. Defined in Black andWiltse (1987), Hristovand Herben (1995), and Ryan et al. (2010), the required phase compensation fi at each resonant elementcan be expressed as Eq. 16 below:

fi þ k Ri þ ri � uoð Þ ¼ 2pn where n ¼ 0, 1, 2, 3 . . . (16)

Transmit array lenses offer the benefits of reduced profile and wide beam steering scanning ranges.However, due to the RLC circuitry of the resonant elements, they are commonly narrowband and mayrequire extensive multilayer fabrication procedures. The gain enhancement effect is highly sensitive to thephysical alignment of the transmit array lens and the broadside antenna, so additional efforts focused oneliminating mechanical jitters and vibrations may be required.

mmWave End-Fire AntennaEnd-fire antenna modes are effective solutions for mmWave device-to-device (D2) communications suchas file transfers or ultrahigh definition (UHD) seamless video streaming scenarios. The axis of the mainlobe is parallel to the plane containing the planar mmWave antenna geometry. The mmWave Vivaldiantennas, (Yngvesson et al. 1989) which are classified as traveling wave antennas, are advantageous whenthe number of available layers within a mmWave antenna module is limited. This type of antennageometry is most compatible with the lateral integration schematic described in the previous subsectionand widely used when the mmWave RFIC in implemented on the same plane of the carrier board. Theantenna gain is proportional to its length which is usually a multiple of the wavelength. For resonant-typeend-fire antenna designs, the mmWave Yagi-Uda and the planar dipole antennas respectively illustrated inFig. 31a and b are by far the most preferred topologies for low-profile applications (Alhalabi and Rebeiz2009). The mmWave Yagi-Uda antenna largely consists of three discrete sub-elements. The drivenelement consists of a planar dipole antenna topology which employs a half-wavelength balun on one ofthe two branches. The balun enables a 180o phase shift and drives the dipole into the desired antennamode. The driver element electromagnetically couples to the driven element and enhances the directivityin the boresight angle of the Yagi-Uda. The reflector suppresses the back radiation of the driven elementwhich is induced by the inherent nature of bidirectional planar dipoles and redirects the radiating field inthe boresight direction. This design is particularly favored for its relatively wide bandwidth and mono-lithic fabrication. The planar dipole antenna topology illustrated in Fig. 31b is typically applied inapplications where an end-fire antenna mode featuring low height profile and wide beamwidth in theelevation plan (“fan-beam patterns” are preferred. Conventional planar dipole antennas feature antenna

Fig. 30 Transmit array antenna (Ryan et al. 2010)


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gain values of less than 2 dBi which in turn results in relatively low EIRP for mmWave communications.The EIRP can be doubled when the mmWave planar dipole antenna is used in conjunction with amulti-RFchain phased-array transceiver IC. The half-wavelength balun is completely eliminated and each branchof the planar dipole antenna is respectively connected to each RFIC signal port. Assuming sufficientangular resolution is available by the RF phase shifter, a 180� phase difference between the two RF signalscan be fed to each branch. The phase difference drives the planar dipole antenna topology into an antennamode with a 3 dB enhancement in the radiated power.

Despite the numerous benefits, Yagi-Uda and planar dipole antennas are often subject to performancedegradations in real-life mmWave applications. One of the most notable issues can be attributed variousmetallic layers which may be implemented underneath the antenna module. For compact mobile terminalssuch as smartphones and wearable devices, the footprint of the wireless module is extremely limited andthe ensuing circuitries are densely spaced. The end-fire mmWave antenna will have to be placed in veryclose proximity to other signal tracelines and metallic layers for reference ground and heat sink. Yagi-Udaand planar dipole antennas inherently require the removal of metallic layers beneath the antenna topologyin order to operate in the end-fire radiation mode. In the presence of a metallic ground underneath themmWave Yagi-Uda and planar dipole antenna, the far-field radiation patterns may experience severedistortions as illustrated in Fig. 32. Methods of mitigating this issue will be discussed in detail in section“Notable mmWave Antenna Applications.”

Pattern-Reconfigurable mmWave AntennasWith the advent of durable, efficient active components such as phase shifters, switches, PIN diodes, andmicroelectromechanical systems (MEMS), reconfigurable antennas are increasingly becoming a practicalsolution for mmWave communications. In Yang et al. (2005), the patch antenna is composed of twoorthogonal quasi-Yagi-Uda antennas printed on the antenna substrate material. Electromagnetic bandgap(EBG) structures are controlled and modified depending on the state of the MEMS switches.

Another state-of-the-art pattern-reconfigurable technique proposed in Marnat et al. (2013) controls theantenna pattern by reorienting the antenna geometry as illustrated in Fig. 33. A broadside antenna mode isenabled when the mmWave bow-tie antenna is placed in parallel to the antenna substrate material. Bycontrolling the buckled cantilever plate (BCP), the designed antenna can be mechanically rotated by 90�

to transform into an end-fire antenna mode. For low-latency Gbps communications, the relatively slow

Fig. 31 (a) Yagi-Uda antenna. (b) Planar dipole antenna


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switching speed needs to be further researched as well as addressing the fragility of the entire topology.For high-volume applications, it is imperative for pattern-reconfigurable antennas to feature a small realestate and efficient packaging schematics. Consequently, future active components required for mmWavepattern-reconfigurable antennas will most likely need to be integrated inside the RFIC as opposed tomajority of the present-day solutions which are based on discrete, off-chip active components. It isimperative that the mmWave antenna topology must be conceived concurrently with the entire RF front-end architecture.

Fig. 32 End-fire pattern distortion when implementing Yagi-Uda and planar dipole antennas above a metallic surface. Solidline: elevation pattern in free space. Dotted lines: elevation pattern with the inclusion of the metallic surface

Fig. 33 Photographs of the pattern-reconfigurable mmWave bow-tie antenna (Marnat et al. 2013)


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Notable mmWave Antenna Applications

This section will present two mmWave antenna designs in detail which are respectively applied toongoing progress of Gbps mmWave communication applications in the global wireless and consumerelectronics industry. Many of the antenna designs introduced in this section have either been incorporatedor are projected to be applied to present and upcoming wireless mobile terminals. The readers arerecommended to refer to Hong et al. (2012, 2013, 2014b) for further detail.

Phased-Array Antenna for 60GHz WiGig/IEEE 802.11ad

Introduction and BackgroundThe explosive proliferation of wireless local area networks (WLAN) and high-quality multimedia datademands Gbps wireless data communication with extremely latencies less than 1 ums. Among numerousIEEE wireless standards, the WiGig/IEEE 802.11ad specializes in ultrafast data transfer and seamlessstreaming of audio and video files at close-range communication scenarios using the unlicensed 60 GHzfrequency band.

The concept and notion of utilizing 60 GHz has been perceived more than two decades ago amongresearch institutions and universities and have been followed with rigorous proof-of-concept (PoC)research and investigations. Over the years, 60 GHz antennas have continued to advance at an unprec-edented pace. Consequently, as antenna element technology begins to mature, the main focus of researchand development efforts has shifted to the realization of low-profile, high-performance 60 GHz antennamodules. Articulating the balance between performance and cost competitiveness is increasingly crucial.

Antenna Module ConfigurationTo mitigate the high cost associated with advanced antenna materials and ball-grid-array (BGA) assem-blies, the state-of-the-art 60 GHz antenna design employs a two-step wireless module integrationschematic introduced in section “Key mmWave Antenna Design Techniques.” Figure 34 illustrates thelamination configuration. The 60-GHz array antenna package is wholly designed using HDI FR-4 PCBsubstrates. Consequently, the antenna module is homogeneously embedded within the 12 layer FR-4carrier board, enabling a simplified, monolithic fabrication and assembly. The 60-GHz RFIC is flip-chipattached to the bottom layer M12 of the antenna module while lumped elements are implemented throughSMT. The sequence between the two assembly processes is interchangeable. The lamination of the

Fig. 34 Lamination configuration of the 60 GHz AiP (Hong et al. 2013)


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antenna module consists of one substrate core and the prepregs. The substrate core is situated at the centerof the lamination. Five prepregs are adhesively bonded together using 0.254 mm thick epoxy resins andstacked above the substrate core. Similarly, another five prepregs are adhesively bonded underneath thesubstrate core. Metal layers from M2 to M11 are implemented using 20 m thick copper. Metal layers M1andM12 are each composed of 25 m thick Au plating and finished with 20 m thick solder resist. Laser andmechanical micromachining processes are utilized for the construction of through, buried, and stacked viaholes for each substrate prior to lamination.

It is rarity for PCB vendors to characterize the electrical properties of the antenna substrate materialbeyond 10 GHz. The inherent frequency-dependent nature of pertinent material properties such aspermittivity and dielectric loss tangent must be well defined prior to the 60 GHz antenna design process.The deviations in the electrical characteristics between the core substrate and prepregs are reported by theFR-4 supplier to be well less than 1 % up to 10 GHz. Therefore for brevity, the FR-4 PCB substrate ismodeled as a homogeneous, isotropic material up to 60 GHz in this subsection. A number of weaklycoupled, coplanar waveguide (CPW)-fed microstrip ring resonators are designed to extract the permit-tivity of the FR-4 substrate. The top view of the ring resonator layout situated on M1 is visualized inFig. 35. Design parameters are optimized to be wr = 0.2 mm, sr = 0.1, tr = 0.2 mm, str = 1.1 mm, andDr = 5.18 mm. The bottom ground is implemented on M4. One such measured and simulated S21amplitudes are presented in Fig. 36. Measurement results indicate the calculated average permittivityer = 3.92 � 0.13 among three identical ring resonator samples at 58.3 GHz. Calculation of the losstangent at 60 GHz is difficult due to limitations in accurate extraction of conductor and radiation loss. As aresult, the simulated loss tangent is iteratively matched to the measured results of various transmissionlines. The loss tangent is confirmed to be tand = 0.027 � 0.07 at 60 GHz.

The antenna module is largely classified into three distinct portions based on functionality. Theantennas are allocated in the upper portion of the module to avoid interference with the RFIC, occupyinglayers fromM1 to M7. The antenna feedlines which route the 60-GHz signal paths from each of the RFICports to the corresponding antenna elements are situated in the mid-portion of the antenna module,ranging from M7 to M9. Lastly, power distributed networks (PDN), control lines, low-speed andhigh-speed interfaces for the RFIC, and digital blocks are located on the bottom portion of the antennamodule occupying layers at M10 and M11. The antenna ports which are each electrically connected tocorresponding RFIC ports through flip-chip assembly are designed onM12. M7 andM9 are designated asRF grounds to maximally isolate the antennas and antenna feedlines from unexpected spurious modes andcross talks that may occur in the bottom portion of the antenna module after integration with the RFIC.

Fig. 35 Topology of the microstrip ring resonator (Hong et al. 2013)


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60 GHz Antenna Element DesignA circular stacked patch antenna model discussed in section “KeymmWave Antenna Design Techniques”is devised to achieve the required �10 dB S11 bandwidth from 57 to 66 GHz and broadside radiationcharacteristic while keeping the lateral footprint to be minimal. Figure 37 illustrates the detailed topology.Two separate circular patches are designed and implemented on M1 and M4 and denoted as the top andbottom patch, respectively. The top and bottom patches are vertically aligned in a way that their centercoordinates are identical in the xy plane. A probe-fed mechanism is realized for the bottom patch byconnecting a signal via hole at a certain distance dp from the center of the bottom patch. As a result, a first

Fig. 36 Measured and simulated S-parameter amplitudes of the microstrip ring resonator (Hong et al. 2013)

Fig. 37 Illustration of the antenna element. (a) 3D view. (b) Top view (Hong et al. 2013)


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resonance is triggered by the bottom patch. The top patch is implemented above the bottom patch with thedistance that equals the combined thickness prepreg substrates between M1 and M4. The two patches arenot physically in contact with one another. The configuration produces a secondary parasitic resonancethat is primarily controlled by the top patch antenna dimension. In comparison to advanced antennamaterials such as LTCC or LCP, FR-4 PCB is relatively more prone to substrate thickness variationsduring the lamination process due to its inferior mechanical and thermal properties such as Young’smodulus and CTE (coefficient of thermal expansion). Any unexpected variations and especially reductionin the combined thickness of the prepreg substrates during the fabrication process will have a direct effecton the coupling capacitance between the top and bottom patches. Therefore a parametric study regardingthe location of the signal via hole (dp) and the radii of the top and bottom patches rtop and rbottom isperformed using numerical simulators to find the optimal design parameter combination for situationswhere the combined prepreg substrates change within the range of�30 um. Minimum bandwidth buffersare incorporated into the target design specification in preparation of any frequency deviations caused byvarious possible scenarios such as fabrication error, measurement discrepancies, and numerical errors.The finalized design parameters of the circular stacked patch antenna are dp = 0.42 mm, rtop = 0.61 mm,and rbottom = 0.64 mm.

An approach involving the utilization of fence of ground via holes surrounding the antenna elementintroduced is adapted in this design due to its relatively small profile and ease of implementation. First, aclearance with a radius rclear = 1.03 mm which encompasses the bottom patch is designed. The regionoutside the clearance in M4 is metallized to function as an RF ground. The dimension of rclear is found tohave limited effect on the reflection coefficient of the antenna element under conditions that remainsgreater than 0.95 mm. Afterward, fences of ground via holes penetrating from M4 to M7 are designed tosurround the antenna element with a circular radius rfence of 1.15 mm and via hole pitch dpitch = 0.344mm. The diameter of the ground via holes in this subsection is set to dvia = 75 um throughout thissubsection. Figure 38 presents the simulated electric fields and the corresponding three-dimensionalradiation patterns of the antenna element as a function of the presence or absence of the designed fence ofground via holes. In the presence of implementing fence of ground via holes, it can be observed that thefringing fields on the edges of the FR-4 substrate are relatively reduced in magnitude. The effect of thefence of ground via holes is further ascertained by the reduction levels of the ripples in the simulated three-dimensional radiation patterns throughout the frequency of interest.

The relatively high material loss tangent of the FR-4 substrate at mmWave frequencies has been theprimary restricting factor for widespread usage at 60 GHz frequency band. Excessive levels of signal lossincurred by the antenna feedline can potentially have an adverse effect on the overall system link budget,resulting in a more stringent design requirement for the RFIC. Therefore it is imperative to first closelyexamine the impedance matching and insertion loss properties of the antenna feedline structure prior tointegration with the antenna element.

A stand-alone topology of the antenna feedline is first designed for verification purpose as illustrated inFig. 39. The antenna feedline consists of two vertical coaxial transitions that are each connected to thecorresponding ends of a stripline structure located in M8. The impedance of the stripline is designed to50 Ω by adjusting the widthWst = 50 um. Fences of ground via holes are implemented along the edges ofthe stripline with via hole-to-stripline distance dstrip = 150 um and minimum via hole-to-via hole pitchdpitch1 � 310 um. Careful modeling of the fence of ground via holes surrounding the stripline isdetermined to be an extremely critical factor in preventing two commonly occurring degradations:(1) anomalous resonances (“suck out”) and (2) mutual coupling and signal cross talk. The two verticalcoaxial transitions are identically designed using one 75 um diameter signal via hole and a series ofground via holes situated around the signal via hole with dpitch2 = 452 um, functioning as a coaxial outerconductor. This configuration translates to a calculated input impedance of less than 26.5 Ω. The


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simulated insertion loss is approximately 0.3 dB due to the small electrical height of the verticaltransitions. Two GSG (ground-signal-ground) ports are connected to each of the vertical transitions atM1 andM12, respectively. Simulation indicates negligible difference when the two GSG ports are locatedon the same layer such as M12. Four antenna feedline test samples with length L = 5, 10, 15, 20 mm arecollectively fabricated. The opposite locations of the two GSG ports hinder the usage of conventional RFprobe stations. A multi-port in-house RF probe station which is capable of performing simultaneousS-parameter measurements in both the front and backside of the DUT (device under test) is custom

Fig. 39 Measured S-parameter amplitudes of the antenna feedlines (Hong et al. 2013)

dB (GainTotal)a

b

4.9941e+0003.4319e+0001.8698e+000

−1.2545e+000−2.8166e+000−4.3787e+000−5.9408e+000−7.5030e+000−9.0651e+000−1.0627e+001−1.2189e+001−1.3751e+001−1.5314e+001−1.6876e+001−1.8438e+001−2.0000e+001

3.0766e−001

dB (RealizedGain)5.5790e+0003.9803e+0002.3816e+000

−8.1575e−001−2.4144e+000−4.0131e+000−5.6118e+000−7.2105e+000−8.8092e+000−1.0408e+001−1.2007e+001−1.3605e+001−1.5204e+001−1.6803e+001−1.8401e+001−2.0000e+001

7.8294e−001

Fig. 38 Simulated electric fields and 3D radiation patterns. (a) Without fence of ground via holes. (b) With fence of ground viaholes (Hong et al. 2013)


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designed and built. Short-open-load-thru (SOLT) on-wafer calibration is performed prior to S-parameterillustration of the antenna feedline: (a) 3D view and (b) top view.

The measured amplitude of the S-parameters of the fabricated antenna feedline test samples are plottedin Fig. 40. It can be observed that the devised antenna feedline structure features less than �10 dB inputreflection coefficients (S11, S22) within the frequency of interest. From the insertion loss (S21, S12) of thefour test samples, the average unit loss is calculated to be 0.55 dB/mm. The relatively high unit losscompared to LTCC- or LCP-based antenna feedlines is primarily attributed by the loss tangent of the FR-4substrate.

Figure 41 shows the measured and simulated far-field radiation patterns of the antenna element. Due tomechanical restrictions of the antenna measurement setup, the radiation patterns are measured within therange of �90� in the elevation plane. The measured antenna element features an omnidirectionalco-polarized radiation patterns in both the E- and H-planes as predicted by HFSS simulations. Cross-polarized measured and simulated radiation patterns indicate a much weaker correlation due to thefollowing attributes: (1) the omission of the antenna measurement environment such as RF cables,adaptors, and chuck in the simulated numerical simulation model due to computation limitations;(2) measurement errors due to various scatterers such as RF cables, chuck, RF probes, etc.; and (3) limiteddynamic range of the antenna measurement setup. Additional measures are being investigated to improvethe accuracy between the two results. A pair of identical standard gain V-band horn antennas is used tomeasure the gain of the antenna element at boresight. First a CW signal is transmitted from the TX hornantenna and the RX horn antenna is used to record the received power. Afterward, the RF horn antenna isreplaced with the AUT, and the received power is recorded in a similar manner. Losses in the RF probe,cables, and connectors are calibrated out during the gain calculation process. The loss incurred in theantenna feedline is not subtracted. The measured average gain is determined to be 4.8 dBi. Using thedirectivity derived from the simulated far-field radiation patterns, the average radiation efficiency of thedevised antenna element is calculated to be approximately 71 % between 57 and 66 GHz.

Phased-Array Antenna ModuleThe designed antenna element and antenna feedline structure is employed to devise a 4 � 2 antenna array,composed of eight antenna elements within the 60 GHz antenna module for phased-array applications.Each of the antenna elements are independently connected with corresponding GSG ports located inM12.Therefore the antenna array also consists of eight independent antenna feedlines. A 20 � 20 mm2 regionat the center of the FR-4 carrier board is allocated for direct integration of the antenna array. Details of the

Fig. 40 Measured S-parameters of the fabricated antenna feedline test samples


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PDN and low-speed and high-speed interfaces are included in the modeling and simulation process of the60 GHz array antenna as illustrated in Fig. 42. The antenna element spacing are configured to dant = 2.5mm in both the x and y direction to achieve minimum mutual coupling while avoiding undesired gratinglobes during beam steering scenarios. The physical dimensions of the antenna elements which composethe array antenna module remain identical.

The antenna array feedlines are designed to exhibit identical phase properties and minimum insertionloss. Figure 43 presents the sliced top view of the modeled antenna array feedline layout. The antennaarray feedlines are modeled using a commercial CAD program in a quadrant symmetric manner. Themaximum deviation among the antenna array feedlines is 14 um. Fences of ground via holes are denselyimplemented with maximum via hole-to-via hole pitch of 310 um. The ground via holes are determined tobe instrumental in shielding the antenna array feedlines from undesired spurious emissions that occur

Fig. 41 Measured and simulated far-field radiation patterns of the antenna element. (a) E-plane. (b) H-plane. Solid: measuredCo-pol. Dash: simulated Co-pol. Dotted: measured X-pol. Dash-dot-dot: simulated X-pol (Hong et al. 2013)

Fig. 42 Simulation setup of the eight-element antenna array. The region of the FR-4 carrier board is not shown for brevity(Hong et al. 2013)


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from various active components that are implemented on the FR-4 carrier board. Simulation andmeasurement indicate the stand-alone model of the antenna array feedlines feature |S11| < �9.6 dB and|S21| < 2.2 dB between 57 and 66 GHz. The maximummeasured phase deviation is determined to be lessthan 7.1�. Amplitudes of the input reflection coefficients of each of the antenna elements are measured oneat a time. During measurement, the rest of the GSG ports are configured to be open. Measurement result ofthe antenna array is presented in Fig. 44. It can be observed that all eight antenna elements within theantenna array exhibit a �10 dB bandwidth of more than 9 GHz from 57 to 66 GHz, sufficiently meetingthe bandwidth specification for WiGig/IEEE 802.11ad applications worldwide. Measurement errorsoccurred during the RF probing procedure and discrepancies between the antenna feedlines are identifiedas the primary cause for port-to-port variations.

Phased-array and beam steering characteristics are evaluated in the anechoic chamber using a far-fieldmeasurement setup. An enhanced serial interface (ESI) control board is connected to the antenna modulethrough wired interface. Custom-made software program installed on the computer that is connected tothe ESI board emulates the digital functions of the 60 GHz transceiver system. Beam steering is realizedby modulating the phase distributions of equal-amplitude 60 GHz signals that feed each of the antennaelements. Eight 2-bit phase shifters within the RFIC are modulated to produce specific phase distributionsfor the corresponding beam steering scenario based on a predetermined beam table. AV-band standard

Fig. 43 Sliced top view of the antenna array. M1 and M8 are overlapped (Hong et al. 2013)

Fig. 44 Measured S11 amplitude of the antenna array (Hong et al. 2013)


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gain horn antenna is used to transmit and receive a series of CW signals within the 57–66 GHz range toand from the DUT. The CW signals are up converted and down converted to 54 MHz intermediatefrequency (IF) in both the transmitting and receiving end. The normalized measured and simulatedfar-field radiation pattern of the antenna module is presented in Figs. 45 and 46 in the E- and H-planefor both boresight and preset beam tilt angle of�45� and 30�, respectively. The designed 60 GHz antennamodule exhibits 3 dB beamwidth of 28� and 50� at boresight in the E- and H-plane, respectively. Themaximum scan range is determined to be approximately 50� in both planes. The measured gain values atpreset beam tilt angle of �45� and 30 o are 11.3 dBi and 11.6 dBi at 61.5 GHz, respectively.

mmWave Antennas for 5G Cellular Phones

Introduction and BackgroundBased on the remarkable technological advancements, the cellular market sector continues to maintain ahealthy growth rate – according to the International Telecommunication Union (ITU), during 2013Mobile World Congress (MWC), worldwide cellular phone subscriptions will exceed world populationby 2014. Numerous approaches are being discussed and researched around the world to create a futurecellular network which would one day succeed the incumbent 4G/LTE network. While the data capacityof the 4G/LTE service continues to improve quite remarkably, classical information theory established byClaude Shannon and Ralph Hartley points out that the bottleneck in achieving the desired cellular Gbpsdata rate mostly lies on the currently crammed spectrum bandwidth. In retrospect, it becomes possible tointroduce an entirely new cellular technology by employing a higher frequency spectrumwhich contains a

Fig. 45 Normalized far-field radiation pattern in the E-plane. (a) Boresight. (b) �45� beam tilt. Solid: measured. Dash:simulated (Hong et al. 2013)


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much larger allocated signal bandwidth. The ongoing proliferation of small base stations denoted asmicrocells and picocells carried out by the cellular carriers becomes enormously instrumental in mitigat-ing the relatively high free-space loss caused by the atmospheric conditions and penetration at extremelyhigh frequencies. Adaptive beam steering technology discussed in section “Theory” can be utilized to getaround obstructions, signal blind spots, and other NLOS propagation conditions. Theoretical study andmeasurement results of a 5G prototype operating at millimeter wave have been presented and discussed indetail in Roh et al. (2014). In the aforementioned study, a 28 GHz prototype exhibits a sharp radiationbeam consisting of an array of multiple antennas which continually sweep the atmosphere to identify thestrongest connection.

The implementation of a mmWave transceiver within a cellular phone introduces unprecedentedchallenges that are far from trivial. The enormity of this task is particularly exacerbated by the uniquenature of user scenarios associated with cellular phones. In comparison to portable computers and wirelessdocking stations, cellular phones are much more compact in size and its user scenarios are highlydiversified owing to the proliferation of smartphones. The 5G cellular phone must guarantee secure andreliable link accessibility of Gbps speed anywhere, anytime. Among key elements constituting themmWave 5G radio, the antenna requires one of the most radical changes design wise. This is due to therelatively simple fact that every cellular standard up to present has operated below the 3 GHz spectrum.Present-day cellular antennas can be classified as electrically small antennas and are inherently omnidi-rectional. The antenna gain of these antennas typically falls in the range of�8 to 0 dBi. The much higher

Fig. 46 Normalized far-field radiation pattern in the H-plane. (a) Boresight. (b) 30o beam tilt. Solid: measured. Dash:simulated (Hong et al. 2013)


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gain that is needed to compensate the higher signal attenuation at mmWave frequencies introduces theconcept of antenna arrays for cellular phones for the first time in the history of wireless communications.In this subsection, pragmatic methodologies to design and selectively utilize up to 32 antenna elementsoperating at 28 GHz within a future 5G cellular prototype are demonstrated. A number of key antennadesign considerations that stem from the unique nature of mmWave bands are discussed in detail. Basedon these considerations, a novel but practical 28 GHz phased array is designed and illustrated. While it isimportant to note that the 28 GHz carrier frequency has yet to be ratified for 5G communications, thisdesign technique can be scaled to other mmWave frequencies in future applications.

Key Considerations for mmWave 5G AntennaDetermining the optimum placement of the 28 GHz antenna within a cellular handset is one of the mostdefining factors prior to the actual design. While it is difficult to accurately envision a future cellularhandset device down to the detail, it is projected that radio functions will closely resemble its present-daycounterpart. A conventional cellular phone chassis is examined and classified into a number of keysegments as presented in Fig. 47. Major segments such as the rear case, battery cover, and the main PCBare predominantly based on polycarbonate and FR-4. For these materials, the loss tangent versusfrequency relation is coherent with studies reported in (Hong et al. 2013). Therefore it would be validto predict that the gain of the 28 GHz antenna will likely experience a non-negligible degradation whenplaced within the cellular handset device. A loss of 1.4 dB in gain of a rudimentary 28 GHz patch antennais calculated when placed within in the top left corner of the presented chassis using electromagneticnumerical simulators. The metallic segments within the cellular handset such as the LCD bracket, LCD,and the battery are expected to be perceived as a finite conductive layer at 28 GHz. Consequently, thedesigned antenna must be implemented within the 5G cellular phone which excludes these regions.

The placement of the 28 GHz antenna is further complicated by the uncertainty of the user’s hand. At28 GHz, the user’s body functions as a formidable absorber which is expected to have negative effects onthe gain, radiation pattern, and return loss of the antenna. Figure 48 presents a simulation of theperformance degradation of a conventional broadside 28 GHz antenna array implemented within thepresented chassis. More than 9.5 dB decrease in gain is estimated when the user’s hand fully encompasses

Fig. 47 Illustration of the material properties of the major segments of a conventional cellular handset (Hong et al. 2014a)


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the region of the 28 GHz antenna array. One way of alleviating the potential performance risk caused bythe user’s hand would be to implement plural number of antennas within the cellular device.

Mobility requires cellular phones to operate virtually anywhere. It is statically and empiricallyimpossible to predict the angle of arrival (AoA) of the strongest signal path. The omnidirectional patternsof 3G/4G cellular phone antennas today are advantageous in this regard. Likewise at mmWave frequen-cies, the incoming waves impinging on the antenna are predicted to be distributed across the entire sphere.This introduces a paradoxical situation for the antenna engineer that given a finite amount of energy, theantenna needs to radiate with extremely high intensity around the sphere. Since simultaneous radiation atall angles violates the classical laws of physics, phased-array beam steering as that employed in theprevious subsection is applied.

In this subsection, Cartesian coordinates (x,y,z) and spherical coordinates y (ranging from 0 to p anddefined as the elevation angle) and j(ranging from 0 to 2p and defined as the azimuth angle) will be usedto describe the maximum allowable trajectory of the main lobe of the phased-array antenna. Based on therealistic physical limitations of antenna placement options within a cellular phone as described earlier,two-dimensional (planar) phased-array topology is deduced. With selected withstanding exceptions, theantenna elements are placed at least half wavelength apart to avoid spatial aliasing or also known asgrating lobes.

For 1 � N antenna array geometries, the scanning angle is a function of y or j. The scanning angle is afunction of y andj for N � M antenna geometries. In addition, the scanning angle range of the main lobeis inversely related to the number of antenna elements (i.e., antenna array gain). Due to the high signalattenuation at 28 GHz, the antenna must be placed in very close proximity to the 28 GHz RFIC and thefront-end module. Implementing the antenna array directly on the PCB of the 5G cellular device willtherefore keep the insertion loss between the antenna and RFIC to be minimal. This implies that aplacement of the RF blocks within the 5G architecture before the intermediate frequency (IF) stage will bedependent on the placement of the 28 GHz antenna array within the cellular phone. Collectivelyconsidering the aforementioned design environment, a minimum set of two 28 GHz antenna arrays areconceived for mmWave 5G cellular phones. As presented in Fig. 49, the two antenna arrays areimplemented on the top and bottom portion of the cellular device. This is somewhat of an intuitiveapproach since similar to 3G/4G antennas, the large LCD display panel and the battery prohibit theantenna arrays from being placed in the center region of the cellular phone. This topology is applicable forboth single-input single-out (SISO) and multiple-input multiple-output (MIMO) systems. Each 5Gantenna array located on the top and bottom portion of the cellular phone is designed to respectively

Fig. 48 Simulation of the effect of the user’s hand on a rudimentary 28 GHz antenna array placed within a cellular handset(Hong et al. 2014a)


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cover the corresponding hemisphere in the form of spatial diversity. For SISO, the secondary antennaarray becomes instrumental when a secure link cannot be established using the primary antenna.

It is a rarity for thickness of cellular phone PCBs to exceed 1 mm. In the fiercely competitive consumerelectronics arena, increasing the PCB thickness is directly related to the rise in production costs andbulkier designs. At 28 GHz, the free-space half wavelength is slightly larger than 10 mm. This discrep-ancy prohibits phased-array beam steering in the elevation plane (zx plane) and leads us to aone-dimensional beam steering approach using a 1 � N antenna array topology. Moreover, a verybroad beamwidth is required in the elevation plane to compensate for the inability to beam steer.

mmWave 5G Mesh-Grid Antenna Array DesignIn order to synthesize a highly directive fan beam, it is imperative to maximize the allowable number ofantenna elements inside the cellular phone. Avoiding unnecessary loss in the antenna radiation is critical atmmWave since this is correlated to the wireless link budget and eventually the power consumption.Therefore each of the antenna elements is designed to be at least a quarter of lg large, which is theminimum threshold before the radiation efficiency starts to sharply deteriorate due to the rise of storedenergy in the near-field region of the antenna. As mentioned in section “Key mmWave Antenna DesignTechniques,” mmWave planar dipoles are by far the most popular fan-beam antennas in the wirelessindustry. However, as it was also discussed in section “Key mmWave Antenna Design Techniques,” thefan-beam radiation characteristics of planar dipole are preserved under the condition that the metallicpatterns and tracelines are removed below the planar dipole topology. The elimination of the metallicpatterns becomes problematic especially with the advent of smartphones due to the increased degree ofintegration of the PCB. Cellular phone PCBs which are less than 1 mm thick typically consists of 6–12layers that are tens of mm in thickness respectfully. Within that space run various signal lines ranging fromlow-speed to high-speed interfaces and power distributed networks. Removal of the signal line traces forthe implementation of the half-guided wavelength planar dipole structure puts a serious strain on the PCBlayout. As an alternative, a low-profile antenna design which can coexist with the signal line traces and yetexhibit a fan-beam radiation characteristic is designed.

Fig. 49 The 28 GHz antenna array configuration for 5G cellular phones and its comparison with 4G standard (Honget al. 2014b)


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Illustrated in Fig. 50, in contrast to planar dipole antennas that spans across the xy plane, the newlydevised structure is situated along the zx plane which can reduce the planar antenna footprint from theprevious half lg to a fraction of a hundreds of a lg. The real estate in the xy plane is solely determined bythe length of the dual antenna signal feed lines denoted as dp. The entire radiating body of the antennastructure consists of tightly spaced via holes creating a mesh type grid as it can be observed from the zxplane. Hence, this antenna topology is named as the mesh-grid patch antenna by the authors. The physicaldimension of the mesh-grid patch antenna ranges from quarter to half of the lg along the x-axis. The heightdenoted as h determines the number of PCB layers required for the specific mesh-grid patch antennadesign and affects the elevation beamwidth in the y angle. The mesh-grid antenna is designed to supportmulti-polarized MIMO based on any combination consisting of vertical, horizontal, and circular polar-izations as illustrated in Fig. 51. Two antenna feedlines are configured to respectively enable horizontaland vertical polarizations. The polarization configurations are summarized in Table 2. Figure 52 presentsthe simulated and measured far-field radiation pattern of the mesh-grid patch antenna in the elevationplane. The measured �10 dB return loss bandwidth is approximately 1 GHz with a center frequency of27.9 GHz.

The mesh-grid antenna elements are expanded and arranged in two sets of 1 � 16 phased arrays in thetop and bottom portion of the cellular phone PCB as illustrated in Fig. 53. The high antenna gain realizedby a reconfigurable and scalable array consisting of 16 antenna elements alleviates the linearity and powerconsumption requirements of the 5G amplifiers. The mesh-grid antenna elements situated at each cornerof the PCB are arranged in slanted angles of approximately 50�. The conformal topology furthermaximizes the range of the beams steering scanning angles in the azimuth plane. Moreover, the slantedtopology conforms to the cellular device and enables the designed mmWave antenna array to appear as anextremely low-profile metallic trace lines which encompasses around the edges of the PCB. The width ofthe trace lines is less than 0.2 mm which is even less than the 1 mm spacing required from the PCB edgesfor conventional surface-mount technologies (SMT). From the vantage point of the hardware layout, theinclusion of a total of 32 mmWave antenna elements requires negligible antenna footprint. Based uponthis antenna solution, a truly massive MIMO antenna system may be realizable for mmWave 5G in thelong term.

Fig. 50 (a) Simplified comparison between conventional cellular antenna and proposed mmWave 5G antenna topology. (b)Detailed view of the mesh-grid patch antenna (Hong et al. 2014b)


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Themeasured and normalized radiation patterns of the mesh-grid antenna array in the azimuth plane areplotted for different main beam scanning angles in Fig. 54. The antenna array prototype is first measuredin a stand-alone state denoted as free space. Afterward, the measurement is repeated after the AUT isintegrated within the cellular phone prototype. Measurements ascertain a scanning angle range ofapproximately �70o. By combining the two antenna arrays, nearly spherical radiation coverage can be

Fig. 51 Illustration of the effect of antenna alignment configuration on the polarization mismatch for mmWave-equippedcellular phones

Table 2 Polarization configuration of the dual-feed mesh-grid patch antenna

Antenna polarization Horizontal polarization feedline state Vertical polarization feedline state

Horizontal On Off

Vertical Off On

Diagonal On On

Circular On On+90�

Fig. 52 Simulated and measured far-field radiation pattern of the mmWave mesh-grid patch antenna in the elevation plane.Solid line: simulation. Dotted line: measurement


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attained. The radiation pattern discrepancies between the free space and integrated condition indicates theoccurrence of pattern deformation caused by diffracted and refracted electromagnetic waves between theantenna elements and the cellular phone chassis. The discrepancy level is expected to vary as a function ofthe design and material composition of the cellular device. The 3 dB beamwidth in the azimuth plane isapproximately 12� generating a null-free beam scanning coverage. The peak gain of the antenna arraywithin the cellular device is calculated to be more than 10.5 dBi. Major loss factors include the 2.8 dB loss

Fig. 53 Photographs of the mmWave 5G antenna system prototype integrated inside a cellular phone and zoomed in views ofthe mmWave antenna region (Hong et al. 2014b)

Fig. 54 Measured and normalized radiation patterns of the 16-element mesh-grid antenna array in the horizontal polarizedconfiguration (Hong et al. 2014b)


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from the mesh-grid antenna feed point to the K-type coaxial connector, 2.2 dB cable loss between theK-type coaxial connector and the RF unit, and the estimated 1.8 dB dielectric loss incurred by thepolycarbonate chassis of the cellular phone prototype used in the measurement.

Biological Implications and ConsiderationsAssessing the biological implications on the user may be one of the most important parameters formmWave 5G cellular devices in the future. The level of electromagnetic absorption by the human body isstrictly and universally regulated by governmental bodies under the specific absorption rate (SAR)guideline. Every cellular device must pass a regulated SAR test prior to certification. Likewise althoughpermissible SAR levels at mmWave frequencies do not exist at this point in time, it is possible to analyzeand compare mmWave SAR levels with that of 3G/4G cellular phones. To investigate this, the measuredradiation patterns of the two sets of mesh-grid antenna arrays are imported into a commercial numericalsimulator.

With the inclusion of a human head phantom, the average SAR levels are calculated. As it can beobserved in Fig. 55, the electromagnetic fields absorbed by the user at 28 GHz tend to be localized incomparison to that of a 3G/4G cellular frequency band. The penetrated skin depth of the absorbed 28 GHzsignal is confirmed to be approximately 3 mm. In comparison the penetrated skin depth at 1.9 GHzexceeds 45 mm. This implies that majority of the absorbed energy is confined to the epidermis atmmWave. The average calculated SAR at 28 GHz for the antenna array situated in the bottom regionof the cellular phone is found to be less than a factor of 18 compared to SAR levels retrieved at 1.9 GHz.The ability to beam steer the main lobe is projected to further reduce the SAR levels in the future.

Conclusion

As discussed in this chapter the exploitation of mmWave frequencies is projected to become the backboneof future mobile networks. mmWave wireless communication aims to enable unprecedented and intuitive

Fig. 55 Analysis and comparison of specific absorption rates of 4G and mmWave 5G antenna systems (Hong et al. 2014b)


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user experience with enhanced accessibility to the general public. However, it is vital to emphasize thatmmWave spectrums will not render presently used microwave spectrum to become obsolete. The inherentlimitations of the mmWave signals imply coexistence with legacy wireless architectures, and integrationswill become an ever-important issue as widespread interest of mmWave communications continue togrow. In severe propagation conditions, mmWave radio systems will be designed to fall back toconventional radio systems to ensure the overall quality of service (QoS) of the wireless network. Asthe number of antennas required inside wireless mobile terminals such as cellular phones and small accesspoints continue to increase on a yearly basis, the limited space constraints for mmWave antennas isprojected to become an major issue. Researching novel mmWave antenna topologies which are easier tointegrate with the rest of the legacy wireless RF system and the microwave frequency spectrum antennaswill be an important area. In addition, the demand for three-dimensional, reconfigurable mmWaveantennas is anticipated followed by the proliferation of wearable communication devices and theadvancements of three-dimensional printing technologies. The mmWave antenna design philosophymust prioritize and focus on the wireless system performance such as link reliability and powerconsumption efficiency.

Another key area where mmWave antenna technologies are projected to be vital is associated with theemergence of multi-node/massive MIMO transmissions and receptions. Massive MIMO antenna archi-tectures are composed of a very high number of antenna elements with the purpose of enhanced wirelesscapacity. They benefit from simplified MAC layers, enhanced latencies, and signal cell channel capacity.Devising efficient mmWave antenna topologies capable of three-dimensional, simultaneous multi-beamsteering that is scalable remains a notable technical challenge. Future massive MIMO antenna technologymust be investigated with full consideration of multi-heterogeneous networks (multi-HetNeT) which iscomposed of multiple carrier frequencies. The increasingly technical complexity related to interoperabil-ity and mutual dependency between the radio hardware and the wireless network researchers underlinesthe multidisciplinary nature of future mmWave antenna research. The anticipated Internet of Things (IoT)era will further introduce numerous wireless devices with diverse shapes and forms. The author predictsmmWave antennas will evolve from the current planar multilayer PCB types to conformal topologiesbased on novel materials and integration concepts.

Cross-References

▶Advanced Antenna Fabrication Process (MEMS/LTCC/LCP/printing)▶Antennas in Hand-held Devices▶Beam-Scanning Leaky Wave Antennas▶Broadband and Multiband Planar Antennas▶Grid Antenna Arrays▶Lens Antennas▶MIMO Systems and Antennas for Terminals (Including Portable Devices Such as Handsets, iPadLaptops)

▶Mm-Wave Sub-mm-Wave Antenna Measurement▶On-Chip Antennas▶ Phased Arrays▶Radio Frequency Beamforming for Scanned and Multibeam Antenna Systems General▶Reconfigurable Antenna Arrays for Wireless Communications▶RF Material Characterization


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http://link.springer.com/SpringerLink:ChapterTarget














▶ Substrate-Integrated Waveguide Antennas▶Taped Slot Antennas▶Waveguide Slot Antennas and Arrays

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