dual, wideband, high output current operational … 2 sbos126i specified package temperature package...

OPA2677

OPA2677

Dual, Wideband, High Output CurrentOperational Amplifier

OPA2677

SBOS126I – APRIL 2000 – REVISED JULY 2008

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Copyright © 2000-2008, Texas Instruments Incorporated

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications ofTexas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

FEATURES WIDEBAND +12V OPERATION: 200MHz (G = +4) UNITY-GAIN STABLE: 220MHz (G = +1)

HIGH OUTPUT CURRENT: 500mA OUTPUT VOLTAGE SWING: ±5V HIGH SLEW RATE: 1800V/µs LOW SUPPLY CURRENT: 18mA FLEXIBLE SUPPLY RANGE:

+5 to +12V Single Supply±2.5 to ±6V Dual Supplies

APPLICATIONS xDSL LINE DRIVER CABLE MODEM DRIVER MATCHED I/Q CHANNEL AMPLIFIER BROADBAND VIDEO LINE DRIVER ARB LINE DRIVER POWER LINE MODEM HIGH CAP LOAD DRIVER

DESCRIPTIONThe OPA2677 provides the high output current and low distortionrequired in emerging xDSL and Power Line Modem driverapplications. Operating on a single +12V supply, the OPA2677consumes a low 9mA/ch quiescent current to deliver a very high500mA output current. This output current supports even themost demanding ADSL CPE requirements with > 380mA mini-mum output current (+25°C minimum value) with low harmonicdistortion. Differential driver applications will deliver < –85dBcdistortion at the peak upstream power levels of full rate ADSL.The high 200MHz bandwidth will also support the most demand-ing VDSL line driver requirements.

Specified on ±6V supplies (to support +12V operation), theOPA2677 will also support a single +5V or dual ±5V supply.Video applications will benefit from its very high outputcurrent to drive up to 10 parallel video loads (15Ω) with< 0.1%/0.1° dG/dP nonlinearity.

The OPA2677 is available in either an SO-8 or QFN-16 and anHSOP-8 PowerPAD package.

OPA2677 RELATED PRODUCTS

SINGLES DUALS TRIPLES NOTES

OPA691 OPA2691 OPA3691 Single +12V Capable

— THS6042 — ±15V Capable

— OPA2674 — Single +12V Capablewith current limit

All trademarks are the property of their respective owners.

Single-Supply ADSL CPE Driver

82.5Ω2kΩ

2kΩ 1µF

17.4Ω

100Ω2VPP

AFEOutput

324Ω

20Ω

324Ω

1/2OPA2677

1/2OPA2677

+12V

1:1.7

15VPP Twisted Pair17.7VPP

20Ω

17.4Ω

+6.0V

PRODUCTION DATA information is current as of publication date.Products conform to specifications per the terms of Texas Instrumentsstandard warranty. Production processing does not necessarily includetesting of all parameters.

www.ti.com

OPA26772SBOS126Iwww.ti.com

SPECIFIEDPACKAGE TEMPERATURE PACKAGE ORDERING TRANSPORT

PRODUCT PACKAGE-LEAD DESIGNATOR RANGE MARKING NUMBER MEDIA, QUANTITY

OPA2677 SO-8 D –40°C to +85°C OPA2677U OPA2677U Rails, 100" " " " " OPA2677U/2K5 Tape and Reel, 2500

OPA2677 HSOP-8 DDA –40°C to +85°C OP2677 OPA2677IDDA Rails, 75" " " " " OPA2677IDDAR Tape and Reel, 2500

OPA2677 QFN-16 RGV –40°C to +85°C OPA2677 OPA2677IRGVT Tape and Reel, 250OPA2677IRGVR Tape and Reel, 2500

ABSOLUTE MAXIMUM RATINGS(1)

Power Supply ............................................................................... ±6.5VDC

Internal Power Dissipation .......................... See Thermal CharacteristicsDifferential Input Voltage .................................................................. ±1.2VInput Common-Mode Voltage Range ................................................. ±VS

Storage Temperature Range: U, DDA, RGV ................. –65°C to +125°CLead Temperature (soldering, 10s) .............................................. +300°CJunction Temperature (TJ ) ........................................................... +150°CESD Rating:

Human Body Model (HBM)(2) ....................................................... 2000VCharge Device Model (CDM) ....................................................... 1000VMachine Model (MM) ..................................................................... 100V

NOTES: (1) Stresses above these ratings may cause permanent damage.Exposure to absolute maximum conditions for extended periodsmay degrade device reliability.

(2) Pins 2 and 6 on SO-8 and HSOP-8 packages, and pins 2 and11 on QFN-16 package > 500V HBM.

PIN CONFIGURATIONS

Top View

ELECTROSTATICDISCHARGE SENSITIVITY

This integrated circuit can be damaged by ESD. Texas Instru-ments recommends that all integrated circuits be handled withappropriate precautions. Failure to observe proper handlingand installation procedures can cause damage.

ESD damage can range from subtle performance degradationto complete device failure. Precision integrated circuits may bemore susceptible to damage because very small parametricchanges could cause the device not to meet its publishedspecifications.

PACKAGE/ORDERING INFORMATION(1)

NOTE: (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site atwww.ti.com.

1

2

3

4

12

11

10

9

NC

–In B

+In B

NC

16 15 14 13

Out

A

NC

+V

S

Out

B

5 6 7 8

NC

NC

–VS

NC

OPA2677RGV

QFN-16

NC

–In A

+In A

NC

1

2

3

4

8

7

6

5

+VS

Out B

–In B

+In B

OPA2677U, DDA

SO-8, HSOP-8

Out A

–In A

+In A

–VS

NOTE: Exposed thermal pad on HSOP-8 and QFN-16 must be tied to –VS.

OPA2677 3SBOS126I www.ti.com

OPA2677U, DDA, RGV

TYP

0°C to –40°C to MIN/ TESTPARAMETER CONDITIONS +25°C +25°C(1) 70°C(2) +85°C(2) UNITS MAX LEVEL(3)

AC PERFORMANCE (see Figure 1)Small-Signal Bandwidth (VO = 0.5VPP) G = +1, RF = 511Ω 220 MHz min B

G = +2, RF = 475Ω 200 170 168 165 MHz min BG = +4, RF = 402Ω 200 170 168 165 MHz min BG = +8, RF = 250Ω 250 225 205 200 MHz min B

Peaking at a Gain of +1 G = +1, RF = 511Ω 0 dB typ CBandwidth for 0.1dB Gain Flatness G = +4, VO = 0.5VPP 80 36 32 30 MHz min BLarge-Signal Bandwidth G = +4, VO = 5VPP 200 MHz typ CSlew Rate G = +4, 5V Step 2000 1500 1450 1400 V/µs min BRise-and-Fall Time G = +4, VO = 2V Step 1.75 ns typ CHarmonic Distortion G = +4, f = 5MHz, VO = 2VPP

2nd-Harmonic RL = 100Ω –72 –70 –69 –68 dBc max BRL ≥ 500Ω –82 –80 –79 –78 dBc max B

3rd-Harmonic RL = 100Ω –81 –80 –79 –78 dBc max BRL ≥ 500Ω –93 –91 –90 –89 dBc max B

Input Voltage Noise f > 1MHz 2 2.6 2.9 3.1 nV/√Hz max BNoninverting Input Current Noise f > 1MHz 16 20 21 22 pA/√Hz max BInverting Input Current Noise f > 1MHz 24 29 30 31 pA/√Hz max BNTSC Differential Gain NTSC, G = +2, RL = 150Ω 0.03 % typ C

NTSC, G = +2, RL = 37.5Ω 0.05 % typ CNTSC Differential Phase NTSC, G = +2, RL = 150Ω 0.01 degrees typ C

NTSC, G = +2, RL= 37.5Ω 0.04 degrees typ CChannel-to-Channel Crosstalk f = 5MHz, Input Referred –92 dB typ C

DC PERFORMANCE(4)

Open-Loop Transimpedance Gain VO = 0V, RL = 100Ω 135 80 76 75 kΩ min AInput Offset Voltage VCM = 0V ±1.0 ±4.5 ±5 ±5.3 mV max AAverage Offset Voltage Drift VCM = 0V ±4 ±10 ±10 ±12 µV/°C max BNoninverting Input Bias Current VCM = 0V ±10 ±30 ±32 ±35 µA max AAverage Noninverting Input Bias Current Drift VCM = 0V ±5 ±50 ±50 ±75 nA/°C max BInverting Input Bias Current VCM = 0V ±10 ±35 ±40 ±45 µA max AAverage Inverting Input Bias Current Drift VCM = 0V ±10 ±100 ±100 ±150 nA°/C max B

INPUT(4)

Common-Mode Input Range (CMIR)(5) ±4.5 ±4.1 ±4.0 ±3.9 V min ACommon-Mode Rejection Ratio (CMRR) VCM = 0V, Input Referred 55 51 50 49 dB min ANoninverting Input Impedance 250 || 2 kΩ || pF typ CMinimum Inverting Input Resistance Open-Loop 22 12 Ω min BMaximum Inverting Input Resistance Open-Loop 22 35 Ω max B

OUTPUT(4)

Voltage Output Swing No Load ±5.1 ±4.9 ±4.8 ±4.7 V min ARL = 100Ω ±5.0 ±4.8 ±4.7 ±4.5 V min ARL = 25Ω ±4.8 V typ C

Current Output VO = 0 ±500 ±380 ±350 ±320 mA min APeak Current Output, Sourcing(6) VO = 0 1.2 A typ CPeak Current Output, Sinking(6) VO = 0 –1.6 A typ CClosed-Loop Output Impedance G = +4, f ≤ 100kHz 0.003 Ω typ C

POWER SUPPLYSpecified Operating Voltage ±6 V typ CMaximum Operating Voltage ±6.3 ±6.3 ±6.3 V max AMinimum Operating Voltage ±2 V typ CMaximum Quiescent Current VS = ±6V, Both Channels 18 18.6 18.8 19.5 mA max AMinimum Quiescent Current VS = ±6V, Both Channels 18 17.4 16.5 16.0 mA min APower-Supply Rejection Ratio (PSRR) f = 100kHz, Input Referred 56 51 49 48 dB min A

THERMAL CHARACTERISTICSSpecification: U, DDA, RGV –40 to +85 °CThermal Resistance, θJA Junction-to-Ambient

U SO-8 125 °C/W typ CDDA HSOP-8 Exposed Slug Soldered to Board 55 °C/W typ CRGV QFN-16 Exposed Slug Soldered to Board 50(7) °C/W typ C

MIN/MAX OVER TEMPERATURE

ELECTRICAL CHARACTERISTICS: VS = ±6VBoldface limits are tested at +25°C.At TA = +25°C, G = +4, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only.

NOTES: (1) Junction temperature = ambient for +25°C specifications.(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature

specifications.(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.

(C) Typical value only for information.(4) Current is considered positive out of node. VCM is the input common-mode voltage.(5) Tested < 3dB below minimum CMRR specifications at ±CMIR limits.(6) Peak output duration should not exceed junction temperature +150°C for extended periods.(7) Not connecting the exposed slug to the –VJ plane gives 75°C/W thermal impedance (θJA).


AC PERFORMANCE (see Figure 3)Small-Signal Bandwidth (VO = 0.5VPP) G = +1, RF = 536Ω 160 MHz min B

G = +2, RF = 511Ω 150 120 118 115 MHz min BG = +4, RF = 453Ω 150 130 128 125 MHz min BG = +8, RF = 332Ω 160 130 128 125 MHz min B

Peaking at a Gain of +1 G = +1, RF = 511Ω 0 dB typ CBandwidth for 0.1dB Gain Flatness G = +4, VO = 0.5VPP 70 23 20 19 MHz min BLarge-Signal Bandwidth G = +4, VO = 2VPP 100 MHz typ CSlew Rate G = +4, 2V Step 1100 830 827 825 V/µs min BRise-and-Fall Time G = +4, VO = 2V Step 2 ns typ CHarmonic Distortion G = +4, f = 5MHz, VO = 2VPP

2nd-Harmonic RL = 100Ω –67 –65 –64 –63 dBc max BRL ≥ 500Ω –71 –68 –67 –66 dBc max B

3rd-Harmonic RL = 100Ω –72 –70 –69 –68 dBc max BRL ≥ 500Ω –74 –71 –70 –69 dBc max B

Input Voltage Noise f > 1MHz 2 2.6 2.9 3.1 nV/√Hz max BNoninverting Input Current Noise f > 1MHz 16 20 21 22 pA/√Hz max BInverting Input Current Noise f > 1MHz 24 29 30 31 pA/√Hz max BChannel-to-Channel Crosstalk f = 5MHz, Input Referred –92 dB typ C

DC PERFORMANCEOpen-Loop Transimpedance Gain VO = 0V, RL = 100Ω 110 72 70 68 kΩ min AInput Offset Voltage VCM = 0V ±0.8 ±3.5 ±4.0 ±4.3 mV max AAverage Offset Voltage Drift VCM = 0V ±4 ±10 ±10 ±12 µV/°C max BNoninverting Input Bias Current VCM = 0V ±10 ±30 ±32 ±35 µA max AAverage Noninverting Input Bias Current Drift VCM = 0V ±5 ±50 ±50 ±75 nA/°C max BInverting Input Bias Current VCM = 0V ±10 ±30 ±40 ±45 µA max AAverage Inverting Input Bias Current Drift VCM = 0V ±10 ±100 ±100 ±150 nA°/C max B

INPUTMost Positive Input Voltage 3.7 3.3 3.2 3.1 V min AMost Negative Input Voltage 1.3 1.7 1.8 1.9 V min ACommon-Mode Rejection Ratio (CMRR) VCM = 2.5V, Input Referred 52 49 48 47 dB min ANoninverting Input Impedance 250 || 2 kΩ || pF typ CMinimum Inverting Input Resistance Open-Loop 25 15 kΩ min BMaximum Inverting Input Resistance Open-Loop 25 40 kΩ max B

OUTPUTMost Positive Output Voltage No Load 4.1 3.9 3.8 3.6 V min A

RL = 100Ω 3.5 3.8 3.7 3.5 V min ALeast Positive Output Voltage No Load 0.8 1.0 1.1 1.3 V min A

RL = 100Ω 1.0 1.1 1.2 1.5 V min ACurrent Output VO = 0 ±300 ±200 ±180 ±100 mA min AClosed-Loop Output Impedance G = +4, f ≤ 100kHz 0.02 Ω typ C

POWER SUPPLYSpecified Operating Voltage +5 V typ CMaximum Operating Voltage 12.6 12.6 12.6 V max AMinimum Operating Voltage +4 V typ CMaximum Quiescent Current VS = ±5V, Both Channels 13.6 14.8 15.2 15.6 mA max AMinimum Quiescent Current VS = ±5V, Both Channels 13.6 12.0 11.7 11.4 mA min APower-Supply Rejection Ratio (PSRR) f = 100kHz, Input Referred 52 dB typ C

THERMAL CHARACTERISTICSSpecification: U, DDA, RGV –40 to +85 °CThermal Resistance, θJA Junction-to-Ambient

U SO-8 125 °C/W typ CDDA HSOP-8 Exposed Slug Soldered to Board 55 °C/W typ CRGV QFN-16 Exposed Slug Soldered to Board 50(4) °C/W typ C

OPA2677U, DDA, RGV

TYP

0°C to –40°C to MIN/ TESTPARAMETER CONDITIONS +25°C +25°C(1) 70°C(2) +85°C(2) UNITS MAX LEVEL(3)

ELECTRICAL SPECIFICATIONS: VS = +5VBoldface limits are tested at +25°C.At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted. See Figure 3 for AC performance only.

MIN/MAX OVER TEMPERATURE

NOTES: (1) Junction temperature = ambient for +25°C specifications.(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +9°C at high temperature limit for over temperature

specifications.(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.

(C) Typical value only for information.(4) Not connecting the exposed slug to the –VJ plane gives 75°C/W thermal impedance (θJA).


TYPICAL CHARACTERISTICS: VS = ±6VAt TA = +25°C, G = +4, RF = 402Ω, and RL = 100Ω, unless otherwise noted.

NONINVERTING SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 100 200 300 400 500

6

3

0

–3

–6

–9

–12

–15

–18

Nor

mal

ized

Gai

n (d

B)

VO = 0.5VPP

See Figure 1

G = +8RF = 250Ω

G = +2RF = 475Ω

G = +4RF = 402Ω

G = +1RF = 511Ω

INVERTING SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 100 200 300 400 500

6

3

0

–3

–6

–9

–12

–15

–18

Nor

mal

ized

Gai

n (d

B)

VO = 0.5VPP

G = –1, RF = 475Ω

G = –2, RF = 422Ω

G = –8, RF = 280Ω

G = –4, RF = 383Ω

See Figure 2

NONINVERTING LARGE-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 100 200 300 400 500

18

15

12

9

6

3

0

–3

–6

–9

–12

–15

Gai

n (d

B)

G = +4, See Figure 1

VO = 10VPP

VO = 8VPP

VO = 2VPP

VO ≤ 1VPP

See Figure 1

INVERTING LARGE-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 100 200 300 400 500

18

15

12

9

6

3

0

–3

–6

–9

–12

–15

Gai

n (d

B)

G = –4RF = 383Ω VO = 8VPP

VO = 10VPP

VO = 5VPP

VO ≤ 1VPP

See Figure 2

NONINVERTING PULSE RESPONSE

Time (5ns/div)

Out

put V

olta

ge (

1V/d

iv)

Out

put V

olta

ge (

100m

V/d

iv)

4VPP

G = +4

200mVPP

Left Scale

Large Signal

Right Scale

Small Signal

See Figure 1

INVERTING PULSE RESPONSE

Time (5ns/div)

Out

put V

olta

ge (

1V/d

iv)

Out

put V

olta

ge (

100m

V/d

iv)

4VPPLeft Scale

Large Signal

Right Scale

G = –4

200mVPP

Small Signal

See Figure 2


TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)At TA = +25°C, G = +4, RF = 402Ω, and RL = 100Ω, unless otherwise noted.

HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

0.1 1 2010

–60

–65

–70

–75

–80

–85

–90

–95

–100

Har

mon

ic D

isto

rtio

n (d

Bc)

VO = 2VPPRL = 100Ω

Single Channel—see Figure 1

2nd-Harmonic

3rd-Harmonic

HARMONIC DISTORTION vs OUTPUT VOLTAGE

Output Voltage (VPP)

0.1 1 10

–60

–65

–70

–75

–80

–85

–90

–95

–100

Har

mon

ic D

isto

rtio

n (d

Bc)

F = 5MHzRL = 100Ω 2nd-Harmonic

3rd-Harmonic


HARMONIC DISTORTION vs NONINVERTING GAIN

Gain Magnitude (V/V)

1

–60

–65

–70

–75

–80

–85

–90

–95

–10010

Har

mon

ic D

isto

rtio

n (d

Bc)

VO = 2VPPf = 5MHzRL = 100Ω 2nd-Harmonic

3rd-Harmonic


HARMONIC DISTORTION vs LOAD RESISTANCE

Load Resistance (Ω)

10 100 1000

–60

–65

–70

–75

–80

–85

–90

–95

–100

Har

mon

ic D

isto

rtio

n (d

Bc)


VO = 2VPPf = 5MHz

2nd-Harmonic

3rd-Harmonic

2-TONE, 3rd-ORDERINTERMODULATION SPURIOUS

Single-Tone Load Power (dBm)

–10 0 5–5 10

–60

–65

–70

–75

–80

–85

–90

–95

–100

3rd-

Ord

er S

purio

us L

evel

(dB

c)

See Figure 1 20MHz

5MHz 1MHz


10MHz

HARMONIC DISTORTION vs INVERTING GAIN

Gain Magnitude |(V/V)|

1

–60

–65

–70

–75

–80

–85

–90

–95

–10010

Har

mon

ic D

isto

rtio

n (d

Bc)

VO = 2VPPf = 5MHzRL = 100Ω

2nd-Harmonic

3rd-Harmonic




MAXIMUM OUTPUT SWINGvs LOAD RESISTANCE


10

6

5

4

3

2

1

0

–1

–2

–3

–4

–5

–6100 1000

Out

put V

olta

ge (

V)

See Figure 1

OUTPUT VOLTAGE AND CURRENT LIMITATIONS

IO (mA)

–600

6

5

4

3

2

1

0

–1

–2

–3

–4

–5

–60 200 400–200–400 600

VO (

V)

RL = 10Ω

RL = 25Ω

RL = 50ΩRL = 100Ω

1W Internal PowerSingle Ch

1W Internal PowerSingle Ch

INPUT VOLTAGE AND CURRENT NOISE DENSITY

Frequency (Hz)

100

100

10

1100k 1M10k1k 10M

Vol

tage

Noi

se n

V/√

Hz

Cur

rent

Noi

se p

A/√

Hz

Inverting Current Noise 24pA/√Hz

16pA/√Hz

2nV/√HzVoltage Noise

Noninverting Current Noise

CHANNEL-TO-CHANNEL CROSSTALK

Frequency (Hz)

1M 10M 100M

–60

–65

–70

–75

–80

–85

–90

–95

–100

Cro

ssta

lk, I

nput

Ref

erre

d (d

B)

Input Referred

RECOMMENDED RS vs CAPACITIVE LOAD

Capacitive Load (pF)

1 10 100 1000

90

80

70

60

50

40

30

20

10

0

RS (

Ω)

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (Hz)

1M

2

0

–2

–4

–6

–8

–1010M 100M 1G

Nor

mal

ized

Gai

n to

Cap

aciti

veLo

ad (

dB)

CL = 10pF

CL = 22pFCL = 100pF

CL = 47pF1/2OPA2677

402Ω

RS

133Ω

1kΩCL

1kΩ is optional.



CMRR AND PSRR vs FREQUENCY

Frequency (Hz)

1k

70

60

50

40

30

20

10

010k 100k 1M 10M 100M

Pow

er-S

uppl

y R

ejec

tion

Rat

io (

dB)

Com

mon

-Mod

e R

ejec

tion

Rat

io (

dB)

CMRR

–PSRR

+PSRR

OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE

Frequency (Hz)

10k 100k 1M 10M 100M 1G

120

100

80

60

40

20

0

Tra

nsim

peda

nce

Gai

n (2

0dB

Ω/d

iv)

0

–45

–90

–135

–180

–225

–270

Tra

nsim

peda

nce

Pha

se (

45°/

div)

CLOSED-LOOP OUTPUT IMPEDANCEvs FREQUENCY

Frequency (Hz)

10k 100k 1M 10M 100M 1G

100

10

1

0.1

0.01

0.001

Out

put I

mpe

danc

e M

agni

tude

(Ω

)

COMPOSITE VIDEO dG/dφ

Number of 150Ω Loads

1 2 3 4 5 6 7 8 9 10

0.14

0.12

0.10

0.08

0.06

0.04

0.02

0

dG/d

φ (%

/°)

G = +2RF = 475ΩVS = ±5V dφ, Negative Video

dφ, Positive Video

dG, Positive Video

dG, Negative Video

8

6

4

2

0

–2

–4

–6

–8

NONINVERTING OVERDRIVE RECOVERY

Time (20ns/div)

Out

put V

olta

ge (

2V/d

iv)

4

3

2

1

0

–1

–2

–3

–4

Inpu

t Vol

tage

(1V

/div

)

G = +4RL = 100ΩSee Figure 1

Input

Output

8

6

4

2

0

–2

–4

–6

–8

INVERTING OVERDRIVE RECOVERY

Time (20ns/div)

Out

put V

olta

ge (

2V/d

iv)

4

3

2

1

0

–1

–2

–3

–4

Inpu

t Vol

tage

(1V

/div

)Input

Output See Figure 2

G = –4RL = 100Ω



TYPICAL DC ERROR DRIFTvs TEMPERATURE

Ambient Temperature (°C)

–55

10

8

6

4

2

0

–2

–4

–6

–8

–10–35 –15 5 25 45 65 85 105 125

Inpu

t Offs

et V

olta

ge (

mV

)In

put B

ias

Cur

rent

(µA

)

Noninverting Bias Current

Input Offset Voltage

Inverting Bias Current

SUPPLY AND OUTPUT CURRENTvs TEMPERATURE

Temperature (°C)

–55

600

550

500

450

400

350

300

250

200

150

100–35 –15 5 25 45 65 85 105 125

Out

put C

urre

nt (

mA

)

50

40

30

20

10

0

Out

put C

urre

nt (

mA

)

Sourcing Output Current

Sinking Output Current

Supply Current

CMIR AND OUTPUT VOLTAGEvs SUPPLY VOLTAGE

Supply Voltage (±V)

2 3 4 5

6

5

4

3

2

1

06

Vol

tage

Ran

ge (

±V) ± Output Voltage

No Load

+V Input Voltage

–V Input Voltage


TYPICAL CHARACTERISTICS: VS = ±6V (Cont.)At TA = +25°C, Differential Gain = +9, RF = 300Ω, and RL = 70Ω, unless otherwise noted. See Figure 5 for AC performance only.

DIFFERENTIAL SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

5 10010 500

3

0

–3

–6

–9

Nor

mal

ized

Gai

n (d

Bc)

RL = 70ΩVO = 200mVPP

GD = +2,RF = 442Ω

GD = +5,RF = 383Ω

GD = +9,RF = 300Ω

See Figure 5

DIFFERENTIAL LARGE-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

4 10010 400

20

19

18

17

16

15

14

Gai

n (d

B)

0.2VPPRL = 70ΩGD = +9

5VPP

2VPP

1VPP

See Figure 5



10 1k100

–65

–70

–75

–80

–85

–90

–95

Har

mon

ic D

isto

rtio

n (d

B) 2nd-Harmonic

3rd-Harmonic

f = 5MHzGD = +9VO = 2VPP

See Figure 5


Frequency (MHz)

0.1 101 100

–60

–65

–70

–75

–80

–85

–90

–95

–100

Har

mon

ic D

isto

rtio

n (d

B) 2nd-Harmonic

3rd-Harmonic

GD = +9

RL = 70ΩVO = 2VPP

See Figure 5


Output Voltage (Vp-p)

0.1 101 100

–64

–66

–68

–70

–72

–74

–76

–78

–80

Har

mon

ic D

isto

rtio

n (d

B)

f = 5MHzG = +9RL = 70Ω

2nd-Harmonic

3rd-Harmonic

See Figure 5

MULTITONE POWER RATIO(VS = ±6V, 13dBm Output Power)

Frequency (kHz)

0 20 40 60 80 100 120 140 160

10

0

–10

–20

–30

–40

–50

–60

–70

–80

Pow

er (

dB)

See Figure 5


TYPICAL CHARACTERISTICS: VS = +5VAt TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω to VS/2, unless otherwise noted.

NONINVERTING SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 50 100 150 200 250

6

3

0

–3

–6

–9

–12

–15

–18

Nor

mal

ized

Gai

n (d

B)

See Figure 3

G = +1RF = 536Ω

G = +2RF = 511Ω

G = +4RF = 453Ω

G = +8RF = 332Ω

INVERTING SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

0 50 100 150 200 250

6

3

0

–3

–6

–9

–12

–15

–18

Nor

mal

ized

Gai

n (d

B)

G = –8RF = 332Ω

G = –4RF = 453Ω

G = –2RF = 511Ω

G = –1RF = 536Ω

See Figure 4

400

300

200

100

0

–100

–200

–300

–400

SMALL-SIGNAL PULSE RESPONSE

Time (5ns/div)

Out

put V

olta

ge (

100m

V/d

iv)

VO = 500mVPP

See Figure 3

1.6

1.2

0.8

0.4

0

–0.4

–0.8

–1.2

–1.6

LARGE-SIGNAL PULSE RESPONSE

Time (5ns/div)

Out

put V

olta

ge (

400m

V/d

iv)

VO = 2VPP

See Figure 3

RECOMMENDED RS vs CAPACITIVE LOAD

Capacitive Load (pF)

1 10 100 1000

50

45

40

35

30

25

20

15

10

5

0

RS (

Ω)

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (Hz)

1M

2

0

–2

–4

–6

–8

–1010M 100M 1G

Nor

mal

ized

Gai

n to

Cap

aciti

veLo

ad (

dB)

CL = 10pF

CL = 22pF

CL = 47pF453Ω

150Ω

804Ω

804Ω

1kΩ

1kΩ Load Optional.

1/2OPA2677

VI

+5V

0.1µF

VORS

CL

0.1µF

CL = 100pF


TYPICAL CHARACTERISTICS: VS = +5V (Cont.)At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω to VS/2, unless otherwise noted.


Frequency (MHz)

0.1 1 2010

–50

–55

–60

–65

–70

–75

–80

–85

–90

Har

mon

ic D

isto

rtio

n (d

Bc)

VO = 2VPPRL = 100Ω to VS/2


2nd-Harmonic

3rd-Harmonic

HARMONIC DISTORTION vs NONINVERTING GAIN

Gain Magnitude (V/V)

1

–50

–55

–60

–65

–70

–75

–80

–85

–9010

Har

mon

ic D

isto

rtio

n (d

Bc)

2nd-Harmonic

3rd-Harmonic

VO = 2VPPf = 5MHzRL = 100Ω to VS/2



Output Voltage (VPP)

0.1 1 2

–50

–55

–60

–65

–70

–75

–80

–85

–90

Har

mon

ic D

isto

rtio

n (d

Bc)

f = 5MHzRL = 100Ω to VS/2


3rd-Harmonic

2nd-Harmonic

HARMONIC DISTORTION vs INVERTING GAIN

Gain (V/V)

–1

–50

–55

–60

–65

–70

–75

–80

–85

–90–10

Har

mon

ic D

isto

rtio

n (d

Bc)

2nd-Harmonic

3rd-Harmonic

VO = 2VPPf = 5MHzRL = 100Ω to VS/2


2-TONE, 3rd-ORDER SPURIOUS LEVEL

Single-Tone Load Power (dBm)

–10 0 5–5 10

–50

–55

–60

–65

–70

–75

–80

–85

–90

3rd-

Ord

er S

purio

us L

evel

(dB

c)

20MHz

5MHz

1MHz


10MHz



10 100 1000

–50

–55

–60

–65

–70

–75

–80

–85

–90

Har

mon

ic D

isto

rtio

n (d

Bc)


VO = 2VPPf = 5MHz

2nd-Harmonic

3rd-Harmonic


TYPICAL CHARACTERISTICS: VS = +5V (Cont.)At TA = +25°C, Differential Gain = +9, RF = 316Ω, and RL = 70Ω, unless otherwise noted.

RLRG

CG

VOVI

GD = 1 + 2 • RF

RG

RF300Ω

RF300Ω

1/2OPA2677

1/2OPA2677

=VO

VI

DIFFERENTIAL SMALL-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

10 100 200

3

0

–3

–6

–9

–12

Nor

mal

ized

Gai

n (d

B)

RL = 70Ω

GD = +2RF = 511Ω

GD = +5RF = 422Ω

GD = +9RF = 316Ω

DIFFERENTIAL LARGE-SIGNALFREQUENCY RESPONSE

Frequency (MHz)

10 100

20

19

18

17

16

15

14

Gai

n (d

B)

RL = 70ΩGD = +9

0.2VPP

1VPP

2VPP

5VPP


Frequency (MHz)

0.1 101 100

–60

–65

–70

–75

–80

–85

Har

mon

ic D

isto

rtio

n (d

Bc)

3rd-Harmonic

2nd-Harmonic

GD = +9RL = 70ΩVO = 2VPP



10 1k100

–65

–70

–75

–80

–85

–90

–95

Har

mon

ic D

isto

rtio

n (d

Bc)

2nd-Harmonic

3rd-Harmonic

GD = +9f = 5MHzVO = 2VPP


Differential Output Voltage (VPP)

0.1 101

–65

–70

–75

–80

–85

Har

mon

ic D

isto

rtio

n (d

B)

f = 5MHzG = +9RL = 70Ω 3rd-Harmonic

2nd-Harmonic

DIFFERENTIAL PERFORMANCETEST CIRCUIT


APPLICATION INFORMATIONWIDEBAND CURRENT-FEEDBACK OPERATION

The OPA2677 gives the exceptional AC performance of awideband current-feedback op amp with a highly linear, high-power output stage. Requiring only 9mA/ch quiescent cur-rent, the OPA2677 swings to within 1V of either supply railand delivers in excess of 380mA at room temperature. Thislow-output headroom requirement, along with supply voltageindependent biasing, gives remarkable single (+5V) supplyoperation. The OPA2677 delivers greater than 150MHz band-width driving a 2VPP output into 100Ω on a single +5V supply.Previous boosted output stage amplifiers typically suffer fromvery poor crossover distortion as the output current goesthrough zero. The OPA2677 achieves a comparable powergain with much better linearity. The primary advantage of acurrent-feedback op amp over a voltage-feedback op amp isthat AC performance (bandwidth and distortion) is relativelyindependent of signal gain. Figure 1 shows the DC-coupled,gain of +4, dual power-supply circuit configuration used asthe basis of the ±6V Electrical and Typical Characteristics.For test purposes, the input impedance is set to 50Ω with aresistor to ground and the output impedance is set to 50Ωwith a series output resistor. Voltage swings reported in theelectrical characteristics are taken directly at the input andoutput pins, whereas load powers (dBm) are defined at amatched 50Ω load. For the circuit of Figure 1, the totaleffective load is 100Ω || 535Ω = 84Ω.

Figure 2 shows the DC-coupled, bipolar supply circuit con-figuration used as the basis for the Inverting Gain ±6VTypical Characteristics. Key design considerations of theinverting configuration are developed in the Inverting Ampli-fier Operation section.

Figure 3 shows the AC-coupled, gain of +4, single-supplycircuit configuration used as the basis of the +5V Electricaland Typical Characteristics. Though not a rail-to-rail design,the OPA2677 requires minimal input and output voltageheadroom compared to other very wideband current-feed-back op amps. It will deliver a 3VPP output swing on a single+5V supply with greater than 100MHz bandwidth. The keyrequirement of broadband single-supply operation is to main-tain input and output signal swings within the usable voltageranges at both the input and the output. The circuit of Figure 3establishes an input midpoint bias using a simple resistivedivider from the +5V supply (two 806Ω resistors). The inputsignal is then AC-coupled into this midpoint voltage bias. Theinput voltage can swing to within 1.3V of either supply pin,giving a 2.4VPP input signal range centered between thesupply pins. The input impedance matching resistor (57.6Ω)used for testing is adjusted to give a 50Ω input match whenthe parallel combination of the biasing divider network isincluded. The gain resistor (RG) is AC-coupled, giving thecircuit a DC gain of +1—which puts the input DC bias voltage(2.5V) on the output as well. The feedback resistor value isadjusted from the bipolar supply condition to re-optimize fora flat frequency response in +5V, gain of +4, operation.Again, on a single +5V supply, the output voltage can swingto within 1V of either supply pin while delivering more than200mA output current. A demanding 100Ω load to a midpointbias is used in this characterization circuit. The new outputstage used in the OPA2677 can deliver large bipolar outputcurrents into this midpoint load with minimal crossover distor-tion, as shown by the +5V supply, harmonic distortion plots.

1/2OPA2677

+6V

+

–6V

50Ω Load

50Ω50ΩVOVI

50Ω Source

RG133Ω

RF402Ω

+6.8µF

0.1µF 6.8µF

0.1µF

+VS

–VS

1/2OPA2677

+5V

–5V

50Ω Load50ΩVO

VI

50ΩSource

RM100Ω

RF402Ω

RF402Ω

Power-supplydecouplingnot shown.

FIGURE 1. DC-Coupled, G = +4, Bipolar Supply, Specifica-tion and Test Circuit.

FIGURE 2. DC-Coupled, G = –4, Bipolar Supply, Specifica-tion and Test Circuit.


The last configuration used as the basis of the +5V Electricaland Typical Characteristics is shown in Figure 4. Designconsiderations for this inverting, bipolar supply configurationare covered either in single-supply configuration (as shownin Figure 3) or in the Inverting Amplifier Operation section.

where the input is brought into the OPA2677. Each has itsadvantages and disadvantages. Figure 5 shows a basicstarting point for noninverting differential I/O applications.

DIFFERENTIAL INTERFACE APPLICATIONS

Dual op amps are particularly suitable to differential input todifferential output applications. Typically, these fall into eitherAnalog-to-Digital Converter (ADC) input interface or linedriver applications. Two basic approaches to differential I/Oare noninverting or inverting configurations. Since the outputis differential, the signal polarity is somewhat meaningless—the noninverting and inverting terminology applies here to

1/2OPA2677

+5V

VS/2806Ω

VI

100ΩVO

806Ω

RF453Ω

RM88.7Ω

6.8µF+

0.1µF

0.1µFRG

113Ω

This approach provides for a source termination impedancethat is independent of the signal gain. For instance, simpledifferential filters may be included in the signal path right upto the noninverting inputs without interacting with the gainsetting. The differential signal gain for the circuit of Figure 5is:

AD = 1 + 2 • RF/RG

Since the OPA2677 is a current feedback (CFB) amplifier, itsbandwidth is principally controlled with the feedback resistorvalue; Figure 5 shows a value of 300Ω for the AD = +9design. The differential gain, however, may be adjusted withconsiderable freedom using just the RG resistor. In fact, RG

may be a reactive network providing a very isolated shapingto the differential frequency response.

Various combinations of single-supply or AC-coupled gaincan also be delivered using the basic circuit of Figure 5.Common-mode bias voltages on the two noninverting inputspass on to the output with a gain of 1 since an equal DCvoltage at each inverting node creates no current throughRG. This circuit does show a common-mode gain of 1 frominput to output. The source connection should either removethis common-mode signal if undesired (using an input trans-former can provide this function), or the common-modevoltage at the inputs can be used to set the output common-mode bias. If the low common-mode rejection of this circuitis a problem, the output interface may also be used to rejectthat common-mode. For instance, most modern differentialinput ADCs reject common-mode signals very well, while aline driver application through a transformer will also attenu-ate the common-mode signal through to the line.

RL

RF300Ω

RF300Ω

–6

+6

1/2OPA2677

1/2OPA2677

RG75Ω

CG

VOVI

GD = 1 + =2 • RF

RG

VO

VI

FIGURE 3. AC-Coupled, G = +4, Single-Supply, Specifica-tion and Test Circuit.

FIGURE 4. AC-Coupled, G = –4, Single-Supply, Specifica-tion and Test Circuit.

FIGURE 5. Noninverting Differential I/O Amplifier.

1/2OPA2677

+5V+VS

VS/2806Ω

100ΩVOVI 57.6Ω

806Ω

RF453Ω

RG150Ω

0.1µF

0.1µF

6.8µF+

0.1µF


SINGLE-SUPPLY ADSL UPSTREAM DRIVER

Figure 6 shows an example of a single-supply ADSL up-stream driver. The dual OPA2677 is configured as a differen-tial gain stage to provide signal drive to the primary windingof the transformer (here, a step-up transformer with a turnsratio of 1:1.7). The main advantage of this configuration is thecancellation of all even harmonic distortion products. Anotherimportant advantage for ADSL is that each amplifier needsonly to swing half of the total output required driving the load.

OPA2677 HDSL2 UPSTREAM DRIVER

Figure 7 shows an HDSL2 implementation of a single-supplyupstream driver.

The two designs differ by the values of their matchingimpedance, the load impedance, and the ratio turns of thetransformers. All these differences are reflected in the higherpeak current and thus, the higher maximum power dissipa-tion in the output of the driver.

The analog front end (AFE) signal is AC-coupled to thedriver, and the noninverting input of each amplifier is biasedto the mid-supply voltage (+6V in this case). In addition toproviding the proper biasing to the amplifier, this approachalso provides a high-pass filtering with a corner frequency,set here at 5kHz. As the upstream signal bandwidth starts at26kHz, this high-pass filter does not generate any problemand has the advantage of filtering out unwanted lower fre-quencies.

The input signal is amplified with a gain set by the followingequation:

GR

RDF

G= + •

12

(1)

With RF = 324Ω and RG = 82.5Ω, the gain for this differentialamplifier is 8.85. This gain boosts the AFE signal, assumedto be a maximum of 2VPP, to a maximum of 17.3VPP.

Refer to the Setting Resistor Values to Optimize Bandwidthsection for a discussion on which feedback resistor value tochoose.

The two back-termination resistors (17.4Ω each) added ateach terminal of the transformer make the impedance of themodem match the impedance of the phone line, and alsoprovide a means of detecting the received signal for thereceiver. The value of these resistors (RM) is a function of theline impedance and the transformer turns ratio (n), given bythe following equation:

RZ

MLINE

n=

2 2 (2)

RG82.5Ω

2kΩ

2kΩ

1µF

0.1µF

0.1µFRM

17.4Ω

100Ω

ZLINEAFE2VPPMax

Assumed

RF324Ω

20Ω

20Ω

324Ω

RF

1/2OPA2677

1/2OPA2677

+12V

1:1.7

17.7VPP

IP = 128mA

IP = 128mA

RM17.4Ω

+6V82.5Ω

2kΩ

2kΩ

1µF

0.1µF

0.1µFRM

11.5Ω135Ω

ZLINEAFE2VPPMax

Assumed

324Ω

20Ω

20Ω

324Ω

1/2OPA2677

1/2OPA2677

+12V

1:2.4

17.3VPP

IP = 185mA

IP = 185mA

RM11.5Ω

+6V

LINE DRIVER HEADROOM MODEL

The first step in a transformer-coupled, twisted-pair driverdesign is to compute the peak-to-peak output voltage fromthe target specifications. This is done using the followingequations:

PV

mW RLRMS

L= • ( ) •

101

2log (3)

with PL power at the load, VRMS voltage at the load, and RL

load impedance; this gives the following:

V mW RRMS L

PL

= ( ) • •1 10 10 (4)

V Crest Factor V C VP RMS F RMS= =• • (5)

with VP peak voltage at the load and CF Crest Factor.

VLPP = 2 • CF • VRMS (6)

with VLPP: peak-to-peak voltage at the load.

Consolidating Equations 3 through 6 allows expressing therequired peak-to-peak voltage at the load as a function of thecrest factor, the load impedance, and the power at the load.

Thus, V CF mW RLPP L

PL

= • • ( ) • •2 1 10 10 (7)

This VLPP is usually computed for a nominal line impedanceand may be taken as a fixed design target.

The next step in the design is to compute the individualamplifier output voltage and currents as a function of VPP on

FIGURE 6. Single-Supply ADSL Upstream Driver. FIGURE 7. HDSL2 Upstream Driver.


the line and transformer turns ratio. As this turns ratiochanges, the minimum allowed supply voltage changes alongwith it. The peak current in the amplifier output is given by:

± = • • •IVn RP

LPP

M

12

2 14 (8)

with VPP as defined in Equation 7, and RM as defined inEquation 2 and shown in Figure 8.

TOTAL DRIVER POWER FOR xDSL APPLICATIONS

The total internal power dissipation for the OPA2677 in anxDSL line driver application will be the sum of the quiescentpower and the output stage power. The OPA2677 holds arelatively constant quiescent current versus supply voltage—giving a power contribution that is simply the quiescentcurrent times the supply voltage used (the supply voltage willbe greater than the solution given in Equation 10). The totaloutput stage power may be computed with reference toFigure 10.

With the previous information available, it is now possible toselect a supply voltage and the turns ratio desired for thetransformer as well as calculate the headroom for theOPA2677.

The model, shown in Figure 9, can be described with thefollowing set of equations:

1) As available output swing:

VPP = VCC – (V1 + V2) – IP • (R1 + R2) (9)

2) Or as required supply voltage:

VCC = VPP + (V1 + V2) + IP • (R1 + R2) (10)

The minimum supply voltage for a power and load require-ment is given by Equation 10.

RM

RM

VLppn VLppRL

2VLppnVpp =

VO

R1

V1

+VCC

R2

V2

IP

FIGURE 8. Driver Peak Output Voltage.

FIGURE 9. Line Driver Headroom Model.

V1 R1 V2 R2

+5V 0.9V 5Ω 0.8V 5Ω+12V 0.9V 2Ω 0.9V 2Ω

TABLE I. Line Driver Headroom Model Values.

V1, V2, R1, and R2 are given in Table I for both +12V and +5Voperation.

FIGURE 10. Output Stage Power Model.

The two output stages used to drive the load of Figure 8 canbe seen as an H-Bridge in Figure 10. The average currentdrawn from the supply into this H-Bridge and load will be thepeak current in the load given by Equation 8 divided by thecrest factor (CF) for the xDSL modulation. This total powerfrom the supply is then reduced by the power in RT to leavethe power dissipated internal to the drivers in the four outputstage transistors. That power is simply the target line powerused in Equation 3 plus the power lost in the matchingelements (RM). In the examples here, a perfect match istargeted giving the same power in the matching elements asin the load. The output stage power is then set by Equation 11.

PICF

V POUTP

CC L= × – 2 (11)

The total amplifier power is then:

P I VICF

V PTOT q CCP

CC L= × + × – 2 (12)

For the ADSL CPE upstream driver design of Figure 6, thepeak current is 128mA for a signal that requires a crest factorof 5.33 with a target line power of 13dBm into 100Ω (20mW).With a typical quiescent current of 18mA and a nominalsupply voltage of +12V, the total internal power dissipationfor the solution of Figure 6 will be:

(13)

P mA VmA

V mW mWTOT = ( ) + ( ) ( ) =18 12128

5 3312 2 20 464

.–

RT

+VCCIAVG =

IPCF


DESIGN-IN TOOLSDEMONSTRATION FIXTURES

A printed circuit board (PCB) is available to assist in the initialevaluation of circuit performance using the OPA2677. Thefixture is offered free of charge as unpopulated PCB, deliv-ered with a user’s guide. The summary information for thisfixture is shown in Table II.

The buffer gain is typically very close to 1.00 and is normallyneglected from signal gain considerations. It sets the CMRR,however, for a single op amp differential amplifier configura-tion. For a buffer gain α < 1.0, the CMRR = –20 • log (1 – α)dB.

RI, the buffer output impedance, is a critical portion of thebandwidth control equation. The OPA2677 inverting inputresistor is typically 22Ω.

A current-feedback op amp senses an error current in theinverting node (as opposed to a differential input error volt-age for a voltage-feedback op amp) and passes this on to theoutput through an internal frequency dependent transimped-ance gain. The Typical Characteristics show this open-looptransimpedance response, which is analogous to the open-loop voltage gain curve for a voltage-feedback op amp.Developing the transfer function for the circuit of Figure 11gives Equation 14:

VV

RR

R RRR

Z s

NGR R NG

Z s

NGRR

O

I

F

G

F IF

G

F I

F

G

=+

++ +

=+ +

= +

•α

α1

1

1 1

1

( )

( )

(14)

This is written in a loop-gain analysis format where the errorsarising from a non-infinite open-loop gain are shown in thedenominator. If Z(s) is infinite over all frequencies, the denomi-nator of Equation 14 reduces to 1 and the ideal desired signalgain shown in the numerator is achieved. The fraction in thedenominator of Equation 14 determines the frequency re-sponse. Equation 15 shows this as the loop-gain equation:

Z sR R NG

Loop GainF I

( )+

= (15)

If 20log(RF + NG • RI) is drawn on top of the open-looptransimpedance plot, the difference between the two wouldbe the loop gain at a given frequency. Eventually, Z(s) rolls offto equal the denominator of Equation 15, at which point theloop gain has reduced to 1 (and the curves have intersected).This point of equality is where the amplifier closed-loop

VO

RG

VI

RI

Z(S) IERR

α

RF

IERR

FIGURE 11. Current Feedback Transfer Function AnalysisCircuit.TABLE II. Demonstration Fixtures by Package.

ORDERING LITERATUREPRODUCT PACKAGE NUMBER NUMBER

OPA2677U SO-8 DEM-OPA-SO-2A SBOU003OPA2677IDDA HSOP-8 Not Available Not Available

OPA2677T SO-16 Not Available Not AvailableOPA2677IRGV QFN-16 Not Available Not Available

This demonstration fixture can be requested at the TexasInstruments web site (www.ti.com) through the OPA2677product folder.

MACROMODELS AND APPLICATIONS SUPPORT

Computer simulation of circuit performance using SPICE isoften useful when analyzing the performance of analogcircuits and systems. This is particularly true for video and RFamplifier circuits where parasitic capacitance and inductancecan have a major effect on circuit performance. A SPICEmodel for the OPA2677 is available through the TI web site(www.ti.com). This model does a good job of predictingsmall-signal AC and transient performance under a widevariety of operating conditions, but does not do as well inpredicting the harmonic distortion or dG/dP characteristics.This model does not attempt to distinguish between thepackage types in small-signal AC performance, nor does itattempt to simulate channel-to-channel coupling.

OPERATING SUGGESTIONSSETTING RESISTOR VALUES TOOPTIMIZE BANDWIDTH

A current-feedback op amp like the OPA2677 can hold analmost constant bandwidth over signal gain settings with theproper adjustment of the external resistor values, which isshown in the Typical Characteristics; the small-signal band-width decreases only slightly with increasing gain. Thesecurves also show that the feedback resistor is changed foreach gain setting. The resistor values on the inverting side ofthe circuit for a current-feedback op amp can be treated asfrequency response compensation elements, whereas theirratios set the signal gain. Figure 11 shows the small-signalfrequency response analysis circuit for the OPA2677.

The key elements of this current-feedback op amp model are:

α → Buffer gain from the noninverting input to the inverting input

RI → Buffer output impedance

iERR → Feedback error current signal

Z(S) → Frequency dependent open-loop transimpedancegain from iERR to VO


frequency response given by Equation 14 starts to roll off, andis exactly analogous to the frequency at which the noise gainequals the open-loop voltage gain for a voltage-feedback opamp. The difference here is that the total impedance in thedenominator of Equation 15 may be controlled somewhatseparately from the desired signal gain (or NG). The OPA2677is internally compensated to give a maximally flat frequencyresponse for RF = 402Ω at NG = 4 on ±6V supplies. Evaluat-ing the denominator of Equation 15 (which is the feedbacktransimpedance) gives an optimal target of 490Ω. As thesignal gain changes, the contribution of the NG • RI term inthe feedback transimpedance changes, but the total can beheld constant by adjusting RF. Equation 16 gives an approxi-mate equation for optimum RF over signal gain:

RF = 490 – NG • RI (16)

As the desired signal gain increases, this equation eventuallypredicts a negative RF. A somewhat subjective limit to thisadjustment can also be set by holding RG to a minimum valueof 20Ω. Lower values load both the buffer stage at the inputand the output stage if RF gets too low—actually decreasingthe bandwidth. Figure 12 shows the recommended RF versusNG for both ±6V and a single +5V operation. The values forRF versus gain shown here are approximately equal to thevalues used to generate the Typical Characteristics. Theydiffer in that the optimized values used in the Typical Char-acteristics are also correcting for board parasitic not consid-ered in the simplified analysis leading to Equation 16. Thevalues shown in Figure 12 give a good starting point fordesigns where bandwidth optimization is desired.

The total impedance going into the inverting input may be

1/2OPA2677

RF392Ω

VO

VI

RG97.6Ω

+6V

–6V

50Ω50Ω Load

VO

Power-supply decoupling notshown.

VI

50ΩSource

RM102Ω

RF

RG= – = –4

FIGURE 13. Inverting Gain of –4 with Impedance Matching.600

500

400

300

200

Noise Gain

0 2510 15 205

Fee

dbac

k R

esis

tor

(Ω)

+5V

±5V

FIGURE 12. Feedback Resistor vs. Noise Gain.

is essential for power-supply ripple rejection, noninvertinginput noise current shunting, and to minimize the high-frequency value for RI in Figure 11.

INVERTING AMPLIFIER OPERATION

The OPA2677 is a general-purpose, wideband current-feed-back op amp; most of the familiar op amp application circuitsshould be available to the designer. Those dual op ampapplications that require considerable flexibility in the feed-back element (for example, integrators, transimpedance, andsome filters) should consider a unity-gain stable, voltage-feedback amplifier such as the OPA2822, because the feed-back resistor is the compensation element for a current-feedback op amp. Wideband inverting operation (and espe-cially summing) is particularly suited to the OPA2677. Figure13 shows a typical inverting configuration where the I/Oimpedances and signal gain from Figure 1 are retained in aninverting circuit configuration.

In the inverting configuration, two key design considerations

used to adjust the closed-loop signal bandwidth. Inserting aseries resistor between the inverting input and the summingjunction increases the feedback impedance (denominator ofEquation 15), decreasing the bandwidth. The internal bufferoutput impedance for the OPA2677 is slightly influenced bythe source impedance looking out of the noninverting inputterminal. High-source resistors have the effect of increasingRI, decreasing the bandwidth. For those single-supply appli-cations which develop a midpoint bias at the noninvertinginput through high-valued resistors, the decoupling capacitor

must be noted. The first is that the gain resistor (RG)becomes part of the signal source input impedance. If inputimpedance matching is desired (which is beneficial when-ever the signal is coupled through a cable, twisted pair, longPC board trace or other transmission line conductor), it isnormally necessary to add an additional matching resistor toground. RG, by itself, is not normally set to the required inputimpedance since its value, along with the desired gain, willdetermine an RF, which may be non-optimal from a fre-quency response standpoint. The total input impedance forthe source becomes the parallel combination of RG and RM.

The second major consideration, touched on in the previousparagraph, is that the signal source impedance becomespart of the noise gain equation and has a slight effect on thebandwidth through Equation 15. The values shown in Figure12 have accounted for this by slightly decreasing RF (fromthe optimum values) to re-optimize the bandwidth for thenoise gain of Figure 12 (NG = 3.98). In the example of Figure13, the RM value combines in parallel with the external 50Ωsource impedance, yielding an effective driving impedance of50Ω || 102Ω = 33.5Ω. This impedance is added in series withRG for calculating the noise gain—which gives NG = 3.98.This value, along with the inverting input impedance of 22Ω,are inserted into Equation 16 to get a feedback


transimpedance nearly equal to the 402Ω optimum value.Note that the noninverting input in this bipolar supply invert-ing application is connected directly to ground. It is oftensuggested that an additional resistor be connected to groundon the noninverting input to achieve bias current error can-cellation at the output. The input bias currents for a current-feedback op amp are not generally matched in either magni-tude or polarity. Connecting a resistor to ground on thenoninverting input of the OPA2677 in the circuit of Figure 13actually provides additional gain for that input bias and noisecurrents, but does not decrease the output DC error since theinput bias currents are not matched.

OUTPUT CURRENT AND VOLTAGE

The OPA2677 provides output voltage and current capabili-ties that are unsurpassed in a low-cost dual monolithic opamp. Under no-load conditions at 25°C, the output voltagetypically swings closer than 1V to either supply rail; tested at+25°C swing limit is within 1.1V of either rail. Into a 6Ω load(the minimum tested load), it delivers more than ±380mAcontinuous and > ±1.2A peak output current.

The specifications described above, though familiar in theindustry, consider voltage and current limits separately. In manyapplications, it is the voltage times current (or V-I product) thatis more relevant to circuit operation. Refer to the Output Voltageand Current Limitations plot in the Typical Characteristics. TheX and Y axes of this graph show the zero-voltage output currentlimit and the zero-current output voltage limit, respectively. Thefour quadrants give a more detailed view of the OPA2677output drive capabilities, noting that the graph is bounded by asafe operating area of 1W maximum internal power dissipation(in this case for 1 channel only). Superimposing resistor loadlines onto the plot shows that the OPA2677 can drive ±4V into10Ω or ±4.5V into 25Ω without exceeding the output capabilitiesor the 1W dissipation limit. A 100Ω load line (the standard testcircuit load) shows the full ±5.0V output swing capability, asshown in the Electrical Characteristics tables. The minimumspecified output voltage and current over temperature are set byworst-case simulations at the cold temperature extreme. Only atcold startup will the output current and voltage decrease to thenumbers shown in the Electrical Characteristics tables. As theoutput transistors deliver power, the junction temperaturesincreases, decreasing the VBEs (increasing the available outputvoltage swing), and increasing the current gains (increasing theavailable output current). In steady-state operation, the avail-able output voltage and current will always be greater than thatshown in the over-temperature specifications, since the outputstage junction temperatures will be higher than the minimumspecified operating ambient. To maintain maximum outputstage linearity, no output short-circuit protection is provided.This is normally not a problem because most applicationsinclude a series-matching resistor at the output that limits theinternal power dissipation if the output side of this resistor isshorted to ground. However, shorting the output pin directly tothe adjacent positive power-supply pin (8-pin package), will inmost cases, destroy the amplifier. If additional short-circuitprotection is required, consider using the equivalent OPA2674that includes output current limiting. Alternatively, a small series

resistor may be included in the supply lines. Under heavy outputloads this will reduce the available output voltage swing. A 5Ωseries resistor in each power-supply lead will limit the internalpower dissipation to less than 1W for an output short circuitwhile decreasing the available output voltage swing only 0.5Vfor up to 100mA desired load currents. Always place the 0.1µFpower-supply decoupling capacitors after these supply currentlimiting resistors directly on the supply pins.

DRIVING CAPACITIVE LOADS

One of the most demanding and yet very common loadconditions for an op amp is capacitive loading. Often, thecapacitive load is the input of an analog-to-digital (A/D)converter—including additional external capacitance that maybe recommended to improve the A/D converter linearity. Ahigh-speed, high open-loop gain amplifier such as theOPA2677 can be very susceptible to decreased stability andclosed-loop response peaking when a capacitive load isplaced directly on the output pin. When the amplifier open-loop output resistance is considered, this capacitive loadintroduces an additional pole in the signal path that candecrease the phase margin. Several external solutions to thisproblem have been suggested.

When the primary considerations are frequency response flat-ness, pulse response fidelity, and/or distortion, the simplest andmost effective solution is to isolate the capacitive load from thefeedback loop by inserting a series isolation resistor betweenthe amplifier output and the capacitive load. This does noteliminate the pole from the loop response, but rather shifts it andadds a zero at a higher frequency. The additional zero acts tocancel the phase lag from the capacitive load pole, thusincreasing the phase margin and improving stability. The Typi-cal Characteristics show the recommended RS vs CapacitiveLoad and the resulting frequency response at the load. Parasiticcapacitive loads greater than 2pF can begin to degrade theperformance of the OPA2677. Long PC board traces, un-matched cables, and connections to multiple devices can easilycause this value to be exceeded. Always consider this effectcarefully, and add the recommended series resistor as close aspossible to the OPA2677 output pin (see the Board LayoutGuidelines section).

DISTORTION PERFORMANCE

The OPA2677 provides good distortion performance into a100Ω load on ±6V supplies. Relative to alternative solutions,it provides exceptional performance into lighter loads and/oroperation on a single +5V supply. Generally, until the funda-mental signal reaches very high frequency or power levels,the 2nd-harmonic dominates the distortion with a negligible3rd-harmonic component. Focusing then on the 2nd-har-monic, increasing the load impedance improves distortiondirectly. Remember that the total load includes the feedbacknetwork—in the noninverting configuration (see Figure 1),this is the sum of RF + RG, whereas in the inverting configu-ration it is just RF. Also, providing an additional supplydecoupling capacitor (0.01µF) between the supply pins (forbipolar operation) improves the 2nd-order distortion slightly(3dB to 6dB).


In most op amps, increasing the output voltage swing in-creases harmonic distortion directly. The Typical Character-istics show the 2nd-harmonic increasing at a little less thanthe expected 2x rate whereas the 3rd-harmonic increases ata little less than the expected 3x rate. Where the test powerdoubles, the difference between it and the 2nd-harmonicdecreases less than the expected 6dB, whereas the differ-ence between it and the 3rd-harmonic decreases by lessthan the expected 12dB. This also shows up in the 2-tone,3rd-order intermodulation spurious (IM3) response curves.The 3rd-order spurious levels are extremely low at low-outputpower levels. The output stage continues to hold them loweven as the fundamental power reaches very high levels. Asthe Typical Characteristics show, the spurious intermodulationpowers do not increase as predicted by a traditional interceptmodel. As the fundamental power level increases, the dy-namic range does not decrease significantly. For 2-tonecentered at 20MHz, with 10dBm/tone into a matched 50Ωload (that is, 2VPP for each tone at the load, which requires8VPP for the overall 2-tone envelope at the output pin), theTypical Characteristics show 63dBc difference between thetest-tone power and the 3rd-order intermodulation spuriouslevels. This exceptional performance improves further whenoperating at lower frequencies.

NOISE PERFORMANCE

Wideband current-feedback op amps generally have a higheroutput noise than comparable voltage-feedback op amps.The OPA2677 offers an excellent balance between voltageand current noise terms to achieve low output noise. Theinverting current noise (24pA/√Hz) is significantly lower thanearlier solutions whereas the input voltage noise (2.0nV/√Hz) is lower than most unity-gain stable, wideband voltage-feedback op amps. This low input voltage noise is achievedat the price of higher noninverting input current noise (16pA/√Hz). As long as the AC source impedance looking out of thenoninverting node is less than 100Ω, this current noise doesnot contribute significantly to the total output noise. The opamp input voltage noise and the two input current noiseterms combine to give low output noise under a wide varietyof operating conditions. Figure 14 shows the op amp noiseanalysis model with all the noise terms included. In thismodel, all noise terms are taken to be noise voltage orcurrent density terms in either nV/√Hz or pA/√Hz.

RG

RF

RS

EO2

Driver

ERS

EN

IN

IN

√4kTRS

√4kTRF

√4kTRFRF

RS

ERS

EN

IN

IN

√4kTRS

√4kTRG

FIGURE 15. Differential Op Amp Noise Analysis Model.

The total output spot noise voltage can be computed as thesquare root of the sum of all squared output noise voltagecontributors. Equation 17 shows the general form for theoutput noise voltage using the terms shown in Figure 13.

(17)

E E I R kTR I R kTR NGO NI BN S S BI F F= + •( ) + + •( ) +

2 2 24 4

Dividing this expression by the noise gain (NG = (1 + RF/RG))gives the equivalent input referred spot noise voltage at thenoninverting input, as shown in Equation 18.

(18)

E E I R kTRI R

NGkTRNGN NI BN S S

BI F F= + •( ) + + •

+

2 2

2

44

Evaluating these two equations for the OPA2677 circuit andcomponent values (see Figure 1) gives a total output spotnoise voltage of 13.5nV/√Hz and a total equivalent input spotnoise voltage of 3.3nV/√Hz. This total input referred spot noisevoltage is higher than the 2.0nV/√Hz specification for the opamp voltage noise alone. This reflects the noise added to theoutput by the inverting current noise times the feedbackresistor. If the feedback resistor is reduced in high-gain con-figurations (as suggested previously), the total input referredvoltage noise given by Equation 18 approaches just the 2.0nV/√Hz of the op amp. For example, going to a gain of +10 usingRF = 298Ω gives a total input referred noise of 2.3nV/√Hz.

DIFFERENTIAL NOISE PERFORMANCE

As the OPA2677 is used as a differential driver in xDSLapplications, it is important to analyze the noise in such aconfiguration. Figure 15 shows the op amp noise model forthe differential configuration.

FIGURE 14. Op Amp Noise Analysis Model.

4kTRG

RG

RF

RS

1/2OPA2677

IBI

EO

IBN

4kT = 1.6E –20Jat 290°K

ERS

ENI

√4kTRS

√4kTRF


As a reminder, the differential gain is expressed as:

GR

RDF

G= + •

12

(19)

The output noise can be expressed as shown below:

(20)

E G e i R kTR iR kTR GO D N N S S I F F D= • • + •( ) +

+ ( ) + ( )2 4 2 2 42 2 2 2

Dividing this expression by the differential noise gain(GD = (1 + 2RF/RG)) gives the equivalent input referredspot noise voltage at the noninverting input, as shown inEquation 21.

(21)

E e i R kTRiRG

kTRGO N N S S

I F

D

F

D= • + •( ) +

+

+

2 4 2 242 2

2

Evaluating these equations for the OPA2677 ADSL circuitand component values of Figure 6 gives a total output spotnoise voltage of 31.8nV/√Hz and a total equivalent input spotnoise voltage of 3.6nV/√Hz.

In order to minimize the output noise due to the noninvertinginput bias current noise, it is recommended to keep thenoninverting source impedance as low as possible.

DC ACCURACY AND OFFSET CONTROL

A current-feedback op amp such as the OPA2677 providesexceptional bandwidth in high gains, giving fast pulse settlingbut only moderate DC accuracy. The Electrical Characteris-tics show an input offset voltage comparable to high-speed,voltage-feedback amplifiers; however, the two input biascurrents are somewhat higher and are unmatched. Whilebias current cancellation techniques are very effective withmost voltage-feedback op amps, they do not generally re-duce the output DC offset for wideband current-feedback opamps. Because the two input bias currents are unrelated inboth magnitude and polarity, matching the input sourceimpedance to reduce error contribution to the output isineffective. Evaluating the configuration of Figure 1, usingworst-case +25°C input offset voltage and the two input biascurrents, gives a worst-case output offset range equal to:

VOFF = ± (NG • VOS(MAX)) + (IBN • RS/2 • NG) ± (IBI • RF)

where NG = noninverting signal gain

= ± (4 • 4.5mV) + (30µA • 25Ω • 4) ± (402Ω • 30µA)

= ±18mV + 3mV ± 12.06mV

VOFF = –29.06mV to +35.06mV

THERMAL ANALYSIS

Due to the high output power capability of the OPA2677, heat-sinking or forced airflow may be required under extremeoperating conditions. Maximum desired junction temperaturesets the maximum allowed internal power dissipation as de-scribed below. In no case should the maximum junction tem-perature be allowed to exceed 175°C. Operating junctiontemperature (TJ) is given by TA + PD • θJA. The total internal

power dissipation (PD) is the sum of quiescent power (PDQ) andadditional power dissipation in the output stage (PDL) to deliverload power. Quiescent power is the specified no-load supplycurrent times the total supply voltage across the part. PDL

depends on the required output signal and load, but for agrounded resistive load, PDL is at a maximum when the outputis fixed at a voltage equal to 1/2 of either supply voltage (forequal bipolar supplies). Under this condition, PDL = VS

2 /(4 • RL)where RL includes feedback network loading. Note that it is thepower in the output stage and not into the load that determinesinternal power dissipation. As a worst-case example, computethe maximum TJ using an OPA2677 SO-8 in the circuit ofFigure 1 operating at the maximum specified ambient tempera-ture of +85°C with both outputs driving a grounded 20Ω load to+2.5V.

PD = 12V • 18mA + 2 • [62 / (4 • (20Ω || 534Ω))] = 882mW

Maximum TJ = +85°C + (0.83 • 125°C/W) = 170°C

This absolute worst-case condition exceeds specified maxi-mum junction temperature. This extreme case is not normallyencountered. Where high internal power dissipation is antici-pated, consider the thermal slug package version.

BOARD LAYOUT GUIDELINES

Achieving optimum performance with a high-frequency am-plifier like the OPA2677 requires careful attention to boardlayout parasitic and external component types. Recommen-dations that optimize performance include:

a) Minimize parasitic capacitance to any AC ground for allof the signal I/O pins. Parasitic capacitance on the output andinverting input pins can cause instability; on the noninvertinginput, it can react with the source impedance to causeunintentional band limiting. To reduce unwanted capaci-tance, a window around the signal I/O pins should be openedin all of the ground and power planes around those pins.Otherwise, ground and power planes should be unbrokenelsewhere on the board.

b) Minimize the distance (< 0.25") from the power-supplypins to high-frequency 0.1µF decoupling capacitors. At thedevice pins, the ground and power plane layout should notbe in close proximity to the signal I/O pins. Avoid narrowpower and ground traces to minimize inductance betweenthe pins and the decoupling capacitors. The power-supplyconnections (on pins 4 and 7) should always be decoupledwith these capacitors. An optional supply decoupling capaci-tor across the two power supplies (for bipolar operation)improves 2nd-harmonic distortion performance. Larger (2.2µFto 6.8µF) decoupling capacitors, effective at lower frequency,should also be used on the main supply pins. These can beplaced somewhat farther from the device and may be sharedamong several devices in the same area of the PCB.

c) Careful selection and placement of external compo-nents preserve the high-frequency performance of theOPA2677. Resistors should be a very low reactance type.Surface-mount resistors work best and allow a tighter overalllayout. Metal film and carbon composition axially leadedresistors can also provide good high-frequency performance.


Again, keep leads and PCB trace length as short as possible.Never use wire-wound type resistors in a high-frequencyapplication. Although the output pin and inverting input pinare the most sensitive to parasitic capacitance, always posi-tion the feedback and series output resistor, if any, as closeas possible to the output pin. Other network components,such as noninverting input termination resistors, should alsobe placed close to the package. Where double-side compo-nent mounting is allowed, place the feedback resistor directlyunder the package on the other side of the board betweenthe output and inverting input pins. The frequency responseis primarily determined by the feedback resistor value asdescribed previously. Increasing the value reduces the band-width, whereas decreasing it gives a more peaked frequencyresponse. The 402Ω feedback resistor used in the TypicalCharacteristics at a gain of +4 on ±6V supplies is a goodstarting point for design. Note that a 511Ω feedback resistor,rather than a direct short, is recommended for the unity-gainfollower application. A current-feedback op amp requires afeedback resistor even in the unity-gain follower configura-tion to control stability.

d) Connections to other wideband devices on the boardmay be made with short direct traces or through onboardtransmission lines. For short connections, consider the traceand the input to the next device as a lumped capacitive load.Relatively wide traces (50mils to 100mils) should be used,preferably with ground and power planes opened up aroundthem. Estimate the total capacitive load and set RS from theplot of Recommended RS vs Capacitive Load. Low parasiticcapacitive loads (< 5pF) may not need an RS because theOPA2677 is nominally compensated to operate with a 2pFparasitic load. If a long trace is required, and the 6dB signalloss intrinsic to a doubly-terminated transmission line isacceptable, implement a matched impedance transmissionline using microstrip or stripline techniques (consult an ECLdesign handbook for microstrip and stripline layout tech-niques). A 50Ω environment is normally not necessary onboard; in fact, a higher impedance environment improvesdistortion (see the distortion versus load plots). With acharacteristic board trace impedance defined based on boardmaterial and trace dimensions, a matching series resistorinto the trace from the output of the OPA2677 is used, as wellas a terminating shunt resistor at the input of the destinationdevice. Remember also that the terminating impedance isthe parallel combination of the shunt resistor and the inputimpedance of the destination device.

This total effective impedance should be set to match thetrace impedance. The high output voltage and current capa-bility of the OPA2677 allows multiple destination devices tobe handled as separate transmission lines, each with theirown series and shunt terminations. If the 6dB attenuation ofa doubly-terminated transmission line is unacceptable, along trace can be series-terminated at the source end only. FIGURE 16. Internal ESD Protection.

Treat the trace as a capacitive load in this case and set theseries resistor value as shown in the plot of RS vs CapacitiveLoad. However, this does not preserve signal integrity as wellas a doubly-terminated line. If the input impedance of thedestination device is low, there is some signal attenuationdue to the voltage divider formed by the series output into theterminating impedance.

e) Socketing a high-speed part like the OPA2677 is notrecommended. The additional lead length and pin-to-pincapacitance introduced by the socket can create an ex-tremely troublesome parasitic network, which can make italmost impossible to achieve a smooth, stable frequencyresponse. Best results are obtained by soldering the OPA2677directly onto the board.

f) Use the –VS plane to conduct heat out of theHSOP-8 PowerPAD package (OPA2677IDDA) or theQFN-16 (OPA2677IRGV). These packages attach the diedirectly to an exposed thermal pad on the bottom, whichshould be soldered to the board. This pad must be connectedelectrically to the same voltage plane as the most negativesupply applied to the OPA2677 (in Figure 6, this would beground), which must have a minimum area of 3.5" x 3.5"(88.9mm x 88.9mm) to produce the θJA values in the Electri-cal Characteristics tables.

INPUT AND ESD PROTECTION

The OPA2677 is built using a very high-speed complemen-tary bipolar process. The internal junction breakdown volt-ages are relatively low for these very small geometry devicesand are reflected in the absolute maximum ratings table. Alldevice pins have limited ESD protection using internal diodesto the power supplies, as shown in Figure 16.

These diodes provide moderate protection to input overdrivevoltages above the supplies as well. The protection diodescan typically support 30mA continuous current. Where highercurrents are possible (for example, in systems with ±15Vsupply parts driving into the OPA2677), current-limiting se-ries resistors should be added into the two inputs. Keepthese resistor values as low as possible, since high valuesdegrade both noise performance and frequency response.

ExternalPin

+VS

–VS

InternalCircuitry


DATE REVISION PAGE SECTION DESCRIPTION

Changed Storage Temperature Range from −40°C to +125°C to−65°C to +125°C.

3 Electrical Characteristics Added Both Channels; Power Supply section under Conditions.

4 Electrical Characteristics Added +5V and Both Channels; Power Supply section under Conditions.

Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

H3/08

7/08 I 2 Abs Max Ratings

PACKAGE OPTION ADDENDUM

www.ti.com 23-Aug-2017

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status(1)

Package Type PackageDrawing

Pins PackageQty

Eco Plan(2)

Lead/Ball Finish(6)

MSL Peak Temp(3)

Op Temp (°C) Device Marking(4/5)

Samples

OPA2677IDDA ACTIVE SO PowerPAD DDA 8 75 Green (RoHS& no Sb/Br)

CU NIPDAU Level-3-260C-168 HR -40 to 85 OP2677

OPA2677IDDAG4 ACTIVE SO PowerPAD DDA 8 75 Green (RoHS& no Sb/Br)


OPA2677IDDAR ACTIVE SO PowerPAD DDA 8 2500 Green (RoHS& no Sb/Br)


OPA2677IDDARG4 ACTIVE SO PowerPAD DDA 8 2500 Green (RoHS& no Sb/Br)


OPA2677IRGVT ACTIVE VQFN RGV 16 250 Green (RoHS& no Sb/Br)

CU NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA2677

OPA2677IRGVTG4 ACTIVE VQFN RGV 16 250 Green (RoHS& no Sb/Br)

CU NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA2677

OPA2677U ACTIVE SOIC D 8 75 Green (RoHS& no Sb/Br)

CU NIPDAU Level-2-260C-1 YEAR OPA2677U

OPA2677U/2K5 ACTIVE SOIC D 8 2500 Green (RoHS& no Sb/Br)

CU NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA2677U

OPA2677UG4 ACTIVE SOIC D 8 75 Green (RoHS& no Sb/Br)

CU NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA2677U

(1) The marketing status values are defined as follows:ACTIVE: Product device recommended for new designs.LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.PREVIEW: Device has been announced but is not in production. Samples may or may not be available.OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substancedo not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI mayreference these types of products as "Pb-Free".RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide basedflame retardants must also meet the <=1000ppm threshold requirement.

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

http://www.ti.com/product/OPA2677?CMP=conv-poasamples#samplebuy









PACKAGE OPTION ADDENDUM


Addendum-Page 2

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuationof the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finishvalue exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on informationprovided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken andcontinues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device PackageType

PackageDrawing

Pins SPQ ReelDiameter

(mm)

ReelWidth

W1 (mm)

A0(mm)

B0(mm)

K0(mm)

P1(mm)

W(mm)

Pin1Quadrant

OPA2677IDDAR SOPower PAD

DDA 8 2500 330.0 12.4 6.4 5.2 2.1 8.0 12.0 Q1

OPA2677IRGVT VQFN RGV 16 250 180.0 12.4 4.25 4.25 1.15 8.0 12.0 Q2

OPA2677U/2K5 SOIC D 8 2500 330.0 12.4 6.4 5.2 2.1 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION


Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

OPA2677IDDAR SO PowerPAD DDA 8 2500 367.0 367.0 35.0

OPA2677IRGVT VQFN RGV 16 250 210.0 185.0 35.0

OPA2677U/2K5 SOIC D 8 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION


Pack Materials-Page 2

GENERIC PACKAGE VIEW

Images above are just a representation of the package family, actual package may vary.Refer to the product data sheet for package details.

DDA 8 PowerPAD TM SOIC - 1.7 mm max heightPLASTIC SMALL OUTLINE

4202561/G

www.ti.com

PACKAGE OUTLINE

C TYP6.2

5.8

1.7 MAX

6X 1.27

8X 0.510.31

2X3.81

TYP0.250.10

0 - 80.150.00

2.62.0

3.12.5

0.25GAGE PLANE

1.270.40

A

NOTE 3

5.04.8

B 4.03.8

4221637/B 03/2016

PowerPAD SOIC - 1.7 mm max heightDDA0008JPLASTIC SMALL OUTLINE

NOTES: 1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing per ASME Y14.5M. 2. This drawing is subject to change without notice. 3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not exceed 0.15 mm per side. 4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm per side.5. Reference JEDEC registration MS-012, variation BA.

PowerPAD is a trademark of Texas Instruments.

TM

18

0.1 C A B

54

PIN 1 IDAREA

NOTE 4

SEATING PLANE

0.1 C

SEE DETAIL A

DETAIL ATYPICAL

SCALE 2.400

EXPOSEDTHERMAL PAD

4

1

5

8

www.ti.com

EXAMPLE BOARD LAYOUT

(5.4)

0.07 MAXALL AROUND

0.07 MINALL AROUND

8X (1.55)

8X (0.6)

6X (1.27)

(2.95)NOTE 9

(4.9)NOTE 9

(2.6)

(3.1)SOLDER MASK

OPENING

( ) TYPVIA

0.2

(1.3) TYP

(1.3)TYP

4221637/B 03/2016


SYMM

SYMM

SEE DETAILS

LAND PATTERN EXAMPLESCALE:10X

1

4 5

8

SOLDER MASKOPENING

METAL COVEREDBY SOLDER MASK

SOLDER MASKDEFINED PAD

NOTES: (continued) 6. Publication IPC-7351 may have alternate designs. 7. Solder mask tolerances between and around signal pads can vary based on board fabrication site. 8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).9. Size of metal pad may vary due to creepage requirement.

TM

METALSOLDER MASKOPENING

NON SOLDER MASKDEFINED

SOLDER MASK DETAILS

OPENINGSOLDER MASK METAL UNDER

SOLDER MASK

SOLDER MASKDEFINED

www.ti.com

EXAMPLE STENCIL DESIGN

8X (1.55)

8X (0.6)

6X (1.27)

(5.4)

(2.6)

(3.1)BASED ON

0.127 THICKSTENCIL

4221637/B 03/2016


2.20 X 2.620.1752.37 X 2.830.150

2.6 X 3.1 (SHOWN)0.1252.91 X 3.470.1

SOLDER STENCILOPENING

STENCILTHICKNESS

NOTES: (continued) 10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate design recommendations. 11. Board assembly site may have different recommendations for stencil design.

TM

SOLDER PASTE EXAMPLEEXPOSED PAD

100% PRINTED SOLDER COVERAGE BY AREASCALE:10X

SYMM

SYMM

1

45

8

BASED ON0.125 THICK

STENCIL

BY SOLDER MASKMETAL COVERED SEE TABLE FOR

DIFFERENT OPENINGSFOR OTHER STENCILTHICKNESSES

http://www.ti.com/lit/slua271

IMPORTANT NOTICE

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