ee241 - spring 2005bwrcs.eecs.berkeley.edu/classes/icdesign/ee241_s05/... · 13 579 fan-in 0.0 1.0...
TRANSCRIPT
![Page 1: EE241 - Spring 2005bwrcs.eecs.berkeley.edu/Classes/icdesign/ee241_s05/... · 13 579 fan-in 0.0 1.0 2.0 3.0 4.0 t p (nsec) t pHL t p t pLH linear quadratic AVOID LARGE FAN-IN GATES!](https://reader034.vdocument.in/reader034/viewer/2022052017/602f74cbc894b912ee3b6c84/html5/thumbnails/1.jpg)
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EE241 - Spring 2005Advanced Digital Integrated Circuits
Lectures 4 and 5:Delay Modeling
2
ISSCC 2005
Keynotes (Monday Morning)Nanoelectronics for the Ubiquitous Information Society, Daeje Chin, Minister of Information and Communications, Korea
Ambient Intelligence: Broad Dreams and NanoscaleRealities, Hugo De Man, IMEC, Katholieke Universiteit Leuven, Belgium
Innovation and Integration in the Nanoelectronics Era, SunlinChou, Intel, Hillsboro, OR
Interesting Short Courses (on Su)Memory forum
“When processors hit the power wall”
3d Integration
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3
Interesting SessionsMo
Non-Volatile MemoriesMultimedia Processing
TuHigh-speed links and clock generatorsMicroprocessors and Signal processingLow-Power wireless and advanced integrationClock distribution and power controlHigh-speed interconnects and building blocks
WeProcessor building blocksPLL, DLL and VCO’sDRAMSRAMClocking and I/O
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Some other stuff
Panels:Mo: Towards the Nanoscale transistor
Tu: SRAM Design in the Nanoscale Era
Th Circuit Design ForumRobust Design for Nanoscale circuits
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5
Projects
Projects info –Target benchmark: ultra low-power 8051 microcontroller (and its components)
Some interesting projects:
Ultra low-power clock dividers/multipliers
Sub-threshold design versus low-threshold design
Low-current voltage converters/multipliers
Design techniques for self-calibration
Energy recovery and reversible computing
The return of “current-driven logic”
Error-resilient circuits and architectures
Small-granularity self-timing / asynchronous
Device Models
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K(VGS –VTHZ) Model
Drain current vs. gate-source voltage
0.0E+00
2.0E-04
4.0E-04
6.0E-04
8.0E-04
0 0.2 0.4 0.6 0.8 1 1.2
V GS [V]
I DS
[A
]
VTHZ
8
Transistor Leakage
-9
-8
-7
-6
-5
-4
-3
0 0.2 0.4 0.6 0.8 1 1.2
V GS [V]
log
I DS
[lo
g A
]
Subthreshold slope
Leakage current is exponential with VGS
VDS = 1.2V
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Transistor Leakage
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Transistor Leakage
Two effects:• diffusion current (like a bipolar transistor)• exponential increase with VDS (DIBL)
0
2
4
6
8
0 0.2 0.4 0.6 0.8 1 1.2 1.4
V DS [V]
I DS
[nA
]
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Subthreshold Current
Subthreshold behavior can be modeled physically
⎟⎟⎟
⎠
⎞
⎜⎜⎜
⎝
⎛−⎟⎟
⎠
⎞⎜⎜⎝
⎛µ=
−−
qkT
Vds
qkTmThVgV
ds eeq
kT
L
WI 1
2
( )S
dsVThVgsV
ds W
WII
γ+−
= 100
0
Or: [Taur, Ning]
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Leakage Components
Courtesy of IEEE Press, New York. © 2000
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Leakage Components
1. pn junction reverse bias current
2. Weak inversion
3. Drain-induced barrier lowering (DIBL)
4. Gate-induced drain leakage (GIDL)
5. Punchthrough
6. Narrow width effect
7. Gate oxide tunneling
8. Hot carrier injection
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Leakage Components
Drain-induced barrier lowering (DIBL)Voltage at the drain lowers the source potential barrier
Lowers VTh, no change on S
Gate-induced drain leakage (GIDL)High field between gate and drain increases injection of carriers into substrate -> leakage (band-to-band leakage)
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DIBL, GIDL, Weak Inversion
Courtesy of IEEE Press, New York. © 2000
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Stack Effect
NAND gate:
Reduction:
Courtesy of IEEE Press, New York. © 2000
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MOS Transistor as a Switch
Discharging a capacitor
• Can solve:
( )DSDD vii =
dt
dVCi DS
D =
• Prefer using equivalentresistances
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MOS Transistor as a SwitchTraversed path
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MOS Transistor as a Switch
Solving the integral:
Averaging resistances:
with appropriately calculated Idsat
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Equivalent Resistance
W/L=1, L=0.25µ
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CMOS Performance
Propagation delay: ( ) LeqnpHL CRt 2ln= ( ) LeqppLH CRt 2ln=
Short channel Long channel
)( DDeq VfR ≠DD
eq VR
1∝
TDD VV >>for
ln2 = 0.7
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MOS Capacitances
CGSO = CGDO
= CoxxdW
= CoW
Gate Capacitance
Overlap Capacitance
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MOS Capacitances
Gate capacitanceNon-linear channel capacitance
Linear overlap, fringing capacitances
Miller effect on overlap capacitance
Non-linear drain diffusion capacitancePN junction
Wiring capacitancesLinear
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Gate Capacitance
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MOS Capacitances
0.25µm process
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Gate and Drain Capacitances
0.13um Cdb/um vs. Vds
0.00E+00
2.00E-16
4.00E-16
6.00E-16
8.00E-16
1.00E-15
1.20E-15
1.40E-15
1.60E-15
1.80E-15
2.00E-15
0.0 0.4 0.8 1.2
Vds (V)
Cdb
(F
)
NMOS VGS=0
PMOS VGS=0
Gate capacitance Drain Capacitance
0.13um Cgs/um vs. Vgs
0.0E+00
2.0E-16
4.0E-16
6.0E-16
8.0E-16
1.0E-15
1.2E-15
1.4E-15
1.6E-15
1.8E-15
2.0E-15
0.0 0.4 0.8 1.2
Vgs [V]
Cgs
[F]
NMOS VDS=VDD
PMOS VDS=VDD
NMOS VDS=0
PMOS VDS=0
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Gate Capacitances
Gate capacitance is non-linearFirst order approximation with CoxWL (CoxL = 2fF/µm)This is an overestimation
Need to find the actual equivalent capacitance by simulating itSince this is a linear approximation of non-linear function, it is valid only over the certain range
Different capacitances for HL, LH transitions and power computation
Drain capacitance non-linearity compensatesBut this changes with fanout
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Gate Capacitance vs. VTh, VDD
Nose, Sakurai, ISLPED’00
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FO4 Inverter Delay
In
tpShapes the input slope to FO4
FO4 load Suppresses Millerkickback
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Calibrating Delays
Step RC delay model is a good first-order approximation
Accuracy can be improved by including:Slope effects
Non-linear capacitive loading
Signal arrival times
Wire models
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Input Slope
Simulated vs. linear model
0
10
20
30
40
50
60
70
0 2 4 6 8 10
FanOut
Del
ay [
ps]
8
4
1
Driving gatefanout
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Input slope
We can model the delay as tp = 0.7*RekvCWhen driving with non-step input, the rise/fall time is absorbed into Rekv
Rekv is different than one extracted straight from I-V
The output delay is linearly dependent on input rise/fall time tp = 0.7RC + ηtS
η = 0.17 in this example (~1/6)The model is limited to a range of fanouts
More accurate delay models propagate two quantities: delay and signal slope
Both can be modeled either as linear or table lookups
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Standard Cell LibraryContains for each cell:
Functional information: cell = a *b * c
Timing information: function of
input slew
intrinsic delay
output capacitance
non-linear models used in tabular approach
Physical footprint (area)
Power characteristics
Wire-load models - function ofBlock size
Fan-out
Library
[from K. Keutzer]
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Synopsys Delay Models
Linear (CMOS2) delay model
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Example Cell Timing
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Delay Dependency on Edge Rate
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Transition Time
Linear: Piecewiselinear:
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Cell Characterization
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Synopsys Nonlinear Delay Model
Delay is a function of:
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Synopsys Nonlinear Delay Model
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Static Timing Analysis
clk
Combinationallogic
clk
Combinationallogic
clk
Combinationallogic
original circuit
extracted block
Combinationallogic
[from K. Keutzer]
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Each Combinational Block
Arrival time in green A
C
B
f
2
2
2
1
0
1
0
.20.20
.20
.10
X
YZ
W
.15
.05
.05
.05
Interconnect delay in red
Gate delay in blue
What’s the right mathematical object to use to represent this physical object?
[from K. Keutzer]
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Problem formulation - 1
C
B
f
X
Y
W
0
.05.1
1
.2
0
0
1
A
.15.20
.20
A
C
B
f
2
2
2
1
0
1
0
.20.20
.20
.10
X
YZ
W
.15
.05
.05
.05
12
2
2
Z
Use a labeled directed graph
G = <V,E>
Vertices represent gates, primary inputs and primary outputs
Edges represent wires
Labels represent delays
Now what do we do with this?
[from K. Keutzer]
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Problem formulation - Arrival Time
Arrival time A(v) for a node v is time when signal arrives at node v
uu
A( ) max (A(u) d )→υ∈ υ
υ = +FI( )
X
Y
A(Z)
Z
x zd →
Y zd →
A(X)
A(Y)
where d is delay from to andu u, {X,Y}, {Z}.υ→ υ υ FI(υ) = =
[from K. Keutzer]
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Static Timing Analysis
Computing critical (longest) path delayLongest path algorithm on DAG [Kirkpatrick, IBM Jo. R&D, 1966]
Used in most ASIC designs today
LimitationsFalse paths
Simultaneous arrival times
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Signal Arrival Times
NAND gate:
1
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Signal Arrival Times
NAND gate:
1
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Simultaneous Arrival Times
NAND gate:
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Impact of Arrival Times
A
B
Delay
0 tA - tB
A arrives early B arrives early
Up to 25%
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Optimization for Performance
Performance critical blocks
Start with a synthesized designEasier to explore architectures
Easy to verify
Provides some level of performance optimization
Understand the limits of synthesized designs
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Optimization for Performance
Options• Technology choice
CMOS, bipolar, BiCMOS, GaAs, Superconducting• Logic level optimizations
logic depth, network topology, fan-out, gate complexity• Circuit optimizations
logic style, transistor sizing• Physical optimization
implementation choice, layout strategy
• Do not ignore wiring!!
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Logic Level Optimizations
R R
Logic Depth
or
Techniques: Restructuring, pipelining, retiming, technology mapping
Well covered by today’s logic and sequential synthesis
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Logic Optimizations (2)
Technique: Removal of common sub-expressionStart from tree structure/output
Fanout
Tp = O(FO) also effects wiring capacitance
Late arriving
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Logic Optimizations (3)
1 3 5 7 9fan-in
0.0
1.0
2.0
3.0
4.0
t p(n
sec)
tpHL
tp
tpLHlinear
quadratic
AVOID LARGE FAN-IN GATES! (Typically not more than FI < 4)
Tp = O(FI2) !Observation: only true if FI
translates in series devices -
otherwise linear
e.g. NAND pull-down
NOR pull-up
Fanin
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Logic Optimizations (4)
Fan-out
t p(p
sec)
t
1 2 3 4 5 6 7
pINVtpNAND
F(Fan-in)
Slope is a function of “driving strength”
pNORt
All the gates have the same drive current
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Technology Mapping for Performance
Alternative coverings
Use low FI modules on critical path(s)Library composition?