ee245 2.4 ghz indoor channel

EE245 WIRELESS DESIGN LABORATORY: 2.4 GHZ INDOOR CHANNEL

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AbstractThis paper presents the results of indoor wireless

channel measurements and characterization of an indoor hallway at the 2.4 GHz ISM band (2.4-2.83GHz). Important channel parameters like the propagation loss factor, Ricean K-factor, delay spread, and coherence bandwidth are calculated and discussed. The measurements were performed for different antenna configurations in a hallway environment. All channel parameters were derived from the power transfer function and its associated channel impulse response. These estimated values were also compared to a 2-D Finite Difference Time Domain (FDTD) model, assuming perfectly conducting walls along the hallway. From the simulation, power profiles in time were obtained, from which path loss, delay spread and coherence bandwidth were calculated. The path loss factor is found to be lower than the free space attenuation. The mean value of the Ricean K-factor is approximately independent of distance and varies for each antenna configuration. The delay spread was found to be about 10 ns for indoor sites, whereas the coherence bandwidth was around 6 MHz.

I. INTRODUCTION

OST of the early research efforts in wireless communication were made on individual radio links from

one transmitter to one receiver. The main problem was to send signals over an unreliable radio link. The signal that arrives at the receiver is greatly distorted due to time-varying and frequency-selective fading. The radio waves are scattered from reflecting obstacles and arrive at the receiver with different delays, producing multi-path fading especially in indoor communication with a dense architecture. These problems have been largely handled after the introduction of digital communication techniques.

In parallel to this evolution, the fixed Local Area Networks (LAN) are becoming wireless. A Wireless LAN (WLAN) is aimed to provide local wireless access to fixed network architectures. WLAN developments are now driven by the success of the Internet with its services and applications. For a

This paper is submitted as Final Report of the course EE245 Wireless

Design Laboratory at Stanford University coordinated by Professor D. Leeson.

The authors are graduate students in the Electrical Engineering Department, Stanford University, Stanford, CA 94305. They can be reached by e-mail at: {eulffe, platinom, barretod}@stanford.edu.

while, the WLAN market did not bloom, probably because the low data rate (2 Mbps) that was offered. In July 1990, the Institute of Electrical and Electronics Engineers formed a working group (IEEE 802.11) to establish a world wide standard for WLANs. By that time, a number of Radio LAN products had been developed in the ISM (Industrial, Scientific and Medical) bands, 902-928 MHz, 2400-2483.5 MHz and 5725-5875 MHz. These frequency bands are unlicensed and can be used for data transmissions if a number of rules are followed. The efforts that the working group IEEE 802.11 made resulted in an approved standard for WLANs in June 1997.

The standard consists of three different physical layer specifications, one with infrared and two with radio transmissions. In 1999 this Task group finished with the specification for a new physical layer (802.11b) that increased the bit rates from 1 and 2 Mbps to 5.5 and 11 Mbps. Most recently, IEEE published the 802.11g standard that supports a data rate of up to 54 Mbps.

The radio standards for WLANs operate in the 2.4 GHz band, the only accepted ISM band available worldwide. In that sense, a careful characterization in the WLAN signal propagation channel is essential for determining optimum methods to achieve the proposed data rates and quality of service. This paper describes results of wireless propagation measurements carried out at a center frequency of 2.45 GHz, which may be useful in design of communication systems operating in such band like those compliant with IEEE 802.11, Bluetooth and HomeRF open wireless standard.

II. MEASUREMENT SYSTEM

A. Measurement Setup

The measurement system for the indoor wireless channel is based in a swept frequency technique called channel sounder. As shown in Figure 1, that system is built around a vector network analyzer (VNA) Agilent 8753ES, which is capable of measuring the S-parameters of any device under test (DUT) connected between its two ports.

Two antennas are connected to the ports of the test set of the VNA with the transmitting antenna associated to the first port and the receiving to the second one. The connection of the antennas was made using coaxial cables. The transmitter line (7.5 ft antenna cable and connectors) and the receiver line

Measuring the 2.4 GHz Band for Indoor Wireless Communications

Enrique Ulffe, Manuel Platino and Daniel Barreto, Stanford University

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(3.75 ft antenna cable) have a loss of 1.55 dB each. The connecting path between the receiver line and the VNA has a loss of 3.6 dB (25 ft coaxial cable) or 17.2 dB (100 ft coaxial cable). In this way, when a VNA is used as part of a channel sounder, the DUT is the wireless channel, which includes transmitting and receiving antennas, and the propagation channel between them. Measuring the S21 parameter, the frequency response of the wireless channel is obtained in a 100 MHz band centered at 2.45 GHz (? 12.2 cm), taking 201 points in the frequency scanning. The 201-points impulse response from 0 to 300 ns was obtained through an inverse fast Fourier transformation performed by the VNA with a frequency span of 1500 MHz (1.5-3.0 GHz).

Network AnalyzerAgilent 8753ES

Transmit AntennaReceive Antenna

2.4-2.5 GHz

Port 1 Port 2

FloppyDisk

PC - Processing

Fig. 1. Measurement setup. The measurements were conducted connecting a standard 0

dBi dipole to port 1 (transmitter) and a receive antenna to port 2 and placing one in front of the other, perfectly aligned. The receive antenna was chosen from a set consisting of another 0 dBi dipole, a 10-element 11 dBi Yagi, and a LHCP 18 dBi Helix, in order to set different scenarios and measure the influence of polarization and directivity of the receive antenna in the characterization of the indoor wireless channel.

Fig. 2. Measuring the channel parameters.

A calibration was first done to measure the effect of auxiliary elements as cables and adapters by connecting transmitting and receiving circuits one to the other, skipping both antennas. In order to ensure that the incident and transmitted waves could be approximated as plane waves, the antennas were placed 30.5 cm (1 ft) apart as point of reference, which is larger enough than the far-field distance of 5.5 cm (r = 2d2/?, where d = length of transmit dipole).

Fig. 3. Frequency Response Measurement. The transmitter antenna was stationary while the receiver

was being moved along a linear path by means of a manual positioning system scaled by the 1-ft square-carpeted floor. Frequency response measurements (transfer functions) as the sample showed in Figure 3 were taken every 5 ft along a 70 ft linear path. Impulse response measurements were taken every 3 ft along a 15 ft linear path (fraction of the first 70 ft path). A typical impulse response measurement (multipath delay profile) is shown in Figure 4. At each position, a set of four measurements spaced equally by a fraction of wavelength were taken for further averaging and processing.

Fig.4. Impulse Response Measurement.

Main Path

Multipath


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B. Environment Description

The measurements were performed on a hallway in the basement of the Packard Building (Electrical Engineering) of Stanford University. The measurement environment was the same for each antenna configuration as shown in Figure 5.

Fig. 5. Hallway environment showing carpeted floor and ceiling covered with polystyrene tiles and the external metallic conduit.

The outer surfaces of the walls in the hallway are made of

plasterboard and a thin layer of paint. The inner part of the structure is made of wood with metal pillar. The floors are protected with a square-grid synthetic carpet. The ceilings are covered with polystyrene tiles and include an external metallic conduit and fluorescent lamps positioned across the corridor. The Hallway contains doors on both sides.

13"19"

49"

Tx

69.5"

38"

19"

6 ft

3 ft

Conduit

70 ft

Tx

6 ft

13"

19" 4 ft

5 ft

33 ft

4 ft 4 ft

9 f t

Front View

Top View

Packard BuildingBasement

Fig. 6. Floor plan of the measurement environment. The transmitter was mounted in a fixed position at a height

of 176.5 cm above floor level and at one end for the hallway propagation measurements. The receiver at the other end and at the same height was moved in steps (5-ft for frequency

response and 1-ft for impulse response) to different fixed locations along the center line shown in Figure 6. Due to the movement of people and equipment indoor propagation channels are in general, time varying. Since the purpose of these measurements was to investigate the wireless channel spatial variability, it was essential to keep the channel stationary during measurements. The experiments took place during the night (1 am to 5 am) to ensure that the channel was free of movement by people and equipment.

C. Antenna Design

The main objective of the project is to compare the parameters of the wireless indoor channel at 2.45 GHz for different transmit-receive antenna configurations. In that sense, we fixed the transmit antenna (Dipole Vertically-Polarized) and measured the impulse response and frequency response of the channel for the following receive antenna scenarios:

a. Dipole Vertically-Polarized b. Yagi Vertically-Polarized c. Helix Left-Hand-Circularly-Polarized d. Yagi Horizontally-Polarized Both transmit and receive antennas were mounted at the

same height above floor level and 125 cm from the ceiling. The innovative mounting structure showed in Figure 5 was designed based on portable and mobile coat hangers, which allowed the mobility required for the different set of measurements. Following is a brief description of the antennas employed in this project.

1) Dipole: The Linearly Polarized half-wavelength Dipole

antennas used in the measurements were obtained from the equipment of the EE245 Wireless Design Laboratory of Stanford University. Both were designed and constructed following the criteria of 0.95?/2 (about 5.80 cm for 2.45 GHz) as the length of the radiating element. Therefore, each pole has a length of 0.4755? (2.9 cm) such that the antenna can generate an omnidirectional pattern with 0 dBi of gain. The transmit dipole is shown in Figure 7.

Fig.7. Dipole 0 dBi antenna.


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2) Yagi: The Linearly Polarized 10-elements Yagi antenna showed in Figure 8 was designed for 2.45 GHz and a bandwidth of 35 MHz following the definitions in [4] and [5] as well as the results of the DOS-based software ANTDL6WU by Gnter Hock.

Fig.8. Yagi Antenna.

The antenna was created with copper material: 1 ft of 3/16 round tube for the boom and 1/8 round tube for the elements. The length of the elements is shown in Table I.

TABLE I

LENGTH OF YAGI ANTENNA ELEMENTS

Element Length (mm)

Director 8 50.54 Director 7 50.54 Director 6 50.98 Director 5 51.42 Director 4 51.86 Director 3 52.75 Director 2 53.87 Director 1 54.77

Driver 58.07 Reflector 63.48

We lineup the tip of the boom at 278.11 mm on a straight edge ruler: 276.11 mm for the director 8 element plus 2mm to space it from the tip of the square tube. Next, we marked on the boom the values indicated in Table II which left approximately 26.69mm after the reflector element to mount the Yagi.

TABLE II

ARRANGEMENT OF YAGI ANTENNA ELEMENTS

Element Reference

Director 8 276.11mm Director 7 235.47mm Director 6 196.55mm Director 5 159.47mm Director 4 124.97mm Director 3 94.03mm Director 2 67.51mm Director 1 45.41mm

Driver 35.71mm Reflector 13.00mm

Each of the elements was soldered to the boom at the respective positions and fixed in place with epoxy superglue which is an almost non-conductive material. Special care was taken to center the elements to the boom. The driver was a ?/2 folded dipole. For impedance matching, the folded dipole was directly connected to a 50O RG-8 coaxial cable. We did not use any special matching system (like the ?/4 balun) since we obtained appropriate values of SWR with the simple configuration mentioned before. As can be seen in Figure 9, the value of SWR at 2.45 GHz is about 1.8 whereas the minimum was reached at 2.485 GHz. The elevation and azimuth Half Power Beam Width (HPBW) were about 30 which are slightly lower than the theoretical value of 35 and 32. In addition, the obtained gain was 5.3 dBi, which is less than the designed value of 11 dBi. The deviations can be explained by the fact that we were not able to cut the elements at the exact lengths indicated in Table I and to solder those elements in the precise positions shown in Table II.

Fig.9. SWR for the Yagi Antenna in the range 2.4-2.5 GHz.. The scale is 0.5 per vertical division where point 2 indicates SWR = 1.12

3) Helix: This left-handed circularly polarized (LHCP) antenna shown in Figure 10 is derived from information on helix antennas in [4], [5] and was built by wrapping a 1/8 copper tube around a dielectric support of 45 cm made of 40mm PVC plumbing pipe. In order to obtain a theoretical gain of 18 dBi, the spiral should consist of 17 turns equally spaced by 2.8 cm. The ground plane for the antenna is an 8 aluminum endcap with flat base since we couldnt find a smaller diameter to satisfy our design requirements. C? = circumference of winding = 0.75? -1.33? S? = axial length of one turn = 0.2126C? - 0.2867C? D = diameter of ground plane = 0.8? -1.1? GdB = 11.8 + 10*log10(n*S?*C?

2) , where n = numbers of turns

HPBW = half power beam width = ll SnC

o

*52

(2.1)


5

Fig.10. Helix Antenna.

To construct the antenna we used epoxy superglue each 4

revolutions to hold the copper tube around the tube, following the spiral spacing. As we approached the base end, we didn't glue down the last turn and did leave some excess wire hanging off the end. In order to match the antenna from its nominal 150 O to the cable 50 O impedance, we should have used a 15-mm-wide triangular strip of copper (impedance transformer) to connect the coaxial line to the copper spiral by soldering the narrow pointy end of the strip to the wire and match up where the lower corner at the large end will solder neatly onto the stub of the BNC connector. However, we used a smaller strip which affected the antenna performance and increased its SWR. We also used epoxy to hold it the matching system in place.

Fig.11. SWR for the Helix Antenna in the range 2.4-2.5 GHz.. The scale is 0.5 per vertical division where point 2 indicates SWR = 2.37

Efficiency of the antenna was obtained by measuring the S11

parameter in the VNA as can be seen in Figure 11. Unfortunately, since we were not able to match the antenna appropriately, we obtained values of SWR in the range 2.37-3.37 which are not excellent but good enough for our project purposes. Although the elevation and azimuth HPBW of this antenna were about 20 which is slightly lower than the theoretical value of 24, the antenna gain obtained (5.3 dBi) was very different to the expected which could be explained by

the exceeded diameter of the ground plane as well as our poor matching system.

D. E-field simulation model

In order to compare the measurements with the ideal case of a hallway with perfectly conducting walls, we developed a numerical model based on the Finite Difference Time Domain method, or FDTD. This method was originally proposed by Yee [13] in 1966, when he described the basis of the FDTD numerical technique for solving Maxwells curl equations directly in the time domain on a space grid. In his proposed method, he described how the space can be divided in a regular grid, where each point has associated values of E and H fields and the whole grid is updated every time step. Tafflove and Brodwin later obtained the correct numerical stability criterion for Yees algorithm, based on the appropriate election of the grid spacing (Dx and Dy) and the time step Dt [14, 15]. Later, in 1994, Berenger [16] introduced the Perfectly Matched Layer (PML) boundary conditions to allow free space propagation in a finite closed grid. In this paper we will implement the Berenger PML boundary conditions, using the field splitting method for y < 0 m and y > 22 m, while the path walls were assumed yo be perfectly conductors at x = 2.4m to the right of the hall and at different locations to the left, depending on the layout profile seen in figures 6 and 13. We are using the same set up as described in section B.

Fig.12. The FDTD grid. In this paper we are presenting a 2D simulation based on the TMz mode, where we calculate the values of Ez, Hx and Hy fields.

To determine the values of Dx, Dy and Dt we will use: Dx = Dy = d/40 = 0.06 m and Dt = Dx / (2 vp) = 0.14 nsec, where d = 7.9 ft = 2.4 m, is the hall width at y = 0 m, and vp = 2.99.10

8 m/sec, is the propagation speed of the electromagnetic radiation in free space. These values were chosen following the criterion of stability developed by Taflove [14, 15, 17], trying to use as much as possible our computational power of memory and processing time. We will launch an incident field separately from the total and scattered fields. This incident field will be launched at x = 1.5 m and y = 0 m:


6

( )

-

--=2

sec6.0sec1.2

exp,5.1),,(n

ntymxtyxEinc d (2.2)

The election of this type of wave to be launched in the hallway was one based on the measurement setup. In effect, as described in the previous section, we mentioned that for the time domain measurements we used a bandwidth of 1.5 GHz. In order to follow this set up as close as possible, we chose a Gaussian pulse of the same bandwidth for the simulation.

We are using the Total Scattered fields method [18] for this simulation, dividing the grid in two regions. For y > 10 cm we will use total fields in the calculations and for 0 cm < y < 10 cm we will use only scattered fields. The incident field will be run separately from the total and scattered fields, and will be added to the scattered field at y > 10 cm. (Refer to Figure 13 for the coordinate system used in the simulations)

Fig.13. Hallway layout, top view used for the numerical simulation. The Point labeled Tx, correspond to the transmission antenna, while the points Rxn, n = 16, are the locations of the receiving antenna, where Ez was recorded, at 20 ft, 30 ft, 40 ft, 50 ft, 60 ft, 70 ft respectively.

As said before, to do the simulation problem we will implement the Berenger PML boundary conditions, with a PML region 12 cells thick. This method requires an implementation of an anisotropic medium to absorb the waves traveling away from the grid, simulating an infinite long hall. In order to do this me use a conductivity profile based on a 4th order polynomial that goes from 0 to a maximum value sxmax or symax, depending on the direction where we want to attenuate the wave (hence the term anisotrpic). In this case the walls along the y =0 and y=2.4 m axis are perfect conductors, therefore we only need to

calculate the PML boundaries for y < 0 m and y > 22 m. In order to do this we must implement the difference equations to separate the conductivity in the x and y axes. We start from the expressions of the two dimensional TMz case [17]:

yH

Et

Ex

HE

tE

xE

Ht

HyE

Ht

H

xzyy

zy

yzxx

zx

zyx

y

zxy

x

-=+

=+

=+

-=+

se

se

sm

sm

*0

*0

(2.3)

Discretizing the previous equation we get the following update equations. These are the update equations we will use here to calculate wave propagation. The indexes i and j are the discrete representation of the grid, they are spatial indexes, while n is the temporal index, representing the progression in time by the time step Dt:

( ))2(

2

2

2

2/1

*0

,1,

2/1

*0

2/1

*02/1

2/1,

2/1

2/1,

ty

tEE

t

tHH

jy

n

jizn

jiz

jy

jyn

jix

n

jix

D+D

D--

-

D+

D-=

+

+

+

+-

+

+

+

sm

sm

sm

(2.4a)

( ))2(

2

2

2

2/1

*0

,,1

2/1

*0

2/1

*02/1

,2/1

2/1

,2/1

tx

tEE

t

tHH

ix

n

jizn

jiz

ix

ixn

jiy

n

jiy

D+D

D-+

+

D+

D-=

+

+

+

+-

+

+

+

sm

sm

sm

(2.4b)

)2(2

2

2

,

2/1

,2/1

2/1

,2/1

,

,

,

1

,

txt

HH

t

tEE

ixji

n

jiy

n

jiy

ixji

ixjin

jizxn

jizx

D+DD

-+

+

D+

D-=

+

-

+

+

+

se

se

se

(2.4c)

( ))2(

2

2

2

,

2/1

2/1,

2/1

2/1,

,

,

,

1

,

ty

tHH

t

tEE

jyji

n

jixn

jix

jyji

jyjin

jizy

n

jizy

D+DD

--

-

D+

D-=

+

-

+

+

+

se

se

se

(2.4d)

1

,

1

,

1

,

+++ +=n

jizyn

jizxn

jizEEE (2.4e)

We see from these update equations that the conductivities along x and y depend only on the respective indexes i and j, as was expected. The electric and magnetic conductivities will have the mentioned before, 4th order polynomial grading as:

[ ]

d

Rmdy

y yy

m

y

+-=

=

0

0maxmax

2

)0(ln)1()(

em

sss (2.5)


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This expression is taken from [17], and was derived from several experiments with the PML, under different conditions. In this equation we used a 4th order polynomial, therefore m = 4. We also used R(0) = e-8 and d = 12.Dx = 0.72 m. With these values we get sy max = 0.0741 S/m. These were the values used for this simulation. The main objective of this gradual increase of conductivity, as Berenger specified, is to avoid reflections from the absorbing layer back into the grid, and at the same time attenuate as much as possible all the waves going into the PML. To simplify the code, we will use the same above update equations for all i and j, defining sx|i = sy |j = 0, sx

*|i+1/2 = sy

*|j+1/2 = 0 and e|i,j = e0, along the hallway, and using the expression above in the PML regions, with e|i,j = e0 in the rest of the space. In order to implement the perfectly conductive walls, we just force the scattered field to be the inverse of the incident field at the walls locations, so we do not have to use sx|i = sy |j =?, which is computationally impossible.

Fig.14. Recorded Electric Field magnitude at the receiving points Rxn, n = 16, located at 20 ft, 30 ft, 40 ft, 50 ft, 60 ft, 70 ft respectively.

We also should mention that, to meet the PML conditions,

in the regions where sx|I, sy |j, sx*|i+1/2 and sy

*|j+1/2? are different from zero, they must fulfill the Berenger conditions for zero reflection [16]:

yyxx sem

ssem

s0

0*

0

0* == (2.6)

Figure 14 shows plots of the total field distribution in time for the hallway, at the locations shown in Figure 13. We must say that the total field is calculated adding the incident field and the scattered field for the region y < 10 cm, where we only have scattered fields. These calculations can then be used to

estimate the measured channels parameters, using the same method we will see in section III.

1) Path Loss

The maximum received electric field magnitude squared, |Ez|2

(the one corresponding to the main path) at each recorded location Rxn is giving us a relative idea of the received power Pr magnitude at each location. We are expecting to have a |Ez|

2 decreasing with distance d as (This expression is similar to (3.1.), used in next section):

|Ez|2(dB) = Ko (dB) n*10log10 d (2.7)

Where n is the exponent of the power-distance relationship and Ko is a constant set by the amplitude of the transmitted pulse. The results of this calculation are shown in Figure 15.

Fig 15. Path loss values obtained from simulat ion. The vertical scale is dB(V/m)

2, not a strict power measure, except for a scale factor, Ko, assuming that the medium is isotropic.

2) Delay Spread

The calculation of the delay spread will be specified in the next section, but we will present some results here from the simulation, therefore it is good to keep in mind that section as reference. Using expressions (3.4.) and (3.5.), we can obtain a value of the rms delay spread, sS, as defined in [3]. Note that these two expressions use the received power profile, which for the simulation was replaced by the electric field square magnitude as a function of sampled time tn, |Ez|

2(tn), the one shown in Figure 14. The results for our simulation are shown in Figure 16.

3) Coherence Bandwidth

The results in time shown in Figure 14 can be used to estimate the Coherence Bandwidth, BC. Performing a Fourier Transform of this data, it is possible to obtain an estimate of BC. Using as reference the correlation method described in next section to calculate this parameter from the frequency spectrum of the signal, we get the values shown in Figure 17.


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Fig.16. Obtained values of rms Delay Spread as a function of distance from the simulated FDTD model.

Fig. 17 Obtained values of Coherence Bandwidth as a function of distance from the simulated FDTD model.

III. CHANNEL PARAMETERS

A. Path Loss

A short duration pulse, with nanosecond duration, is used by the VNA to probe the propagation channel. The duration of the single pulse, inversely proportional to the bandwidth of the transmission, determines the multipath resolution, i.e., the minimum discernible path between individual multipath components. The period of the periodic pulse signal transmission determines the maximum measurable multipath delay. Hence, successive multipath components with differential delay greater than the width of the pulse and within one period of the periodic pulse transmission can be measured unambiguously. The single multipath profile of 300 ns mentioned in section II.A and showed in Figure 4 can capture two successive probing pulses which is enough for the purposes of the measurement. Figure 4 also shows that the response of the first probing pulse has decayed before the next pulse arrives at the receive antenna which clearly indicates the presence of path loss.

Fig.18. Path loss exponent for different antenna configurations.

The received power for each frequency of interest at each

point from the transmit antenna was extracted by averaging the 4 equally-spaced (by a fraction of wavelength) frequency response data sets for each position. As we know, received power (Pr) decreases with the distance (d) as

Pr = Po.d-n or Pr(dB) = Po (dB) n*10log10 d (3.1)

where n is the exponent of the power-distance relationship and Po is a constant set by transmitted power and measured system gain. n is also known as the path loss exponent.

In this project, the values of n were extracted from the slope of the line that better fits the measured data of receiver power with distance, both in logarithmic (dB) scale. Figure 18 presents a comparison of the path loss for the four


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measurement setup for the indoor environment at 2.45 GHz and shows the reasonable well-fit log-log line with the respective values of drop-off exponent.

B. Ricean K-factor

Fast fading or small scale fading in the signal is the result of small variation in the spatial separation between the receiver and transmitter which generates depth changes in the signal amplitude and phase. In case a dominant Line-Of-Sight (LOS) component is present, the amplitude distribution of fast fading is characterized by the Ricean probability distribution function (pdf). Otherwise, the Rayleigh pdf is the best characterization of that small scale fading. In an indoor environment, a LOS path between transmit and receive antenna is expected. The Ricean K-factor is defined as the ratio between the LOS signal power and the power of the multipath component [3]. The Ricean pdf is given as [3]

0.7

0

5

10

15

20

25

30

35

0 10 20 30 40 50 60 70Distance (feet)

Co

her

ence

Ban

dw

idth

(MH

z)

Dipoles Dipole-Yagi

Dipole-Yagi-Cross Dipole-Helix

Corr > 0.9

0

2

4

6

8

10

12

0 20 40 60 80Distance (feet)

Co

her

ence

Ban

dw

idth

(M

Hz)

corr > 0.5

0

10

20

30

40

50

0 20 40 60Distance (feet)

Co

her

ence

Ban

dw

idth

(M

Hz)

Fig. 22. Coherence Bandwidth vs. Distance for a 0.7, 0.9 and 0.5 correlation threshold.

Table III summarizes the measurement results under the

different setups. Depending on the configuration and on the correlation threshold, we estimated the Coherence Bandwidth of the channel to be between 3 and 27 MHz for our experiments. Under a more conservative stand we would estimate that the Coherence Bandwidth is between 3 and 7 MHz. Thus, the general believe that Bc = 1/ss is off by a factor of ten in our measurements, since we estimated the delay spread to around 8nsec. Based on our results, this relationship would be closer to Bc 1/(10ss).


11

0

0.2

0.4

0.6

0.8

1

1.2

0 10 20 30 40X=Coherence Bandwidth

Pro

bab

ility

x 0.5)

IV. CONCLUSIONS

Results confirm the impact of the different antenna configurations on the channel parameters. Antenna directivity improves multipath characteristics of the wireless channel. Circularly polarized antennas additionally show more stable performance since there is no dependence on the relative angle between transmit and receive antennas. In general, as received power increases the channel performs better showing higher coherence bandwidth and lower delay spread values.

The path loss factor is found to be lower (n = 1.7-1.8) than the free space attenuation in a hallway environment. However, comparing with the results of the simulation model, the characteristics of the hallway differ from the ideal conductive waveguide.

Dominant Line-Of-Sight path is expected in indoor environments like hallways. The mean value of the Ricean K-factor is approximately independent of distance and varies from 16 dB to 30 dB for different antenna configurations. These high values confirm Ricean Fading characteristics.

Comparing the measured results of Delay Spread and Coherence Bandwidth, we find that the relationship between Bc and s s is not exactly inverse but BC 1/(10 s s). This is also found in the numerical simulations, even thought the values of simulated Delay Spread are higher than the measured ones, and the values of simulated Coherence Bandwidth smaller than the ones measured. Based on our measurements, a conservative estimate of the Coherence Bandwidth of the channel is between 3 and 7 MHz based on the antenna configuration.

The numerical simulation shows that the ideal, perfectly conducting walls is not a faithful representation of the walls in the hallway. This conclusion is derived mainly from the path loss factor obtained for the simulation (n 1), which is the one we would obtain modeling the hall as a waveguide. In this case the energy is concentrated in the y direction, allowing for a higher delay spread, and a decrease of power as a function of distance d, as d-1, instead of d-2, which is usually the case in free space.

REFERENCES [1] T.A. Wysocki and H. Zepernick, Characterization of the Indoor

Radio Propagation Channel at 2.4 GHz , Journal of Telecommunications and Information Technology, 2000, vol. 3, no. 4, pp. 84-90.

[2] J. S. Davis and J. M. Linnartz, Indoor Propagation at 2.4 GHz in Wireless Communications Demo Edition 1997, unpublished.

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ee245 2.4 ghz indoor channel

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