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Effects of Matching on RF Power Amplifier

Efficiency and Output Power

The effects of matching the large-signal input and load impedances of an active device during RF

power amplifier design

Francisco Javier Ortega Gonzalez, Jose Luis Jimenez Martin and Alberto Asensio Lopez

April 1, 1998

Effects of Matching on RF Power Amplifier Efficiency and

Output Power

To date, one of the most important design methods for RF power amplifiers (PA) still consists of

matching the large-signal input and load impedances of an active device. These large-signal

impedances are supplied by the manufacturer of the active devices or are measured directly by the

user. In both cases, the output power level and efficiency are associated with those large-signal

impedances. Nevertheless, in practice, the output power, efficiency and even the operation mode

depend not only on the large-signal impedances but also on the matching networks used to provide

the recommended loads for the transistor. This phenomenon is caused by the high frequency behavior

of the matching networks. The usual practice of using a short-circuitedl/4 line located at the

collector or drain of the transistor to supply DC power causes similar frequency-limiting effects due to

the behavior ofl/4 lines at harmonics of the fundamental frequency.

Francisco Javier Ortega Gonzalez and Jose Luis Jimenez Martin

EUIT TelecomunicacinMadrid, Spain

Alberto Asensio Lpez

ETSI Telecomunicacin (GMR)

Madrid, Spain

The most popular methods for designing RF PAs utilize load-pull techniques in different sophistication levels,

techniques based on load-line theory and techniques based on nonlinear simulation. By far, the load-pull

design techniques are still the most widely used by most engineers and, hence, are the focus of this article.

However, this methods popularity does not mean the other design techniques are not important.

Load-Pull Methods
http://www.microwavejournal.com/authors/519-francisco-javier-ortega-gonzalez-jose-luis-jimenez-martin-and-alberto-asensio-lopez/articleshttp://www.microwavejournal.com/http://www.microwavejournal.com/authors/519-francisco-javier-ortega-gonzalez-jose-luis-jimenez-martin-and-alberto-asensio-lopez/articleshttp://www.microwavejournal.com/


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The design methods of PAs based on load-pull measurements (matching of large-signal impedances) consist

of providing a concrete load impedance at the output of a transistor and simultaneously matching the input

impedance of the device. The transistors large-signal impedances are provided by the manufacturer or are

measured directly by the user. After the large-signal impedances are determined, it is necessary to employ

matching networks to provide these impedances at the input and load of the active device.

In addition, it is important to determine the usefulness of the load and input impedances provided by themanufacturer or in-house measurements. Usually, these impedances are suitable for the operating conditions

for which the transistor was designed. The impedances are selected to maximize features such as gain or

output power. In most cases, other important features such as collector/drain or added efficiencies are

secondary.

Large-signal impedances are typically measured only at the fundamental operating frequency. The data

provided by the manufacturer always are measured at the operating frequency. Only sometimes does the

manufacturer show the test fixture used to measure the large-signal impedances. This test fixture can provide

insight into the high frequency behavior of the loads recommended by the manufacturer.

In most cases, the designer is able to measure the large-signal impedances of a device only at the fundamental

frequency. Although an increasing number of papers about multiharmonic load-pull measurements appear in

technical literature every day,1,2 these works are still a minority. In the near future, this situation most likely

will change because some manufacturers are beginning to offer automatic multiharmonic load-pull

measurement systems. Nevertheless, the cost of these systems is still high. In the end, the output impedance

of the generator and the load must be transformed into the required large-signal impedances at the input and

output of the transistor.

Matching

The problem of determining how to design the matching networks of any amplifier can be complex,

particularly if wideband matching is required. Usually, the operating frequency determines the use of matching

networks using descrete components (capacitors and coils) or transmission lines. During the 1960s and

1970s, various manufacturers published application notes and tables to design matching networks for RF

power amplifiers.3,4 The results obtained for transmission line or descrete matching networks with regard to

the focus of this article are exactly the same. Therefore, this work will concentrate on analyzing matching

networks with descrete components. The results obtained can be extrapolated to any network with the same

behavior in the frequency domain.

In his application note, Davis describes four matching networks consisting of three elements each. 3 The three-

element matching networks are useful for narrowband matching. For the same application, matching networks

different from those described by Davis are necessary to match the large-signal impedances provided by the

manufacturer. However, designers have found that the resulting output power and efficiency can be

substantially different.


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The efficiency and output power of RF PAs are not the only parameters that

depend on the matching network used to match the large-signal impedances

provided for the device. The mode of operation (C or C-E) also can change. Three

of these matching networks, shown in Figure 1 , will be analyzed from this point of

view, emphasizing their high frequency behavior.

Most PA devices use loads and have input large-signal impedances lower than 50W . For this reason, the coil L1 should be attached directly to the collector/drain at

the output or the base/gate at the input of the device. In this sense, network A

begins with a coil. The first thing the transistor sees is a coil, which causes the high

frequency performance of the matching network to be inductive and tending toward

an open circuit. (An exception occurs if the coil exhibits self resonances near the operating frequency.) In this

way, although the correct load can be achieved at the fundamental frequency, a high reactive impedance

tending toward an open circuit is provided for the harmonics of the fundamental frequency.

Network B starts with a capacitor connected to ground. Obviously, the reactance of this capacitor decreases

when the frequency increases. This case is the opposite of the one mentioned previously. At high frequencies,

this network tends toward a short circuit. The high frequency behavior of this network is completely different

from the behavior of network A. Nevertheless, in many cases, this network is able to match the input and

load impedances provided for the device. Network B is recommended to match tube amplifiers.3

Network C begins with a coil. It is different from network A in that it has two coils. In this sense, its high

frequency behavior is similar to but better than the high frequency of network A and is said to achieve high

collector efficiencies.3

Table 1lists the frequency behavior of the three matching networks that are used to provide a proper loadimpedance at the output of a transistor. The quality factor Q of the three networks is 3. The load to be

matched is ZLOAD = 12.5 + j9.

Table I

Matching Network Frequency Behavior

Network A Network B Network C

L1(mH) 8.0 2.2 8.0

L2(mH) -- -- 10.4

C1(pF) 2.7 28.1 6.6


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C2(pF) 3 22 --

Local Impedence (W )

At 900m MHz 12.0 + j9.4 12.0 + j9.0 13.0 + j9.0

At 1800 MHz 8.4 + j68.0 0.01 j3.7 0.8 + j74.0

At 2700 MHz 5.6 +j118.0 0 j2.3 0.2 + j12.5

As can be seen, all the matching networks are able to provide the required load impedance at 900 MHz.

Nevertheless, their high frequency behavior is completely different. Matching networks A and C tend toward

high inductive impedances (open circuit) as the frequency increases. Network B tends toward a low

capacitive impedance (short circuit). This high frequency behavior will influence not only the devices output

power and efficiency performance, but also the class of operation.

Figures 2, 3 and 4 show nonlinear time-domain simulations where the output waveforms of current and

voltage of the same transistor matched with networks A, B and C are displayed. The behavior of the device

loaded at the output is the only case addressed in this article.

Table 2lists the performance obtained with networks A, B and C. It is apparent that the same load

impedance at the fundamental frequency yields very different results.

Table II

Amplifier Performance vs. Output Matching


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Network A Network B Network C

Pout (W) 8.87 4.71 8.20

Pin (W) 1 1 1

Power gain

(dB)

9.4 6.7 9.1

Collector Effic.

(%)

77.0 20.0 88.5

Added Effic.

(%)

71 19 80

Pdc (W) 11.47 24.00 9.25

In network A, the transistor used in the amplifier to perform this work has its base connected to ground.

Many designers may think this configuration is equivalent to working in class C, but class C requires collector

current flowing in less than 180 of the signal period as well as short circuits for harmonics of the fundamental

frequency. The high frequency trend of network A is exactly the opposite of the required behavior for class C

operation. The data show that the collector current and VCE are exactly opposite of what is required for class

C operation. Rather, network A forces the transistor to work in a mixed-C mode5 or class C-E.6 The data

show a very high efficiency for network A as a result of the mixed-C mode or class C-E amplifier

performance.

The high frequency behavior of network B is the opposite of the high frequency behavior of network A. It

offers short circuits at the harmonics of the operating frequency. In this way, the amplifier is expected to

function in a true class C operating mode. The output waveforms of the amplifier are completely different

from regular output waveforms of class C operation. This effect is due mainly to the nonlinear output

capacitance of the device (apart from other considerations).5 Nevertheless, it is believed that the inductance

of the package (bonding) plays a very important role in this effect, particularly at high frequencies. This effect

tends to preclude the C operating mode because the conditions required cannot be achieved. In fact, it isalmost impossible to achieve true class C operation with RF power bipolar transistors. The data show that

the output power and efficiency for these networks are very low.


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The high frequency behavior of network C is similar to but better than the high frequency behavior of network

A (at least at the second harmonic). The output waveforms are also similar to the waveforms obtained with

network A. The operation mode is also mixed C or C-E. Network C exhibits weak inductive behavior for

the third harmonic. However, the resulting collector efficiency is the best of the three configurations. This

result enforces the theory that the value of the load at the second harmonic is crucial for efficiency and the

load at the third harmonic only refines the main performance.8

From these three examples, it is easy to conclude that the high frequency behavior of the matching networkscauses not only important changes in RF PA efficiency and output power, but also that the operation mode

depends on these networks. These results have been obtained for one load impedance only. It is useful to

determine if these results can be reproduced for any load on the Smith chart. Figures 5, 6, 7 and 8 show

load-pull diagrams obtained on the Smith chart that resulted from various loads provided by networks A and

C. Not all of the loads could be matched with network B.

Analyzing these data, it is easy to derive constant output power and constant collector efficiency hc load-pull

contours. The output power contours obtained with networks A and C are very similar. The constant

efficiency contours also are very similar for networks A and C, but some differences in the maximum values

exist. The collector efficiency obtained with network C is slightly better.

These results agree with the values obtained for efficiency and output power. Apparently, matching network

C yields better collector efficiency as predicted.3 This result also confirms that the mixed-C or C-E mode is

similar to class E.6,7 For this kind of amplifier operating in class E, C-E or mixed-C mode (from a frequency

point of view), once the harmonic load impedance is set highly inductive the collector efficiency depends

mainly on the load at the fundamental frequency.8 On the other hand, most of the so-called C amplifiers are

actually E, C-E or mixed-C amplifiers. Many RF amplifiers function in this way, particularly at VHF and


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UHF. The test circuits of the manufacturers only need to be examined to confirm this fact.

The Short-circuited l /4 Transmission Line

Determining how to feed PAs is another routine problem faced by RF PA

designers. l /4 transmission lines are used frequently at high frequencies (for

example, 900 MHz). The transmission line is short circuited at one of its

extremities. This type of line usually is located at the collector/drain of a

transistor. The l /4 lines exhibit infinite impedance at the fundamental frequency

and odd harmonics, and a short circuit at the even harmonics. This behavior at

high frequencies influences the devices collector efficiency and output power.

Figure 9 shows the analyzed PA with a short-circuited l /4 line attached

directly at the collector and at the output of matching network C.

The output waveforms (VCE/VDS and iC /iD ), shown in Figures 10 and 11 , were obtained using a coil to

feed the collector of the transistor. The same output waveforms (VCE/VDS and iC/iD) are displayed whenthe coil is replaced by a l/4 line attached at the collector/drain or output of the matching network. Table 3

lists the performance of the amplifier using a l /4line at the collector to feed the amplifier (location a) or after

the matching network (location b).

Table III

Amplifier Performance vs. Transmission Line Location

Location A Location B

Pout (W) 3.84 8.43

Pin (W) 1 1


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Power gain (dB) 5.8 9.2

Collector Effic. (%) 17 91

Added Efficiency (%) 16 87

Pdc (W) 22.6 9.2

It is apparent that the first location of the l /4 line (at the collector) leads to a situation similar to using

matching network B. This similarity occurs because the l /4 line provides a short circuit at the second

harmonic of the operating frequency. If the l /4 line is located after the matching network (location b), the

high frequency behavior of the matching network predominates over the high frequency behavior of the line.

As a result, the performance is similar to that obtained with matching networks A and C. A similar situation

occurs when the collector coil self resonates at frequencies lower than the third harmonic of the operating

frequency.

The Transistor Package

Most PA designers use packaged transistors. Many packaged power transistors have built-in matching

networks to increase the operating bandwidth of the device. The behavior of built-

in matching networks is similar to that of any matching network. Therefore, built-in

matching networks can determine the operating mode of the final amplifier, limiting

output power and, especially, collector efficiency. In this sense, a packaged

transistor with a built-in matching network most often functions like an amplifier

rather than a transistor. Furthermore, bonding and dynamic and stray capacitances

of the device also limit the performance of the design. Figure 12 shows the area in

which the entire Smith chart is converted when it is seen at different harmonics

through the built-in matching network of a commercial power transistor.

Conclusion

The effects of matching networks on RF PA efficiency and output have been analyzed. The high frequency

behavior of the matching networks influences not only the performance of RF amplifiers, but also their

operating modes. It was determined from the high frequency behavior of the matching networks that the load

at the second harmonic is crucial to improving efficiency. The load at the third and higher harmonics only

contributes to refining the potential high efficiency behavior. As a result, the large-signal impedances offered

by the manufacturers should be viewed with caution. In order to achieve the specified performance for any

device, the design of the amplifier should closely resemble the test circuit recommended by the manufacturer.

If this configuration is not possible, the high frequency behavior of the matching networks at least must

reproduce as closely as possible the high frequency behavior of the networks used in the manufacturers test

fixture (at the input and output of the transistor).

The effects associated with short-circuited l /4 lines used for feeding purposes also have been analyzed. The


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www microwavejournal com/ar ticles/print/2303 effects of matching on rf power amplifi er efficiency and output power 9/9

high frequency behavior of these lines modifies the efficiency and output power of PAs, depending on the

location selected (attached directly to the collector or after the matching network). Again, the reason is due to

the high frequency behavior of the l/4 transmission lines. Similarly, the package of the transistor and built-in

matching network (if it exists) influences the operating mode and performance of the amplifier. Manufacturers

usually offer little, if any, information about this important topic.

This article has analyzed only the problem at the output of the device. The effects of matching networks at the

input of the device will be covered in future work.

Acknowledgment

The authors wish to thank Jose Manuel Ledo for the load-pull simulations described in this article.

References

1. J. Staudinger, "Multiharmonic Load Termination Effects on GaAs MESFET Power Amplifiers,"

Microwave Journal, April 1996, pp. 6077.

2. F. Blache, J.M. Nebus, P. Bouysse and J.P. Villote, " A Novel Computerized Multiharmonic Load-

pull and Power Measurement System," Proceedings of the Microwave Theory and Techniques

Symposium (MTT-S), Orlando, FL, 1995.

3. F. Davis, "Matching Networks Designs with Computer Solutions," Application Note ANA267,

Motorola Semiconductors.

4. R. Hejhall, "Systemizing RF Power Amplifier Design," Application Note AN282A, Motorola

Semiconductors.

5. H.L. Krauss, C.W. Bostian and F.H. Raab, Solid State Radio Engineering, Wiley, New York, NY.

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