feedback vs feedforward common-mode control: a comparative study

8/17/2019 Feedback vs feedforward common-mode control: a comparative study

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Feedback vs feedforward common-mode control:

a comparative study

J.M. Carrillo,

J.L.

Ausin, P. Merchh, and J.F.Duque-Carrillo

Dept. of Electronics and Electr. Eng.

University of Extremadura

(06071) Badajoz, Spain

Tel.-Fax: +34-24-289544;duque@pizarro.

unex.es

Abstract

A

comparison among feedforward (CMFF) and the

traditional common-mode feedback (CMFB) loops,

based on the most frequently used common-mode (CM)

signal detectors for CM control in fully-differential (FD)

circuits,

is

carried out. Simulated results confirm that

CMFF shows a better performance in terms of induced

nonlinear signal distortion, speed, and amplifier output

signal swing. It is demonstrated that feedforward

approach results very attractive for low-voltage

applications.

1.

Introduction

In the last years, fully-differential (FD) signal processors

have been widely used, mainly to take advantage of the

reduced available signal swing imposed by the fast scaling

of CMOS technologies. Besides,

as

compared to the

single-ended counterparts, FD structures provide other

non-negligible advantages such as more reduced harmonic

distortion, higher rejection capability to power-supply and

substrate coupling noises, and higher design flexibility.

However, the main disadvantage consists of the need of an

extra negative feedback (CMFB) loop that controls the

output common-mode (CM) amplifiers voltage, fixing it

to an appropriate dc reference voltage

(V,J,

usually at the

middle value between supplies.

The design of any CMFB loop must be carried out very

carefully to avoid, as much as possible, the interaction

with differential-mode (DM) loops, since, otherwise, the

performance of the F circuit can be degraded

[I].

This

requirement is more and more difficult to fulfill

as

the

total supply voltage is scaling down. Therefore, designing

continuous-time CMFB circuits that are both linear and

operate with low power-supply voltages, is an area of

continuing research.

Very recently some approaches have been reported in

order to avoid the need of CMFB loop in

FD

circuits [2

-

41, however, all the proposed techniques present their own

pros and cons.

In this work, it is shown how the control of the CM

component based on feedforward (CMFF) provides

advantages respect to the traditional CMFB networks. The

CMFF performance is discussed and compared with the

feedback counterpart based on the most frequently used

CM signal detectors (differential pairs, source followers,

and triode-operated MOS), according to three figures of

merit: induced distortion, speed, and output signal swing.

Simulated results to illustrate the amplifier performances

are also shown.

2.

Feedback and feedforward

CM

control

Figure 1 shows a generic FD transconductance amplifier

(enclosed by a Gaussian surface) with CMFB. The CM

signal detector senses the output voltages and, ideally,

provides a voltage V, proportional to the output CM

component,

Vo,c,fi.

his output CM voltage is compared

with the reference voltage in a gain stage, forcing Vo,c,,,o

the desired value V,


2/4

Current current I

I

Fig.

2.

Generalized

FD

transconductance amplifier

with

CMFF.

Io++ I,

1

Fig. 3. Circuit implementation of the feedforward

section with frequency compensation.

summing node voltage to

V,cf

The generated current I is

substracted from the amplifier output currents, performing

a feedforward cancellation of the output CM voltage and

setting

Vocm

o

Vmf

A circuit implementation of CMFF control is shown in

Fig. 3. Basically, it consists of a self-biased two-stage

ampliflier. In order to ensure the stability of the global

feedback loop, the positive input of the differential second

stage is connected to the output of the first stage.

Therefore, as the second stage is non-inverting, a

compensation technique, more robust for this case than the

traditional Miller compensation, is used

[ 5 ]

Frequency

compensation is realized by means of the capacitor C

This compensation capacitor has no effect on the pole

associated with the output of the first stage, but affects the

positions of the second amplifier stage and generates a

zero in the global feedback loop. Assuming that the poles

&-e widely spread, a direct analysis of the equivalent

small-signal circuit of the general feedback loop, leads to

the following expressions for the critical frequencies

(poles and zeros) of the loop gain:

(14

Gl

Cl Cc

z=-

where

go,,

nd go are the small-signal output conductances

associated with the output nodes of the first stage and

error amplifier, respectively,

C,,

and

Cp o

re the parasitic

capacitances of these nodes, and g, o is the

transconductance of the error amplifier. The poles p 2 and

p 3 correspond to the lower and high unity loop gain

frequencies of the local feedback loop around the error

amplifier, respectively. The pole-zero pair pz-z is

generated by the compensation capacitance C,. To

guarantee the stability, the zero is placed relatively close

to the pole p, , since a large mismatch is tolerable while the

phase shift is maintained with enough safety margin.

Notice that the position of the pole p I s independent on RI

and C and therefore, the zero can be realized close to this

pole by proper choice of such parameters. As example, for

gfi0 300

pAN,

C = 1 pF,

RI

= 10

IGR

nd C=

20

pF,

the phase margin of the loop changes just in 4 when

z

moves from lSp, up to Sp,. The gain-bandwidth product

of the general loop is given by

where

g, ,

s the transconductanceof transistor Mi.

3.

Performance comparison

CMFB loops frequently include CM sense circuits based

on differential pairs, source followers, and triode-operated

MOS (Fig. 4). The two first structures provide a voltage

V,, proportional to the amplifier output CM voltage, while

the triode-operated MOS, used to degenerate the amplifier

current sources, provides a current

Z,

Next, both

approaches (CMFB based on the above CM sense circuits

and CMFF) are compared according to several f igures of

merit: induced distortion, speed, and amplifer ouput

swing. All the simulated results have been obtained with

the

FD

amplifier connected in unity-gain DM resistive

feedback configuration, 3-V total supply voltage, and a

biasing current of

40@.

i Induced distortion: In CMFB loops, most of CM sense

circuits provide an output signal which, along with the

ideal voltage or current component proportional to the

amplifier CM output voltage, contains some nonlinear

Vo,d,nerms due to the nonlinear I-V characteristic of the

active devices. This fact generates nonlinear distortion on

the DM signal, which is proportional to the CM detector

nonlinearity NL) and signal swing [l]. This nonlinearity

can be expressed as

364


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J

L

Differential Pair

Source Followers

Triode MOS

Feedforward

C)

NL

8

I,

1 1

1

___

2 B 2 R +

E)z

0

Fig.

3.

CMFB is usually based on differential pair (a), source

followers (b), and triode-operated MOS (c) CM detectors.

Table 1.Nonlinearity for the different approaches.

(3)

where

a

is the coefficient of the linear term of the output

of the CM sense block.

Table

1

shows the second-order nonlinear term for the

CM sense circuits, as well as the nonlinearity of generated

current

I

by CMFF. Figure

5

shows the total harmonic

distortion (THD) induced by the different CM controls in

the output signal of the FD amplifier. The frequency of

the sine-wave differential input signal applied is 1 kHz.

As observed, CM control by feedforward introduces a

lower distortion in the output signal as a consequence of

its linearity and also, due to appreciable voltage swing

does not exists in the summing current node, where the

CM output voltage is indirectly sensed. The dc small-

signal gain of the error amplifier included in the

feedforward scheme is about 40 dB.

2

vs Cis =

a

0,~m+ N L .

,,dm

ii) Speed:

The speed limit in the response of any circuit is

given the maximum achievable gain-bandwidth product

with reasonable phase margin. In CMFB loops, the

dominant pole arises at the amplifier output node, while

secondary poles are in the order of gn/Cp, where C

represents the total parasitic capacitance associated each

0.6

0 4

0.2

OURCE FOLLOWERS

IFFERENTIAL PAIR

INEAREEDFORWARDOS

0.b

'

0 4

'

0.8

'

1.5

'

1

. 6

V i n . dm

Fig.

5.

THD induced distortion on a 1-kHz sine-wave

input signal. THD is proportional to NL see Table 1).

internal nodes. However, unlike the DM loop, CMFB loop

introduces, at least, an extra pole (p due to CM sense

circuit. This pole limits the maximum gain-bandwidth

LGBW,) of the CMFB loop. Table 2 shows these critical

frequencies for the different approaches. The position of

p0 , ,

at high frequencies requires large biasing currents.

This fact increases the dc voltage drop in the devices and

can tradeoff the input linear range of the CM detector (see

also Table

3)

and hence, the minimum supply voltage. In

the case of the proposed feedforward scheme, the

frequency response is basically the frequency response of

a two-stage amplifier, which in general

is

slower than the

one-stage amplifier counterpart. Nonetheless, due to the

absence of the CM sense pole, no tradeoff exists between

the frequency response on the one hand, and the linear

range and minimum supply voltage on the other.

iii)

Output

signal swing:

Feedback based CM control

requires common-mode signal detectors with large input

range, to avoid the operation out of the linear region (slew

Po,,,

c,

c,

2 P lL 2

ifferential

Pair

( C , and

C,,

represent a well-substrate and resistor layer-substrate

parasitic capacitances, respectively, and

C

is the load

capacitance).

Table

2.

Critical frequencies for the different approaches.

365


4/4

zone) and hence, the output voltage becomes

assymmetrical. This is a critical design constraint

especially for low-voltage applications, where taking

advantage of available voltage room is a more and more

demanded design feature. Table

3

shows the limits impose

in the amplifier output swing due to the limited input

linear range of the different CM signal detectors,

as

well

as for the feedforward scheme.As shown, in general large

input linear range

in

CM sense circuits requires large bias

current and in some cases, large device area as well. In the

particular case of CM detectors based on source followers

with passive resistors (Fig. 4(b)) [13 such requirements

can be relaxed by using large resistor values. However, as

stated in Table 2 , besides slowing down the speed of the

CMFB loop, for large resistor values the transistor current

sources can go into the linear region before the CM

detector reaches its slew zone. In this case, the input l inear

range for the source follower CM sense circuit, is

given by the second expression

Vo,dmlmax

Differential

Pair

Table 3. Maximum differential output signal swing

in theFD mplifier.

1 5

V o . d m

OURCE FOLLOWERS

DIFFERENTIAL PAIR

LINEAR MOS

FEEDFORWARD

0 0

- 1

5

-

V i n , d m

Fig. 6. Limited input range of CM detectors generates very

assymetrical output swing. CMFF allows rail-to-rail

symmetrical output signal

in

the amplifier.

shown in Table

3.

The resistor value from which this

effect is the limiting one for the linear range, can be easily

derived, resulting:

On

the contrary, a rail-to-rail DM amplifier output range

is allowed by CMFF without tradeoff between swing and

power-area consumptions. Figure

6

illustrates the

simulated results of the FD amplifier curve transfer

characteristic with the different approaches for CM

control. It shows how the limited input linear range of the

CM detectors, shifts the output CM voltage and generates

very assymetrical amplifier output swing, while

feedforward scheme allows a rail-to-rail variation, as

desired for low supply voltage applications.

4.

Conclusions

The control of the CM voltage in

FD

amplifiers by means

of feedforward, rather than the traditional CM feedback

loops, has been proposed, which results very appropriate

for low-voltage applications. Due to the absence of a

specific CM voltage sense circuit,

CMFF

improves the

induced nonlinear distortion, output signal swing, and, in

some cases, the maximum achievable speed of the control

circuit.

References

[l] J. F. Duque-Carrillo, “Control of the common-mode

component in CMOS continuous-time fully differential

signal processing,” Analog Integr. Circ. and Sip. Proc.,

Kluwer Academic Publishers, vol. 4, pp. 131-140,

September 1993.

[2] A. Wyszynski and R. Schaumann, “Avoiding common-

mode feedback in continuous-time

g,”-C

filters by use of

lossy integrators,”

in

Proc.

IEEE Int.

Symp. Circuits

Syst.,

[3] P. D. Walker and M. M. Green, “An approach to fully

diferential circuit design without common-mode feedback,”

IEEE

T. Circuits and Systems

I,

vol. 43, pp. 752-762,

November 1996.

[4]

P.-H.

Lu, C.-Y. Wu, and M.-K. Tsai ‘The design of fully

diferential CMOS operactional amplifiers

without

extra

common-mode feedback circuits,” Analog Integr. Circ. Sig.

Proc.,

Kluwer Academic Publishers, vol. 4, pp. 173-186,

September 1993.

[5 ] Z.

Y.

Chang and W. Sansen, Low-Noise Wide-Band

Amplijiers in Bipolar and CMOS Technologies,

Kluwer

Academic Publishers, 1991, pp. 116.

vol.

5

pp.

281-284, 1994.

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feedback vs feedforward common-mode control: a comparative study

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