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This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/JSEN.2017.2698263, IEEE SensorsJournal

Sensors-16938-2017.R1 1

Abstract— Wireless monitoring systems are becoming a

popular technology in noninvasive diagnosis. Based on current technological trends, low-power consumption, flexibility, small size, disposability and low-cost are the most important parameters to consider when designing sensors for biomedical applications. Echoing this trend, this paper presents a flexible wearable/implantable biocompatible sensor using wireless passive detection scheme, for future studying of the dynamics of an implanted abdominal mesh. Recording information, about mechanical behavior of the implanted mesh helps in understanding hernia recurrence, among other post-surgical problems. Microfabricated sliding interdigitated gold plates and a circular coil (parallel inductor-capacitor tank, LC) on a flexible substrate constitute the sensor’s main core. The capacitance and in consequence the resonant frequency of the sensor will vary according to the sliding magnitude, because the overlapping area between the interdigitated plates changes on sliding. Thus, the monitored resonant frequency is the sensor response. Different thicknesses Cyclo Olefin Polymer was used as dielectric layer. Phantom was used to the ex-vivo characterization of tissue behavior. The capacitor was tested in a 0 to 600 µm displacement’s range. The measured sensitivity was 0.04 and 0.85 pF/50 µm. in simulated conditions, with a resolution of less than 1 µm. The maximum sensitivity in the frequency range was 410 kHz/ 50 µm. The frequency shift can be detected by an external readout coil up to 20 mm in air and 10 mm in phantom. The experimental results confirm the scalability of the sensor.

Index Terms— Microfabricated flexible sensor, interdigitated parallel plate capacitor (IPPC), microdisplacements, technical feasibility, frequency shift.

I. INTRODUCTION NTERNET of Things (IoT) promises a new era of communication and effective real time data transmission,

bringing a flood of new and innovative applications to biomedical and health related industries [1]. The large number of great possibilities are offered in Welfare, Aging and many

Manuscript received January 9, 2017. N. M. Cerón-Hurtado and J. Aguiló-Llobet are with Microelectronic and

Electronic Systems Department at the Autonomous University of Barcelona, and with the Biomonitoring Group at CIBER – BBN. 08193 Bellaterra, Spain. ([email protected]; [email protected]).

J. Aguiló-Llobet is also with the Integrated Systems Department. Microelectronics Institute of Barcelona, IMB-CNM. 08193 Bellaterra, Spain.

M. H. Zarifi and M. Daneshmand are with Department of Electrical and Computer Engineering at the University of Alberta, Edmonton, Alberta, Canada T6G 2V4 ([email protected]; [email protected]).

others, which justify the importance of IoT-related healthcare applications [2]–[5]. Following this idea, a number of IoT-enabled sensors (stationary, wearable, implantable, or ingestible) has been developed offering a new level of diagnostics, monitoring and medical delivery methods.

While energy-harvesting, battery or wire-powered devices can be the right choice for certain applications, in the biomedical area, the wireless powered sensors offer a major versatility and are crucial in implantable types. Wireless electronics provide mobility to a patient and avoid serious health risks and infections through the skin, especially for implantable sensors. The passive wireless LC-sensors are potential alternatives since they are battery-free and have an "on-demand" source of energy, thus called "zero-power" sensors.

RF-powered sensors can play a crucial role in the IoT-related healthcare [6]–[9]. The technology consists of a transmitter inductive coil and a passive LC- tag which is based on a LC resonant circuit. The wireless passive LC resonant tags have been widely studied mainly for use in hostile environments [10] such as: high temperatures [11], corrosive media [12]–[14] or biomedical applications (especially implantable) [15]–[17]. The low complexity, adaptability and low cost are some of the most striking features of those passive sensors.

The principle of operation for wireless LC sensors is based on the change in capacitance or inductance of the tag which can be transferred to resonant amplitude, frequency or quality factor change in the resonant circuit. A secondary coil with inductive coupling to the tag’s coil enables the hands free measurement of electrical characteristics’ variation in the tag. [17][18]. The simplicity, inexpensiveness and robustness of these sensors in performance and implementation make them very attractive and popular for passive telemetry applications [19][20].

In this paper, we present the design, simulation, fabrication and test of a LC sensor prototype to monitor the strain suffered by a mesh which can be implanted in the abdomen of a patient. The main sensing element is an interdigitated parallel plate capacitor (IPPC) with sliding electrodes. The capacitor consists of a pair of interdigitated parallel gold plates (IPPs), which are deposited on a flexible Cyclo Olefin Polymer (COP) substrate. This material is biocompatible, transparent shows low water absorption and good mechanical properties [21]. The IPPs are overlapping and separated by a dielectric with

Flexible Microdisplacement Sensor for Wearable/Implantable Biomedical Applications

Nathalie M. Cerón-Hurtado, Mohammad H. Zarifi, Senior Member, IEEE, Mojgan Daneshmand, Senior Member, IEEE, Jordi Aguiló-Llobet

I



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thickness between 20 and 188 µm. The measurement results of the fabricated biocompatable

sensor demonstrate the feasibility of the proposed LC-sensor to remotely detect micrometer changes on an implantable mesh.

II. PRINCIPLE OF OPERATION The sensing principle of the wireless LC resonant sensor is

based on the relationship between the resonant frequency shift and the applied strain. The expected mesh deformation will produce a relative displacement of one of the capacitor’s plates in a parallel direction to the surface.

Fig. 1 shows the schema of wireless passive sensing. Ls and Cs, are the inductor and variable capacitor on the tag. This resonant circuit transfers the mechanical strain in a capacitance variation which in turn shifts the circuit’s resonance frequency. Le is the “reader” coil which couples to the tag inductance and enables wirelessly reading of the changes on the tag frequency.

To test the device’s concept, for biomedical application purposes a demonstrator is developed. This demonstrator is designed to monitor non-invasively, the stress/strain of an abdominal mesh that could potentially be applied to patients after hernia surgical corrections. As the abdominal wall behavior is modified by the implanted mesh, a better knowledge of the biomechanical response after the implant could be helpful to further improve the treatment of incisional hernias. As the abdomen is not stress free due to physiological movements, the surgical mesh is subject to strain differences that could be measured with this new device.

III. THEORETICAL APPROXIMATIONS: CAPACITOR RESPONSE The capacitor has parallel plates with total length of LT and

each plate has same number of fingers N, with the length l and width w. The spacing between the fingers is considered as s and the spacing between the plates is d (Fig. 2b).

An IPPC may simply model a parallel arrangement of multiple capacitors whose total capacity is the algebraic sum of their individual capacitances; therefore, the capacity for N finger pairs is given by equation 1:

dA

NNCC rii

0´εε

== (1)

where Ci’ is the ideal capacitance for a parallel plate

capacitor (PPC) of surface A = l * w. The expression of Ci’ above is valid when A is quite large and d relatively small to avoid fringing field at the edge of the electrodes [22]; however the electrical response is not automatically scalable when lowering the dimensions of the structure. If the geometric parameters are reduced, the fringing field effects are no longer negligible. Equation 1 can be used only under the condition of a homogeneous electric field, and this is true in any space other than the edges. As we move toward the edges of the capacitor, the curvature of the electric field lines increases resulting in an additional “fringing capacity” [22].

Thus, in order to obtain a more accurate model for the total capacitance calculations, it is necessary to add to the Ci, the fringing capacitance provided by the transverse and longitudinal edges of each pair of fingers, and the interaction effects with adjacent fingers.

That is, the capacitance may be expressed by the following expression:

( )∑=

++=N

npfninT CCCC

1

(2)

where CT is the total capacitance of the capacitor, Cin the ideal or geometric capacitance for each pair of fingers and Cfn the fringe capacitance. The term Cfn is including effects from their own edges, and the interactions with adjacent fingers. Cp, is the system parasitic capacitance. Cfn will be a large part of CT while Cp will only experiment minor variations, thus it will not be taken into account in the analysis.

Fig. 3 shows a schematic view of the finger pair capacitances in the overlapping condition. Cw represents the contribution of the transverse edges (along w), Cwt,wb the longitudinal edges (along l), and Cint represents the interaction effects due to the adjacent finger pairs. All of them are fringe effects which produce a nonlinearity effect on the displacement and capacitance ratios.

Fig. 1. Representative scheme of wireless passive sensing.

(a)

(b)

Fig. 2. Representative schematic of an interdigitated parallel plate capacitor (IPPC) (acting as the capacitive sensor in the schematic presented in Fig. 1) (a) 2D view and (b) 3D view.



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To make a rough estimate of the capacitance, several authors have proposed different methods, based on conformal mapping method [23]–[25] or empirical or analytical solutions using rational functions [26]–[28], in order to model the nonlinear behavior of the capacitance. This characteristic behavior of IPPCs is linked to its relationship w/s, and the working principle in which each movement produces a change in capacitance. The plates relative displacement x, brings additional complexity to (2). This new variable represents the displacement and the resulting changes in the overlapping area. As a consequence of the shared area variation between the interdigitated plates, the capacitance value is decreasing from a maximum (full overlap) to a minimum value when x = p/2. If we could disregard edge effects, the capacitance will be 0 at a fully misaligned position, therefore the ideal design for IPPCs will have the ratio w/s=1. Since the fringe effects are more important for smaller values of w, a suitable relationship between w and s will produce a maximum variation of capacitance for a given displacement, thus for a resolution and sensitivity defined by the application, the sensor design will be optimized by adjusting this ratio.

IV. SENSOR DESIGN AND FABRICATION These edge effects also add complexity when trying to

establish a standard model for the design of such sensors, which imposes the use of finer simulations to reach better approaches to an optimization of the same model.

In order to have a simulation model allowing adjustments for certain range of displacement, a 2D approximation is used. The electric simulations of the IPPC are made in COMSOLMultiphysics® 4.0 (licensed by the National Center microelectronics-CNM-IMB).

The sensor has to suit the application therefore the geometric and technological parameters must be defined, along with the sensor’s value or variation range. A summary of the design parameters for the interdigitated parallel plate capacitor (IPPC) with sliding plates is presented in Table I.

Due to the limited space imposed by the application, we defined the total length of the device between 2 and 4 cm and width 1 to 1.5 cm, since the first demonstrator will be placed on 5 cm x 7 cm abdominal meshes (See Fig. 8).

The coil design is made according to the regulations for inductive applications established by the Electronic Communications Committee [29]. The coil radios between 5 and 7 mm, wire thickness between 0.25 and 0.5 mm, and different turn ratios, in order to achieve resonant frequencies between 11 and 16 MHz are considered. As reported by Jow et al [30], the power loss within the surrounding tissue can be ignored if the operating frequency is chosen below 20 MHz, which also justifies the choice of operation frequency range.

The fabrication of the IPPCs is carried out in a clean room of the National Microelectronics Centre (CNM-IMB) using the method developed by Illa et al. [21]. In this process, the following materials are used: Cyclo Olefin Polymer - COP (thickness 100 µm) as a substrate to provide good mechanical properties and biocompatibility, and the photoresist AZ5214E to carry out the photolithographic process. Gold metal is used for the interdigitated capacitor plates, which provides biocompatibility and low resistivity. This metal layer is deposited on top of a thin layer of titanium in order to promote adhesion between the metal and the substrate [31]–[33]. For printing purposes, chromium masks are used. The fabrication method for this sensor is simple, reproducible and low cost, resulting in a contribution of competitive advantage towards the final sensor structure.

The Fig. 4 shows the most relevant steps of the photolithographic process. The COP (100 µm) sheets were cut in a 4 inch wafer shape, cleaned by a piranha etch (1:3), H2O2 (45%) - H2SO4 (96%), for 10 min. After rinsing the wafers in deionized water and drying them in a spin dryer, the COP wafers were covered by 1.5 mm of AZ5214E photoresist and baked at 90 ºC for 60 s on a hotplate (See Fig. 4a). Finally, the COP sheets were developed on a wet bench using a commercial developer (See Fig. 4b). Once the lithography step was finished, a combination of Au (100 nm) and Ti (20 nm) were evaporated by e-beam (QCL 800, Wordentec Ltd., United Kingdom) over the photoresist pattern (See Fig. 4c). Afterwards, the lift-off was performed in acetone in order to strip the underlying photoresist (See Fig. 4d).

Fig. 4. Fabrication process scheme of the IPPCs (a) Photolithography process to structure the AZ5214E image reversal photoresist on top the COP sheet (b) wet bench (c) deposition of Ti/Au (20nm/100nm) layer by e-beam evaporation and (d) photoresist removal with acetone.

Fig. 3. The schematic view of the capacitances between two finger pairs in the overlapping condition for an IPPC.

TABLE I CAPACITOR DESIGN PARAMETERS

Symbol Description Value/Range w Finger Width 50 to 300 μm l Finger Length 5000 μm s Space between fingers 100 to 1400 μm N Number of finger pairs Variable d Dielectric Thickness 20 to 188 μm p Pitch size (w + s) Variable LT Total length of Capacitor N*(w+s) Variable x Non-overlap distance between fingers Variable ε Dielectric Constant (COP) 2.3 tc Conductors Thickness 0.2 μm ts Substrate Thickness 100 μm



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The design of the different geometric patterns is made in Clewin® 4.0. Fig. 5 shows the fabricated capacitors.

V. TEST BENCH To perform the electrical characterization of IPPCs, the two

interdigitated plates need to be aligned and in contact with the dielectric layer (keeping d) during the sliding while at the same time, the microdisplacements need to be controlled. The setup consists of a positioning and sliding guide coupled to a micrometer. Also, the capacitor plates are aligned and placed in an envelope of transparent flexible polymer support with open ends as shown at the cross-sectional view in Fig. 6a and the entire configuration in Fig. 6b. The electrical response of the capacitor is measured using a 4284A LCRMeter.

Fig. 7 (a) shows the set up of the entire system. The test bench consists of a three-axis positioner, a polycarbonate support for the sensor (custom designed and manufactured), a portable microscope to verify the shifting in the mover plate, and a PIXE vector network analyzer VNA 1075 NI 8135 National Instruments embedded controller with a frequency range 300 kHz to 8 GHz. The positioner controls the microdisplacements and the distance between coils. The polycarbonate support has two pieces, one is movable with clamps to hold the sliding interdigitated plate; the other one is fixed and provides the guide and the alignment to the other components in the sensor (see Fig.6b). The reading coil is connected to the VNA for measurements. For the ex-vivo test and in order to create the artificial abdominal wall, a Petri dish with phantom is placed between the sensor and the reading coil (Fig. 7c). For the mechanical test, the set up and the LC-Tag are shown in Fig. 8. The silicone encapsulation of the tag has been broken in order to show the interior.

VI. SIMULATION AND EXPERIMENTAL RESULTS

A. Capacitor designed: Study of the fringe contribution in the capacitive response Due to the structure and dimensions of the IPPCs, we

should consider the fringe effects along fingers. Calculated fringe capacitance from the experimental data is shown in Table II. It is expected that the measured capacitance is greater than the ideal capacity. To find the percentage contribution of the fringe capacitance, Ci is calculated by (1) and subtracted from the experimental value. Regarding the separation between fingers, a range between 100 and 1400 µm is considered while the other parameters are kept constant. Ci is the same in all cases because the overlapping area corresponds to 45 finger pairs of equal width and length.

As the ratio w/s becomes smaller, CT increases due to the fringing field effects. When the spacing between fingers increases, the field generated by each finger pair is less disturbed by the field of the adjacent finger pairs. By weakening the interactions between them, prevailing interactions between each pair are isolated. Otherwise, when the spacing between fingers is very small, the result is a considerable decrease in the total effect of the Cf. The closer the fingers are, the more lower the fringe effect becomes. When this happens, the ideal case of two infinite plates can be applied and the capacitance value will be Ci. Note that Cf in all cases studied contributes to over 35% of the total capacitance, which confirms the importance of its contribution to CT.

Fig. 8. Test bench configuration Uniaxial test. (View of the LC-Tag)

Fig. 7. Test bench configuration (a) entire system configuration, (b) LC-Tag set-up and (c) ex-vivo configuration.

(a)

(b)

Fig. 6. (a) IPPC cross-sectional view and (b) Entire configuration

Fig. 5. Photos of the interdigitated plates (a) big IPPs design and (b) small IPPs with an abdominal mesh sample on top.



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The measured capacitance for all the fabricated IPPCs is summarized in Table II. The field distribution obtained by multiphysics simulation of the I100/100 and I100/500 IPPC models is shown in Fig. 9.

In contrast with conventional PPC, the proposed capacitor structure has a higher capacitance variation with the same amount of shared area. Thus, we can adjust its resolution by changing the ratio w/s, therefore this ratio is a design parameter. The pitch size also plays an important role in determining the sensor response.

Due to the simplicity in the assembly of the slider and the capacitor (see Fig. 6), it cannot be guaranteed that the plates are completely in contact with the dielectric. This is evidenced by the differences between the experimental and simulation results. Such differences are attributed to a small air layer between the IPPs and the dielectric. Accordingly, the hypothesis of a second dielectric is taken into account in the simulation model.

B. Capacitor designed: Study of the IPPC sensitivity The correlation between the capacity variation and the

dielectric thickness is one of the determinant parameters of the IPPC sensitivity. Therefore, for this study, the IPPC I100/500 (See Table II) is chosen using different COP-sheet thicknesses (20 to 188 µm) between the plates.

The simulation model has been adjusted according to the hypothesis of a second dielectric, as depicted in the inset of Fig. 10 (a), upper right portion.

To best fit the experimental results, we plot different capacitance vs. air thickness curves obtained by simulation shown in Fig. 10 (b). The setting value is obtained interpolating the experimental value in the simulation curve, thus determining the air thickness.

All the curves in Fig. 10 (b), exhibit a drop in the capacitance value as the air layer thickness increases, this tendency is even more pronounced for the lower the thicknesses.

Fig. 10 (a) shows how the experimental trace has been correlated with the adjusted simulation values. These results are in agreement with the expectations after incorporating the second dielectric hypothesis. The drop in the traces is not linear. When the dielectric thickness increases 5 times, the capacitance decreases only 2 times, therefore the reduction in capacitance is lower in IPPCs. This result confirms once again that the fringe effect has a strong impact on capacitance values, being more evident for lower dielectric thickness. Accordingly, Fig. 10 (a) is the characteristic curve for IPPCs.

The experimental and simulation results are summarized in Table III. The proposed simulation model corresponds with relative errors less than 8%, thus it will be a benchmark in optimizing this type of capacitors.

From Fig. 10 (a), it is clear that in order to increase the sensor sensitivity, the dielectric thickness should be as small as possible [34]. Since the commercial COP with the minor thickness is 20 µm, for practical reasons we use it in this study as a minimum value.

It should be noted that regardless of the presence or absence of air gap between the interdigitated plates junction, it is possible to determine the sensor sensitivity supported by the calibration curves obtained from simulation.

TABLE II EXPERIMENTAL RESULTS

Fixed Parameters

ID Iw/s

s (μm) w/s CExp

(pF) Ci

(pF) Cf

(pF) Cf

(%) I100/Var N: 45 w: 100 μm d: 100 μm L: Variable

I100/100 100 1.00 6.63

4.18

2.49 37.5 I100/200 200 0.50 7.62 3.44 45.1 I100/500 500 0.20 8.32 4.14 49.8 I100/700 700 0.14 8.49 4.31 50.8 I100/1000 1000 0.10 8.66 4.48 51.7 I100/1400 1400 0.07 8.83 4.65 52.7

Fig. 9. Capacitance trace of the experimental results in the overlapping condition, keeping w, l, N and d constant, with an s range between 100 and 1400µm. The distribution of the electric field in the simulation model of two IPPC is also illustrated but partially. The figure shows a small segment of the IPPC.

(a)

(b)

Fig. 10. Comparison between capacitance values in the ideal condition overlap in a I100/500 capacitor featuring 45 fingers and COP varying dielectric thickness between 20 and 188μm (a) experimental and simulation results for dielectric thickness and (b) simulation with air thickness.



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C. Capacitor designed: Finger width influence in the capacitive response Table IV summarizes the simulation and experimental

results for different IPPCs with different w values. In all cases, we work with 45 fingers of equal length and COP 100 µm is used as dielectric. The first three capacitors have the same w/s ratio and the last two the same s value.

Overall, the results show that the capacitance does not vary proportionately with the change in the shared area between interdigitated plates, as it happens in the conventional PPCs. The results indicate once again that edge effects should not be neglected in the IPPCs.

To estimate how much is the contribution of fringe effects; the calculated value Ci is subtracted from the experimental capacitance value. For capacitors, whose ratio w/s is 0.5 (s=2w), I100/200, I200/400 and I300/600, the edges contribution is 3.4, 3.86 and 4.94 pF respectively. These results show that although the Ci increases proportionally with the increasing w (Ci100, 2Ci100, 3Ci100), CT does not increase in the same way. Such nonlinear behavior is due to Cf, its relations increasing with nonlinear coefficients; for instance 1.13Cf100 for the I200/400 and 1.45Cf100 for I300/600.

Table IV gives a comparison of the capacitance values, pitch size and w/s ratio. When both w/s and p increases, CT values also increase. When s and N are kept constant, if w increases CT also increases however, the edge effect contributions do not follow this same trend. Cf for I100/500 has a value of 4.14 pF and 4.05 pF for I200/500. This is because when the area increases, Ci increases much faster than Cf meaning that the fringe capacitance is not strongly dependent on the finger width. This result is also evidenced by Meijs and Fokkema[35] after comparing the numerical results of formulas given by Chang [25], Sakurai [28] and Ruelhi and Brennan [27].

The capacitance value of the IPPCs will depend on how the w and s dimensions are correlated between them and on the length LT that is normally limited by the application.

D. Displacement Detection with designed capacitor The displacement detection defines the working principle of

the LC sensor. Once the initial characteristics are known, we design a demonstrator in which the microdisplacements are performed by means of a micrometer attached to a polycarbonate support. The scheme used for the alignment and plates positioning is shown in Fig. 7.

The experimental results for some IPPCs studied are summarized in Table V, and the experimental traces are observed in Fig. 11.

Table V gives a comparison between the maximum and minimum capacitance for different capacitor designs. The values are obtained experimentally under two conditions, the first one when the fingers are fully overlapped and the second one when they are fully misaligned. The difference between these two capacitances will be ΔCmax. In the design of these capacitors, such difference needs to be as large as possible, because this way the sensor sensitivity increases. ΔC is strongly dependent on the dielectric thickness and the optimal w/s ratio.

The ΔCmax happens when the displacement is equal to (w+s)/2, that is when the sliding plate is completely misaligned from the fixed plate. For example, in the IPPC I200/400, we have:

ΔCmax = Cmax - Cmin = (12.22 - 10.19) pF = 2.03pF However, to calculate the maximum sensitivity of the IPPC,

we have considered the regions with the most abrupt changes and therefore with maximum slope (see Fig. 11). The sensitivity values, for a displacement of 50 µm, are in the range between 0.04 and 0.85pF (see Table V).

As stated earlier, one of the variables that determine the sensor sensitivity is the distance between plates (d). Reducing the dielectric thickness and ensuring a perfect joint between plates (airless among them) allows for maximum sensitivity values to be obtained.

TABLE III RESULTS: DIELECTRIC THICKNESS

Fixed Parameters

d (μm)

w/d CExp (pF)

CSim (pF)

Error (%)

I100/500 N: 45 w: 100 μm s: 500 μm L: 26500 μm

20 5.00 16.61 17.87 7.05 40 2.50 12.34 13.29 7.15 60 1.66 10.12 10.77 6.04 80 1.25 9.19 9.64 4.67

100 1.00 8.32 8.73 4.70 120 0.83 7.76 8.15 4.79 160 0.62 6.90 7.24 4.70

188 0.53 6.60 6.88 4.07

TABLE IV EXPERIMENTAL AND SIMULATION RESULTS

Fixed Parameters

ID (Iw/s) w/s p

(μm) Ci

(pF) CExp (pF)

CSim (pF)

Error (%)

N: 45 d: 100 μm

I100/200 0.5 300 4.18 7.58 7.95 4.6 I200/400 0.5 600 8.36 12.22 12.78 4.4 I300/600 0.5 900 12.54 17.48 18.21 4.0 I100/500 0.2 600 4.18 8.32 8.73 4.7 I200/500 0.4 700 8.36 12.46 13.00 4.2

Fig. 11. Experimental capacitance vs. displacement traces for the IPPCs I200/400 and I300/600.

TABLE V CAPACITANCE/DISPLACEMENT

Fixed Parameters ID Cmax

(pF) Cmin (pF)

ΔCmax (pF)

x (µm)

Sensitivity (pF/50µm)

N: 45 d: 100 μm

I100/200 7.62 7.43 0.19 150 0.04 I100/500 8.32 7.20 1.12 300 0.30 I200/400 12.22 10.19 2.03 300 0.44 I300/600 16.48 11.42 5.06 450 0.85



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Table VI shows the simulation results of IPPC I100/500 under ideal conditions for different dielectric thicknesses of COP. As soon as the ratio w/d begins to decrease, the ΔCmax increases rapidly, even though nonlinear variation is present.

From the results for thicknesses of 100 and 20 µm, the sensitivity in 50 µm of displacement is 0.33 pF for the first one and 6.19 pF for the second one. Such difference is 19 times higher than in the case of a conventional PPC which is only 5 times higher. This shows that the fringe effects induce a nonlinear capacitance behavior for the IPPCs.

To study the fringe effect in a single finger pair (when the slider plate is moving), both the influence of the pair’s edges and the field interactions with neighbored pairs must taken into account. The simulation results for the IPPC I100/500 (under ideal conditions) are limited to a single finger pair contribution. Figure 12 shows the capacitance variation curves, in a displacement range 0 to x=(w+s)/2, with ± 50 µm intervals. In the range of dielectric thickness variation of 20 to 200 µm, the center capacitance varies from 0.69 pF to 0.15 pF at zero displacement, indicating that the amplitude decreases as the dielectric thickness increases, which occurs in a nonlinear relationship, as observed in Fig. 10 (a). As the movable plate is displaced right or left of the center point, the capacitance value decreases to reach a minimum, different from the zero level, corresponding to the Cmin, and its position on the abscissa is p/2. ΔCmax has, in this case, a period of 300 µm and in this position the capacitance values remains between 0.16 pF and 0.14 pF. Yu et al [36] and Kim et al [37] report Cmin as a residual capacitance that is maintained at a level above zero. Moreover, the continuity and symmetry are both confirmed for the capacitance variation principle.

From this study, we can observe how the geometric parameters affect the sensor response, concluding that the

dielectric thickness d, the ratio w/s and the space to be occupied L are the most important parameters to determine the sensitivity and resolution of the proposed sensor. In addition, it should be noted that the fringe capacitance is an effect to consider, especially if scaling-down the design is considered.

E. Device and system characterization We integrate the IPPC elements with an inductive coil to

form a LC resonant circuit which has a characteristic resonant frequency. The wireless operation is achieved through near field electromagnetic coupling between the resonant LC-Tag and an external hand-wound reading coil (see Fig. 7). The reading coil is connected to a VNA with a SMA connector to implement the readout system. To characterize the device, the set up shown in Fig. 7 is used. The LC-Tag components are an IPPC I100/500 and a circular coil BwireN21 (7 mm diameter). The maximum sensing distance is characterized between 15 and 20 mm for the Tags with IPPCs I100/s and I200/s, respectively, nevertheless the prototype provides the feasibility of modifying sensor parameters to increase the sensing distance for the required monitoring specifications.

The scattering parameter measured (S11), once the longitudinal displacement of the IPP starts, is shown in Fig.12 (a) and (b). These frequency responses correspond to two different values of the dielectric thickness, 100 and 20 µm respectively.

The resonant peaks are detected at a distance of 8 mm from a single readout coil in an air environment. The resonant frequency shift of sensors validates the presented design methodology. The expected distance in practical wireless sensing will be 8 to14 mm.

We track the displacement changes from an initial overlapping position by a wireless monitoring resonance peak. Microdisplacements are performed pulling one end of the sliding plate (with the clamps, see Fig. 6) in the maximum displacement range of 0 ≤ x ≤ p. As soon as the plate is displaced, the overlapping area between IPPs is reduced, resulting in the capacitance value reduction, thus the resonance frequency of the circuit increases; this is easily observable by the shift to the right in the curves in Figs. 13 (a) and 13 (b).

The frequency response of the sensor can be represented as shown in Fig. 13 (c). It is noted that the frequency variations for the same displacement are much greater for a dielectric thickness of 20 µm than for a 100 µm. Fig. 12 also shows that by decreasing the dielectric thickness, the capacitance value increases, therefore the initial frequency will be reduced.

When the ratio w/d becomes smaller, the minimums shift further to the right side favoring the sensor sensitivity. For this reason, a 300 µm displacement in the IPPC I100/500 (d = 100 µm) generate a Δf of 1.37 MHz and a sensitivity of ≈ 230 kHz/50µm. In the case of the capacitor with a dielectric of 20 µm, the Δf is 2.45 MHz and the sensitivity is ≈ 410 kHz/50µm.

We can conclude that as the ratio d/w decreases, the same displacement causes a greater variation in the frequency Δf, and the sensor sensitivity increases.

Fig. 12. Capacitance vs. displacement curves obtained by simulation of IPPC I100/500 with different dielectric thicknesses normalized to a single finger pair.

TABLE VI CAPACITANCE/DISPLACEMENT

Fixed Parameters d (μm) w/d ΔCmax,

300µm (pF) Sensitivity (pF/50µm)

I100/500 N: 45 w:100 μm s: 500 μm L: 26500 μm (w+s)/2:300 μm

20 0.2 23.50 6.19 40 0.4 11.34 2.16 60 0.6 7.07 1.08 80

100 120 160

0.8 1.0 1.2 1.6

4.75 3.49 2.35 1.28

0.64 0.33 0.24 0.13

200 2.0 0.60 0.04



Sensors-16938-2017.R1 8

F. Ex-vivo characterization Due to the application characteristics such as the

implantable abdominal mesh, a sensor characterization in an ex-vivo model is required. In order to demonstrate the ex-vivo wireless longitudinal strain sensing feasibility of such a flexible LC sensor, the testing is conducted with an experimental set up as shown in Fig. 7 (c). To simulate the "implanted" condition, the sensor is placed under 3 and 6 mm liquid phantom (3.90g / L NaCl and 1.96 g / L CuSO4) used as simulator tissue. The reader coil is placed at the same distance as the volume level of the phantom. The dynamic characteristics of the pair LC-Tag/mesh, a bending/stretching cyclic study is carried on using the set up shown in Fig. 8.

On-bench experimental studies indicate that the LC-Tag sensor - reader coil coupling is substantially affected by the

surrounding medium. Fig. 14 shows the frequency response of the sensor (I100/500 + Bwire21) characterized in two mediums, air and phantom. The sliding IPP is placed in two positions, x=0 and x=(w+s)/2, the second one corresponds to the misaligned position; in this case the distance is 300 µm.

Comparisons between the S11 values obtained with air and phantom in two levels, 3 and 6 mm, show that proximity to the phantom creates some mismatch decreasing the sensing distance.

The most relevant and significant result in this study is the correlation between capacitance and frequency response, showing a dependence on the micrometer displacement between IPPs. The experimental results are placed in a three-axis graph as shown in Fig. 15.

The overlapping regions are highlighted with shaded rectangles. These regions are determined by the geometric characteristics of the capacitors. Fig. 15 represents the sensor response I100/500 + BwireN21 using COP 100 µm as dielectric. The overall sensor response describes a harmonic behavior, akin to the electrical response of the capacitor. When the two responses overlap, direct correspondence can be obtained between electrical variables and displacements.

Monitoring the S11-parameter at the mechanical cyclic-tests shows the reliable and repetitive response; however, the dynamic characteristics are strongly related to the characteristics of the external packaging adopted by the specific application.

Based on these results, the longitudinal strain in an abdominal mesh can be modeled as follows

Fig. 14. I100/500-BwireN21 sensor response in frequency range in air and phantom media.

(a)

(b)

(c)

Fig. 13. Measured S11 parameter versus frequency response for an IPPC I100/500 and the coil BwireN21. The displacement of the sliding plate is 300 μm while two dielectric thickness slides are used (a) 100 µm and (b) 20 µm. (c) frequency traces of the measured sensor response at two dielectric thicknesses (20 and 100 µm).

Fig. 15. The relationship between the electric response of the capacitor and the frequency response for the LC sensor I100/500-BwireN21 for a longitudinal displacement of 600 µm.



Sensors-16938-2017.R1 9

( )( )

( )

2,

-

21

-21

00

swxifCC

C

CL

CCLxf

xf

ss

s

ss

sssx

+≤∆

∆=

∆=

=∆∆

π

π

(3) where ΔCs is the decreased electrical capacitance from the

IPPC by increased longitudinal strain applied on the sensor. As the measuring range is designed to be 10 mm, and the

dynamic range is defined as a ratio of the maximum measurable displacement to the resolution of the sensor, the dynamic range of the LC-Tag is over 1x104.

We demonstrate the ability of this system to track micro displacements with a wireless sensor through simulated tissue. In addition, the measurements in real time are possible thanks to the reading coil. However, future work should be focused on reader optimization. The starting point will be to increase the coupling coefficient between the coils. Some loss-compensation method would be required for the final telemetric system.

VII. CONCLUSION This study presents several contributions. From a

conceptual perspective, the proposed IPPC design is enabled via a simulated model that allows the capacitance values’ adjustment in accordance with the hypothesis of introducing a second dielectric for air gap influence consideration. The greatest deviation between electromagnetic simulations and measured capacitance value is observed with an error < 8%. The advantage of this type of capacitor design over conventional PPCs is that highest capacitance values can be obtained with the same area, in addition to high capacitance changes once a displacement occurs. The optimal w/s ratio for the IPPCs designed is 0.28.

From a biomedical perspective, an ex-vivo testing with this system is demonstrated for real-time micro displacement monitoring through a simulated tissue. The LC-Tag positioned under the phantom liquid, wirelessly detects the electrical response with a reading coil. The initial resonant frequency in the ex-vivo test is shifted due to the saline interface. The reduction in the coupling factor is expected due to the dielectric constant and loss tangent of the saline phantom as compared with air environment. The maximum detection distance is 20 mm in air and 10 mm in phantom for the tags with IPPCs I200/s. The LC-Tag response in the dynamic test of bending/stretching cycles describes the same repetitive harmonic behavior. The frequency results show that the sensor sensitivity increases when d/w ratio in the IPPC design decreases. The improvement of the coupling and the detection distance should be investigated in future research.

From an application perspective, the proposed LC sensor design and its wireless passive scheme detection emerge as an alternative for potential study of the dynamics of an implanted abdominal mesh while showing technical feasibility. The obtained results point to many possible uses of this kind of

sensors for continuous monitoring where miniscule displacement measures are required. Other applications could emerge from magnitudes that can be related to microdisplacements such as pressure, weight and forces in general, respiratory rate with a thoracic band pulse rate, etc.

ACKNOWLEDGMENT The authors would like gratefully acknowledge Xavi Illa,

for the sensor’s fabrication, Sergi Sanchez and Jordi Sacristán of the IMB-CNM–Spain, for their valuable technical input and assistance. We would also like to express our gratitude to Juan Manuel Bellón, Gemma Pascual and Barbara Pérez for the CIBER Project ABDOMESH. We are also immensely grateful to Estefania Peña and Begoña Perez for their support during the mechanical tests, at the Unizar-Spain. The authors also acknowledge the UAB-Grant awarded to Nathalie Cerón.

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Nathalie Marcela Cerón Hurtado received the B.Sc. degree in Engineering Physics from University of Cauca, Colombia in 2008, the first M.Sc. degree in Engineering from Balseiro Institute (IB) - National University of Cuyo (UNCuyo), Argentine in 2010, and a second M.Sc. degree in Micro and Nanoelectronics Engineering from Autonomous University of Barcelona (UAB), Spain in 2012. From 2011 to 2015, she was a Trainee Research Staff (PIF), and from 2015-2016, research assistant support at the UAB. She is currently pursuing the Ph.D. degree in Microelectronic and Electronic Systems at UAB, and is research assistant at Biomedical Applications Group - Bioengineering program of the CIBER-BBN. Her research focus includes sensor design for biomedical applications and biomonitoring, healthcare applications of the Internet of Things, passive-RF for wireless communication systems and gas sensors. Mohammad Hossein Zarifi, received the B.Sc., M.Sc. and Ph.D. degree in electrical and computer engineering from the University of Tabriz, Iran in 2004, 2006 and 2009 respectively. He is currently a postdoctoral fellow at the University of Alberta, Canada. His research focus includes design of high-speed and low-power analog circuits, analog-to-digital converters for biomedical and communication applications and microwave planar structures for sensing application. Dr. Zarifi received CMC-NRC first place award, on industrial collaboration, for the innovative microwave sensors, in Canada. He is a senior member of the IEEE Solid-State Circuits Society, and the IEEE Circuits and Systems Society and served as a reviewer for different journal and conferences. His current research is investigating the new emerging technologies for the state of the art sensors, with focus on microwave resonators. Mojgan Daneshmand (Ph.D, P.Eng) received the B.Sc degree from Iran University of Science and Technology (IUST), Tehran, Iran in 1999, the M.Sc degree from the University of Manitoba, Winnipeg, Canada in 2001 and Ph.D degree from the University of Waterloo in 2005 all in Electrical Engineering. For 2006-2008, she has been awarded Natural Sciences and Engineering Research Council of Canada (NSERC) and also Canadian Space Agency (CSA) Postdoctoral fellowship. She is currently an associate professor at the University of Alberta holding Canada Research Chair Tier II position on Radio Frequency Nano/Micro -systems for communication and sensing. She is working towards advancing Microwave to Millimeter-wave (M2M) lab at the University of Alberta with the focus on Microwave to Millimeter-wave device characterization, design and fabrication. She has been authored and coauthored more than 80 scientific publications including book chapters, accredited journal papers and several patents. She is co-chairing the IEEE Northern Canada Joint APS/MTTS Chapter. She is reviewer for various accredited journals such as Transactions on Microwave Theory and Technique, Journal of Microelectromechanical Systems, and Transactions on Antenna and Propagation. Dr. Daneshmand is also associate editor of IEEE Canadian Journal of Electrical and Computer Engineering. Jordi Aguiló LLobet received the B. Sc and Ph.D degree in Physics Sciences from the Autonomous University of Barcelona (UAB) in 1972 and 1976 respectively. He is currently a coordinator of the Biomedical Applications Group (GAB) at the IMB-CNM (CSIC), the Bioengineering program of the CIBER-BBN and the postgraduate studies at the Microelectronic Systems Department at the UAB. He has been Full Professor at the Engineering School of the UAB since 1987. His research interests are in the fields of Micro and Nano Systems with special focus on Biomedical Applications such as Advanced Micro-Multisensors, Microelectrodes, Neural interfaces, Telemetry systems and Monitoring devices. He has participated as a researcher in numerous European projects. Among them, he led the MicroCard and MicroTrans projects within the EU-IST programme on medical applications for cardiovascular surgery and graft transplants. He participated in the AAL Chronic and Persona projects. He was one of the five representatives of EU institutions related to technology transfer in the EXIF project led by CERN. He participated as expert in the European initiative on Converging Technologies for Food: Nano- Bio-Info and Cognitive Sciences, the EPoSS, scientific committee for medical applications and the External Advisory Group of the FET Flagship Mid-term Pilots. Concerning industrial R&D he led the southern office of MEDICS, the European centre of Competence on Microsystem’s Biomedical Applications.

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