hid lamp electronic ballast based on chopper converter

8/21/2019 HID LAMP Electronic Ballast Based on Chopper Converter

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 59, NO. 4, APRIL 2012 1799

HID Lamp Electronic Ballast Basedon Chopper Converters

Fabio Luis Tomm, Member, IEEE , Álysson Raniere Seidel, Member, IEEE , Alexandre Campos, Member, IEEE ,

Marco A. Dalla Costa, Member, IEEE , and Ricardo Nederson do Prado, Member, IEEE

Abstract—This paper proposes some topologies for developingelectronic ballasts, supplying high-intensity discharge lamps fedby a pulsewidth-modulation ac–ac converter, implemented withbidirectional switches. The lamp operates directly from the acmains; thus, operation with low frequency so as to prevent theoccurrence of destructive acoustic resonance is provided. Thefeatures of the proposed solution are high efficiency, high powerfactor, low cost, and the absence of electrolytic capacitors. This pa-per includes the design of passive elements, the transfer function,and the development of the control strategy. The experimentalresults qualify the viability of the system feasibility.

Index Terms—AC–AC conversion, acoustic resonance (AR)phenomenon, ballasts, current control, high-intensity discharge(HID), ignition, lamps.

I. INTRODUCTION

H IGH-INTENSITY DISCHARGE (HID) lamp lightingsystems are widely used to provide a high-intensity lightlevel for outdoor usages such as vehicle headlight, industrial,and street lighting. These lamps are cheap, their average lifespans are more than 26 000 h long, and they produce light withefficiency higher than 90 lm/W, which make them indispensable

for the aforementioned usages. Traditionally, magnetic ballastswere the usual choice available for HID lighting systems, butthey use bulky and heavy line-frequency inductors to limit thelamp current and to provide ignition. Electromagnetic ballastspresent low cost and are simple and reliable [1]. However,they have several drawbacks, including large size and weight,low power factor, low efficiency, poor power regulation, andsensitivity to line voltage sags. Nowadays, high-efficiencyelectronic HID ballasts are also available and can provide anincreased lighting quality. At the same time, a reduction of the aforementioned drawbacks of magnetic ballasts can beachieved. Unfortunately, electronic ballasts still have higher

Manuscript received September 30, 2010; revised December 23, 2010and February 8, 2011; accepted February 25, 2011. Date of publicationMarch 28, 2011; date of current version November 1, 2011. This work wassupported by the National Council for Scientific and Technological Develop-ment (CNPq) under Proclamation 479046/08-5 and Pró-Publicacões Interna-cionais/PRPGP/UFSM.

F. L. Tomm is with the Federal University of Pampa (UNIPAMPA), Bagé96413-170, Brazil (e-mail: [email protected]).

Á. R. Seidel, A. Campos, M. A. Dalla Costa, and R. N. do Prado are withthe Electronic Ballast Research Group (GEDRE), Federal University of SantaMaria (UFSM), Santa Maria 97105-900, Brazil (e-mail: [email protected];[email protected]; [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2011.2134061

Fig. 1. Functional block diagram of classic LFSW electronic ballasts.

Fig. 2. Functional block diagram of ac–ac electronic ballasts.

cost and lower reliability, mainly due to the use of electrolyticcapacitors [2].

The conventional way of supplying HID lamps by meansof electronic ballasts is applying a low-frequency square-waveform (LFSW) current in order to avoid the acoustic res-onance (AR) phenomenon [3], [4]. These electronic ballastsdemand three power stages: power factor correction, lamppower control provided by a dc–dc converter, and inverterstage. Moreover, any of these stages also require a driver,which increases the overall cost of the circuit [5]–[8]. Fig. 1shows the block diagram of classic LFSW electronic ballasts.Therefore, many efforts are being made in order to integrate

stages [9], [10]. However, it is well known that the integrationof stages results in voltage or current stress in the shared switch,and these circuits still need the presence of large electrolyticcapacitors.

This paper brings a solution to this and other problemsby feeding the lamp with an ac line-frequency sine-waveformcurrent [11] in order to maintain the stability and maintain thepower ripple below the AR threshold requirement. The switch-ing frequency is still maintained high in order to guarantee theefficiency and to minimize the converter magnetic elements.

The proposed solution to achieve this electronic ballast isby using an ac–ac converter, as shown in Fig. 2 [12]. Thetopologies which have the potential to achieve this goal are

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1800 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 59, NO. 4, APRIL 2012

the buck, buck–boost, SEPIC, zeta, and Ćuk converters. Withthe exception of the buck converter for this application, all of them allow reaching a high power factor and a reduction on thenumber of active switches to only two. Moreover, electrolyticcapacitors are not necessary in this application.

The buck ac–ac converter was applied to supply a 250-W

high-pressure sodium (HPS) lamp in [13]. However, this elec-tronic ballast supplies the lamp with an ac line-frequencywaveform without filtering the high-frequency (HF) switchingcomponent. According to Dalla Costa et al. [3], power ripplesabove 5% can cause AR in the lamp. Therefore, the maincontribution of this work is the analysis of several topologiesto achieve ac–ac electronic ballasts, which supply the lampwith the ac line-frequency current sine waveform without thepresence of the HF switching component.

The drawback of such a solution is that, when powering thelamp at low frequency, the lamp impedance becomes nonlinear.Therefore, some characteristics of the electromagnetic supplyare present, like reignition process each half line cycle and lightflicker at double the line frequency. Moreover, due to the factthat the proposed topologies do not present electrolytic capac-itors, which are energy-storage elements, the line variationsreach the lamp. Therefore, the control feedback must be fastenough to compensate such variations.

Experimental results for an HPS 70-W lamp powered fromthe mains (220 V ± 20% at 60 Hz) are presented to validatethe proposed ballast. The converter was designed to operate at33 kHz and feeding the lamp at 60 Hz, using only two con-trolled power switches.

The controller is implemented with a 32-b/38 MIPS micro-controller (ColdFire Microcontroller V1), which contributes to

reducing the number of parts and the system’s final cost.This paper is organized as follows. Section II presents the

basic concepts of the proposed solution. Section III analyzesthe operation modes. A SEPIC ac–ac test topology is presentedin Section IV. The control design is shown in Section V. Theexperimental results are shown in Section VI. This paper isconcluded in Section VII.

II. BASIC CONCEPTS

This section presents the basic concepts required to providethe correct supplying of the HID lamp proposed in this pa-

per. These concepts are related to the AR phenomenon, theproposed methods to avoid it, and the lamp operating stages(ignition, warm-up, and steady state), and a review about ac–acconverter systems supplying HID lamps is performed.

The lamp discharge tube is a mechanical element thatpresents natural oscillation frequencies. In the case of HIDlamps, these frequencies start at 1 kHz and do not presenta theoretical upper limit. However, it has been proved thatextrahigh-frequency (above 1 MHz) electronic ballasts avoidthe AR rising [4].

The AR occurrence is mostly determined by the fact that oneor more power harmonic frequencies produced by the ballastmatch the resonance frequencies of the lamp gas pressure oscil-

lations, which produce distortions in the discharge patterns. Theperiodic lamp power fluctuation results in a gas pressure ripple

Fig. 3. Igniter circuit.

of the same frequency. If this frequency is equal to the naturaloscillation frequency of the particular discharge tube, standingwaves are generated. These waves can lead to light flicker, arcextinction, lamp life decrease, color shift, or discharge tubedestruction [3].

Several methods have been used to try to prevent the ARoccurrence. A revision of the main methods currently knownto avoid the resonance phenomenon is presented in [1] and [4].Electromagnetic ballasts work at mains frequency, preventingthe AR rising. This means that it is a good method to avoidthe detrimental effects of the AR phenomenon. However, thissolution presents several drawbacks, including large size andweight, low power factor, low efficiency, poor power regulation,and sensitivity to line voltage sags.

The use of ac–ac converters is a solution to solve the elec-tromagnetic ballast problems because it works at HF. Someexamples are presented in [13] and [15], where the buck ac–acconverter was employed to supply an HPS lamp. In this paper,

this solution will be extended to the buck–boost, SEPIC, Ćuk,and zeta converters. These topologies demand four-quadrantactive switches due to that the input voltage and output currentin the lamp must be ac sinusoidal. The phase difference betweenvoltage input and output current is near to 0◦ or 180◦ in theseconverters.

The electronic transformers formed by the buck, Ćuk, andzeta converters have their outputs connected directly to thelamp by an inductor and a capacitor. This arrangement makesthese converters capable of implementing a resonant ignition,thus making the converter work in the output filter resonancefrequency, creating the voltage output required for ignition [13],

[16]. However, in order to provide a comparison among allstudied topologies, an external igniter was used, which is shownin Fig. 3.

In the next sections, the proposed strategy to supply the lampduring all operational stages (ignition, warm-up, and steadystate) provided by the proposed topologies is presented.

A. Lamp Ignition

Previous to ignition, the converter must charge rapidly thecapacitor (Ci), through the resistor (Ri), to reach the SIDAC(silicon bilateral voltage-triggered switch) breakdown voltage.During the SIDAC triggered instant, the capacitor voltage is

applied to the transformer primary winding, which applies1.8–2.5 kV at the HPS lamp, through its secondary winding.


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TOMM et al.: HID LAMP ELECTRONIC BALLAST BASED ON CHOPPER CONVERTERS 1801

Fig. 4. Control phases [17].

Fig. 5. Quadrilateral diagram for a 70-W HPS lamp.

The energy stored in Ci is discharged on the lamp. The SIDACsemiconductor returns to blocking state, and the transformerprimary current is null.

B. Lamp Warm-Up

After the ignition, the lamp equivalent impedance drasticallydecreases to a value around 10% of the steady-state impedancevalue. Fig. 4 shows the suggested control modes for each op-eration stage [17]. The proper design of the igniter inductance,discharging the energy accumulated on capacitor Ci, must limitthe current during the ignition in 200% of the rated current.Therefore, there are not excessive electrode detritions at thisstep.

During the lamp heating phase, the ballast keeps the currentbelow twice the nominal current until the lamp power reaches150% of the nominal power. The output voltage should bemonitored so that, when reaching the minimum voltage givenby the standard EN60662 and NBR IEC 662 quadrilateraldiagram in Fig. 5, the power is reduced to nominal. Then, thesteady-state phase takes place.

C. Lamp Steady State

Fig. 5 shows the required characteristics of a lamp supplyduring steady state. During the lamp aging, its operating voltage

is being enhanced by the electrode deteriorations. Furthermore,if the lamp voltage reaches the upper limit, the ballast must beturned off.

In steady state, the lamp current presents a sinusoidalpattern, and the lamp voltage is close to a square waveform.Moreover, during the lamp supply zero crossing, the arc is

extinguished, causing reignition and light flicker. Therefore, theproposed topologies must increase the lamp power deliveringduring the zero crossing in order to minimize the drawbacks of low-frequency ac supply.

Buck–boost, Ćuk, SEPIC, and zeta converters can applyoutput voltages higher than the input voltage. However, the fastcorrection and accurate control of the duty cycle are very dif-ficult in the system design. Experimentally, it was verified thatthe controller must be able to compensate for any disturbanceof the network in less than 0.1 ms in order to avoid perceptiblelight flicker.

III. CIRCUIT OPERATION

The converter operation steps are presented in this section.It is important to denote that all capacitors work at ac volt-age, avoiding the use of electrolytic capacitors. Moreover, theswitch switching frequency is much higher than the ac supplyfrequency. Then, the mains and lamp voltages are consideredconstant during a switching period.

The converter design must pay attention to the dead timebetween the main switches because, if the dead time is too large,overvoltages will occur at these switches.

A. ´ Cuk Converter

The Ćuk chopper is an integration of a boost and a buckconverter. This topology, together with its four working stages,is shown in Fig. 6, where the four-quadrant switches are usedto provide the electronic ballast correct working. Comparingthe ac–ac converter with the conventional dc–dc Ćuk converter,it can be stated that switch S 1 plays the role of the active switchand S 2 plays the role of the diode. This topology was presentedin [1].

When S 1 is in O N state (D · Ts), the source voltage V in(t)is applied to the inductor L1, which can be positive [Fig. 6(a)]

or negative [Fig. 6(c)], depending on the mains half cycle. TheL2 current circulates through S 1 in series with capacitor C 2.The instantaneous voltage on C 2, in steady state, is the sum of the instantaneous input and output voltages.

When S 2 is in ON state [(1 −D) · Ts], L1 dischargesthrough capacitor C 2 [Fig. 6(b) and (d)]. The lamp is in parallelwith capacitor C 1, and the output voltage is applied to inductorL2. A difficulty of this configuration is the C 2 high voltage.However, due to L2, which is in series with the lamp, thevalue of C 2 is lower than those in the SEPIC and buck–boostconverters.

Equation (1) shows the duty cycle used in the ac–ac Ćuk,buck–boost, zeta, and SEPIC converters when they operate with

the inductor current in continuous conduction mode (CCM).The duty cycle value must be updated by the control circuit


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Fig. 6. AC–AC Ćuk converter.

each switching period because it depends on the instantaneousline and lamp voltages

D(t) =

|V out(t)|

|V in(t)| + |V out(t)| . (1)

Fig. 7 shows the theoretical waveforms that can be found inthe ac–ac Ćuk converter in CCM.

B. Buck–Boost Converter

The ac–ac buck–boost converter is shown in Fig. 8, whereL1 and C 2 comprise a filter in order to filter the input currentharmonics. Capacitor C 2 must handle the input voltage peakplus the ripple. Then, L2 represents the buck–boost inductor.

During D · Ts, switch S 1 is turned on, and the voltage of

V C 2 is applied to inductor L2, which can be positive or negativedepending on the mains half cycle. The L2 current circulates

Fig. 7. Ćuk theoretical voltage and current waveforms.

Fig. 8. AC–AC buck–boost converter.

through S 1. This current is the sum of the input and outputcurrents.

When S 2 is in ON state [(1 −D) · Ts], L2 discharges on thecircuit consisting of capacitor C 2, and the load is submitted tothe output voltage.

An inconvenience of this circuit is that the output currentis discontinuous and, because of that, requires a higher output

capacitor than that in [15]. On the other hand, only one largeinductor on the converter is necessary.


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Fig. 9. AC–AC zeta converter.

Fig. 10. AC–AC SEPIC converter.

C. Zeta Converter

The ac–ac zeta converter is shown in Fig. 9, where the inputand output have the same polarity. The zeta converter alsoneeds two inductors and a series capacitor, which is sometimescalled a flying capacitor (C 2). A good characteristic of the zetaconverter is the absence of a right-half-plane zero. This makes

its dynamic transfer function be more easily compensated inorder to achieve a wider loop bandwidth and better load-transient results with smaller output-capacitance values.

During D · Ts, switch S 1 is turned on, and inductors L1 andL2 store energy from the input supply; during this stage, theload is supplied by the C 2 voltage. The C 2 instantaneous volt-age has the same value with the instantaneous output voltageV out(t).

When S 2 is in ON state [(1 −D) · Ts], L1 dischargesthrough C 2, while the energy of inductor L2 is transferred toC 1 and the load.

An inconvenience of this circuit is that the input currentis discontinuous, with a peak value equal to the sum of the

currents of inductors L1 and L2 [8], and thus needs a largerelectromagnetic interference (EMI) filter. Unlike the SEPICconverter, which is configured similar to a standard boostconverter, the zeta converter is configured like a buck converter,where the main switch is in series with the input source.

D. SEPIC Converter

The ac–ac SEPIC converter is shown in Fig. 10, where thevoltage rating of capacitor C 2 is the input voltage peak. Asthe input voltage is sinusoidal, the capacitors must be of dual-polarity ceramic with high lifetime.

DuringD · Ts, switchS 1 is turned on, and the source voltageV in(t) is applied to inductor L1. The L2 current is also con-

TABLE IDESIGN PARAMETERS

ducted by S 1 charging capacitor C 2. The instantaneous voltageon C 2, in steady state, is the instantaneous input voltage.

When S 2 is in ON state [(1 −D) · Ts], L1 discharges on thecircuit consisting of C 2 and the load in parallel with C 1. Theparallel inductor L2 to C 1 is submitted to the output voltage.The L2 current to the output of this converter is discontinuous

and, because of that, requires a high output capacitor, whereasthe input current is continuous, and the use of a large EMI filterto reach the electrical interference standard requirements is notnecessary.

IV. DESIGN EXAMPLE

The ac–ac SEPIC converter was chosen in order to presentthe design example and experimental results. The ballast mustfeed the HPS lamp respecting the quadrilateral diagram inFig. 5, which shows that the minimum lamp steady-state op-erating voltage in 70 W is 76.5 V and the maximum is 130.2 V.

Table I was defined based on the quadrilateral diagram, and itpresents the worst case for not isolated electronic ballast ac–acSEPIC design.

The current ripple in the inductors must be minimized,thus to reduce the capacitor size. The buck–boost and SEPICconverters are the circuits that impose greater stresses at theoutput capacitor because of the output current discontinuity.

The passive elements of the SEPIC converter are given by (2)and (3) based on the values presented in Table I

C 1 = IoutPeak · DAVGα · V outPeak · f

(2)

L1 = L2 = (DAVG · V inPeak)2

IoutPeak · V outPeak · f . (3)

Obtaining these values, the magnetic core required for theinductors is given by (8), shown at the bottom of the nextpage, where K u is the window area utilization rate, which isusually around 0.8. Therefore, the missing parameters in orderto design inductors L1 and L2 are the peak and rms currents,which are shown in (4)–(7), shown at the bottom of the nextpage. The current density J depends on the conductive materialtype applied to the winding, where copper is around 450 A/cm2

from where comes the factor 104 to convert this constant to m2.Bmax is the maximum flux density in the magnetic core without

the saturation effects; it is a value specified by the manufacturer,but for commercial ferrites, it is around 0.25 T.


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TABLE IILIST OF THE COMPONENTS FOR THE SEPIC CONVERTER

Switch S 1 employs insulated-gate bipolar transistor technol-ogy, and switch S 2 employs the CoolMOS technology. It wasperformed in order to improve the converter efficiency becausethe rms current through S 2 is much higher than that in S 1.The SEPIC converter components are listed in Table II. Thesemiconductor S 1 and S 2 drive circuits are optocoupled withvoltage and current amplifiers in order to enable the rapid dutycycle variation [18].

V. CONTROL SYSTEM

The necessary characteristics for the proposed control systemare the following: satisfactory lamp power regulation, gooddisturbance rejection from the supply voltage, and to be fastenough to rapidly compensate the lamp impedance nonlinear-ity. Moreover, the control system was built for maintainingsinusoidal current in closed loop and regards the quadrilateraldiagram (Fig. 5). After arc ignition and stabilization, the lamp

Fig. 11. Control model [20].

effective impedance is roughly equivalent to a resistor with 10%of the equivalent resistance of the normal operation. Moreover,a nonlinearity and small series inductance must be considered.

The proposed digital controller, presented in (13), is obtained

through the analysis of the ac–ac SEPIC converter small-signalmodel, presented in (12). Fig. 11 shows the block diagram of the proposed control system for simulation, and Fig. 12 showsthe proposed control algorithm. The controller transfer functionwas obtained by Routh–Hurwitz analysis and simulated usingthe program Matlab. The ac–ac SEPIC converter small-signalmodel is developed using the dynamic array of the converter,shown in (9), where a is shown in (10)

•iL1•iL2•vC 1•vC 2

=a

iL1iL1vC 1vC 2

+

1L1

0

00

vin (9)

a =

0 0 D−1L1

D−1L1

0 0 1−DL2

−DL2

1−DC 1

D−1C 1

− 1C 1·Z Lamp

01−DC 2

D

C 20 0

. (10)

I L1_Peak = V inPeak · DAVG

2 · L1 · f +

DAVG

1 −DAVG

2V inPeak · IoutPeak

V outPeak(4)

I L2 = V inPeak · DAVG

2 · L2 · f + IoutPeak (5)

I L1rms =

Iout2Peak · D

2AVG

2 · (1 −DAVG)2 +

V out2Peak + D3AVG · V in

2Peak − D

3AVG · V out

2Peak

3 · L21 · f 2

(6)

I L2rms =

Iout2Peak

2 +

V out2Peak + D3AVG · V in

2Peak − D

3AVG · V out

2Peak

3 · L21 · f 2

(7)

Ae · Aw = L_ · I L_Peak · I L_rms · 104K u · J · Bmax

(8)


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Fig. 12. Control strategy for the microcontroller.

The ac–ac SEPIC converter small-signal model, shown in(12), is obtained solving the Taylor series (11) of system (9).Therefore, the obtained equation has as entry the duty cycle,

which is the control variable, and the lamp current as output, asshown in (12) [20]

f (x1, x2, . . . , xn)

= f (x1, x2, . . . , xn) + ∂f

∂x1

x1

(x1 − x1)

+ ∂f

∂x1

x2

(x2 − x2)+ · · ·+ ∂f

∂xn

xn

(xn−xn). (11)

The ac small-signal model is studied by setting ∆D = u,and the plant becomes the array (12) for small perturbations

Fig. 13. Prototype.

around the equilibrium point. It is called the linearized state-space representation by a Jacobian technique

•iL1•iL2•vC 1

•vC 2

= a

iL1iL2vC 1vC 2

+

V inD

L1(1−D)

V inD

L2(1−D)

0V in(D−2D

2)

C 2Z Lamp(D−1)2

u. (12)

This plant model was discretized at twice the switch fre-quency. The average current-mode control was built in orderto get an equation that can be run in a low-cost ColdFire

MCF51QE32 microcontroller, being fast enough to keep thelamp flicker to a minimum during the zero crossing of themains. The block “Gc(z)” performs the compensate calculationof the output current error given by

Gc(z) = 105z + 100

100z2 − 194z + 94. (13)

Regarding the control implementation by the microcontrollershown in Fig. 12, the I Ref input is a sine waveform thatcontrols the output current to maintain it sinusoidal even withthe randomness of the lamp operating at 50/60 Hz as load. Thereference is a table of 1100 values in 12 b multiplied by theK Pow gain. When the input voltage crosses zero, the referencetable is accessed in the first value. Moreover, if the outputvoltage exceeds the maximum of the quadrilateral diagram orthe current exceeds the maximum allowed in the diagram, I Ref is also reduced.

The lamp current sensor is a pulse transformer with a full-wave rectifier with an offset of 1.5 V. Then, a band stop filter

of second order tuned to 800 Hz is added in order to not benecessary to filter the plant natural frequency in the control law.The controller equation in (13), applied to the plant, makes theplant stable in closed loop. The variables behind the driver areinitialized with high values in the block output of “Boot” inFig. 12 in order to minimize the ignition impact on the plant.

VI. EXPERIMENTAL RESULTS AND D ISCUSSIONS

A prototype of the electronic ballast projected in Section IVwas implemented in laboratory and shown in Fig. 13, wherethe igniter and auxiliary source have a double Thornton FerriteCore NEE-25/10/6 IP6. The controller and switch of the auxil-

iary flyback power supply (less than 1 W), with three outputs,were integrated with the ON Semiconductor NCP1010. The


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Fig. 14. Voltage and current on the mains and the lamp.

Fig. 15. Normalized power spectrum on the lamp.

first output is for supplying the control circuit on 3 V regulated.Two isolated outputs are used for driving the switches on 15 Vwith the Hewlett-Packard HCPL3120. The mains voltage isconditioned for the microcontroller by a resistive divider with a

small 10-nF filter capacitor. The dead time is fixed at 150 ns onthe center-aligned 16-b pulsewidth modulation.The experimental results of the SEPIC ac–ac electronic

ballast are shown in Fig. 14. This figure denotes the ballasthigh power factor, where all input current harmonics attendedto the IEC 61000-3-2 class-C requirements. Moreover, it is alsoshown that the ballast supplies the lamp with a low-frequencysine-waveform current without high-order harmonics, avoidingARs and low flicker. The action of the proposed controller canbe verified in the HPS and metal halide lamp waveforms, whereno large reignition voltage is perceived during the mains zerocrossing.

Fig. 15 shows the measurements of the output power spec-trum and confirms that no high-order harmonics are present,which could cause AR at the lamp. Several tests were per-formed, and the input disturbance rejection of 220 V ± 20%is shown in Fig. 16, where the system control feedback com-pensates the output current during the input voltage disturbance.The measured efficiency of the system is about 89.1% in steady-state operation; the power factor is 0.92, and the total harmonicdistortion is about 27.4%.

VII. CONCLUSION

This paper has explored electronic ballasts for HID lamps

based on a direct ac–ac conversion, implemented with four-quadrant switches. A proposed design procedure for the power

Fig. 16. Response for the mains disturbance on the current lamp.

topologies and control for the converters has been presented.The control solution was focused not only on the converterperformance but also on its tailoring to be implemented in a

low-cost microcontroller. Because of the ac–ac converter, thereis no need of a dc bus, and the consequent usual electrolyticcapacitors are avoided, which considerably increases the ballastlifetime [19]. The HF power ripple was kept below 5%, in thecase of the presented prototype; thus, no AR was observed.The flicker levels in the lamp were significantly reduced whencompared to that in [15].

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Fabio Luis Tomm (S’08–M’10) was born in SantoÂngelo, Brazil, in 1980. He received the B.S. andM.S. degrees in electrical and electronic engineeringfrom the Federal University of Santa Maria (UFSM),Santa Maria, Brazil, in 2004 and 2007, respectively.

From 2008 to 2010, he was a Professor with theCatarinense Federal Institute, Rio do Sul, Brazil.Since 2011, he has been with the Federal Universityof Pampa (UNIPAMPA), Bagé, Brazil, where he iscurrently a Professor in the renewable energy engi-

neering degree. His research interests include elec-tronic ballasts, high-intensity discharge lamps, dimming systems, and modelingand simulation of power converters.

Álysson Raniere Seidel (M’03) was born in SãoPedro do Sul, Brazil, in 1975. He received the B.S.and Ph.D. degrees in electrical engineering from theFederal University of Santa Maria (UFSM), SantaMaria, Brazil, in 1999 and 2004, respectively.

From 2004 to 2008, he was an Associate Professor

with the Department of Electrical Engineering, Uni-versity of Passo Fundo, Passo Fundo, Brazil. He iscurrently with UFSM, where he is a Professor withthe Colégio Técnico Industrial de Santa Maria anda Researcher with the Electronic Ballast Research

Group (GEDRE). His research interests include resonant converters, dimmingsystems, simulations, and discharge lamps.

Alexandre Campos (S’89–M’94) received theB.Eng. degree in electrical engineering from theFederal University of Santa Maria (UFSM), SantaMaria, Brazil, in 1981, the M.Eng. degree in elec-trical engineering from the Federal University of Santa Catarina (UFSC), Florianópolis, Brazil, in1986, and the Ph.D. degree in electrical engineeringfrom Concordia University, Montreal, QC, Canada,

in 1994.He is currently an Associate Professor withUFSM. He has extensive experience in electrical and

computer engineering, specially on measuring, controlling, and compensatingpower electronic systems. His main topics of interest are static power convertersapplied in electronic ballasts, power filters, and static compensators. Also, hisspecial interests are lightning systems, embedded systems, and signal process-ing. He does work with innovation and entrepreneurial education techniques,being a Specialist on CEFE (http://www.cefe.net) methodology.

Marco A. Dalla Costa (S’03–M’09) was born inSanta Maria, Brazil, in 1978. He received the B.S.and M.Sc. degrees in electrical engineering from theFederal University of Santa Maria (UFSM), SantaMaria, in 2002 and 2004, respectively, and the Ph.D.degree (with honors) in electrical engineering fromthe University of Oviedo, Gijón, Spain, in 2008.

From 2008 to 2009, he was an Associate Professorwith the Universidade de Caxias do Sul, Caxias doSul, Brazil. Since 2009, he has been an AssociateProfessor with UFSM. He is the author of more than

25 journal papers and more than 50 international conference papers. He is theholder of two Spanish patents. His research interests include dc–dc converters,power factor correction stages, dimming systems, high-frequency electronicballasts, discharge-lamp modeling, electronic starters for high-intensity dis-charge lamps, light-emitting-diode systems, and renewable energy systems.

Dr. Dalla Costa was the recipient of the Second Prize Paper Award of the2005 IEEE Industry Applications Society Meeting from the Production andApplication of Light Committee. He also serves as Reviewer for several IEEE

journals and conferences in the field of power electronics.

Ricardo Nederson do Prado (M’00) was born inItapiranga, Brazil, in 1961. He received the B.Sc.,M.Sc., and Ph.D. degrees in electrical engineer-ing from the Federal University of Santa Catarina(UFSC), Florianópolis, Brazil, in 1984, 1987, and1993, respectively.

From 1987 to 1992, he was an Assistant Professorwith the Electronics Department, Federal Universityof Minas Gerais, Belo Horizonte, Brazil. Since 1993,he has been with the Federal University of SantaMaria (UFSM), Santa Maria, Brazil, where he is

currently an Associate Professor with the Electrical Energy Processing Depart-ment. In 1997, he founded the Electronic Ballast Research Group (GEDRE).From 2005 to 2006, he was a Postdoctoral Research Scholar at the FraunhoferInstitute, Sankt Augustin, Germany. He has authored more than 180 tech-nical papers published in international journals and conference proceedings.

His research interests include high-frequency high-density power converters,fluorescent and high-pressure lamps, dimming systems, luminous efficiency,electronic ballasts, LED as a light source, and power factor correction.

Dr. do Prado is a founding member of the Brazilian Power ElectronicsSociety and a member of the Brazilian Automatic Control Society and severalIEEE societies. He is a Reviewer in the Brazilian Power Electronics Society,the Brazilian Automatic Control Society, and several IEEE societies.

hid lamp electronic ballast based on chopper converter

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