Microstrip filter and power divider with improved

out-of-band rejection for a Ku-band inputmultiplexer

Sergio Llorente-Romano*, Alejandro Garcia-Lamperez*, Magdalena Salazar-Palma*,Ana Isabel Daganzo-Eusebiot, Juan Sebastian Galaz-Villasantet, Manuel Jesus Padilla-Cruz t

*Dpto. Sefiales, Sistemas y Radiocomunicaciones, Universidad Politecnica de Madrid,ETSI Telecomunicacion, Ciudad Universitaria s/n, 28040 Madrid, SpainTel: 34-91-336-7358, Fax: 34-91-336-7362, e-mail: lamperez@ieee.org

t Alcatel Espacio, c/ Einstein s/n, PTM, Tres Cantos, 28760 Madrid, Spain.

Abstract- Two microstrip power combiners/dividers with fil-tering capabilities for a Ku-band Input Multiplexer (IMUX) hasbeen developed and measured. The requirements consist of lowinsertion loss in a wide passband, high selectivity and restrictiveout-of-band attenuation characteristics. The devices are competi-tive with similar waveguide structures, while achieving significantimprovements in dimensions and weight, at a much lower cost.

I. INTRODUCTION

This paper describes the design and construction of a deviceconsisting of two filters and a power divider for an on-boardKu-band IMUX for a satellite communications system. Theobjective of this device is to generate two duplicates of thereceived signal, filtered within the band of interest, in order toapply a different treatment to each one. More precisely, oneof the outputs is used to separate the even channels of theIMUX, while the other corresponds to the odd channels. Thisprevious division of the signal allows the IMUX channel filtersto have lower selectivity and be simpler, since the channel-to-channel guard bands are greatly increased (fig. 1). The designalso covers the more general possibility of two different inputsignals from two receivers.

IMUX channels

Even channels

2 4 6

Odd channels

1 3 5

Fig. 1. Reduction of the required selectivity for the IMUX filters throughthe separated processing of even and odd channels.

This kind of devices is traditionally implemented usingwaveguide components in order to achieve low insertion lossand high selectivity of the filter response. However, waveguidetechnology elements are typically heavy, bulky, and difficult tointegrate with other elements such as amplifiers. Our objective

is to develop a lower cost equivalent device using microstriptechnology. This approach presents some important problems,that usually limit the application of microstrip technology tolower frequency bands:

* At frequencies over the X-band, conductor, dielectric andradiation losses can be very significant, degrading thequality factor of the resonators, the insertion loss andthe out-of-band attenuation.

. The response can be affected by higher order modespropagated through the microstrip and/or casing structure.

In order to effectively substitute the waveguide technology,the original requirements must be maintained. Only a meticu-lous design allow all the microstrip capabilities to be exploited,in order to achieve this specifications. The careful designmust be extended to the casing structure and the input/outputconnectors.

II. DESIGN

The device consists of a pair of identical passband filters,connected to two common ports through a 3 dB hybrid (fig. 2).Theoretically, the port-to-port response of the whole structure(from input ports 1-1F to output ports 2-2') is the same of oneisolated filter, with a 3 dB increase of the insertion losses dueto the power distribution on the hybrid.

1'2'

1 2

Fig. 2. Block diagram of the designed device.

The original specifications for the device are as follows:* Passband frequencies from 10.70 to 12.75 GHz.. Passband return loss < 20 dB.. Rejection > 50 dB for frequencies from 13.80 GHz to

20 GHz.. Input-to-input isolation > 25 dB.

This combination of passband and out-of-band attenuationspecifications requires a filter with an achievable order n > 7,that can be reduced to n > 5 by inserting one transmission

33rd European Microwave Conference - Munich 2003

1

zero at finite frequency. However, the large relative bandwidthcorresponds to input and output coupling coefficients muchhigher than the ones that can be reasonably obtained formicrostrip resonators, with external quality factors as low asQe 6.The original specified passband is intended to cover the

complete band assigned to satellite communications systems,so that only one design is needed to satisfy all the possibleapplications. This ambitious goal can be eased by splitting theentire passband into two subbands. In this case, two differentdesigns must be obtained. The new specifications for thetwo devices are shown in table I. Their relative bandwidthsare 9.8% and 8.6%.

A. Coupled-Line Filter with Capacitive Gap Cross-Coupling

The selectivity specifications from table I can be satisfiedusing Chebyshev function filters of order n > 7, with insertionloss in the passband higher than 2 dB. In order to reduce the fil-ter order and therefore the insertion loss, quasielliptic functionfilters with one transmission zero in the upper stopband arechosen. The position of this transmission zero is optimized sothat the transfer function satisfies the attenuation requirements.The required filter order is reduced to n = 5, at the costof requiring asynchronously tuned resonators, and one cross-coupling between them [1].

Figure 3(a) shows the chosen coupling topology, that allowssymmetrical configurations. The filters consist of parallel-coupled A/2 resonators, with the cross coupling implementedas a capacitive gap between resonators 2 and 4 (fig. 3(b)).

the slight detuning effect on resonators 2 and 4 due to thefringing field in the capacitive gap [3].The fabrication process that has been used limits the nar-

rowest line width and the narrowest spacing between lines to100 ,um. Due to the wide bandwidth of the filters, very stronginput and output couplings (Ms, and M5L) are required.Therefore, those couplings cannot be implemented as induc-tively coupled transmission lines. One possible alternative isthe use of tapped lines [5]. A better geometrical configurationis achieved by implementing the input and output couplingsas high impedance line sections with electrical length approx-imately equal to A/4 at the filter central frequency. This isthe chosen solution because the linear geometry of the filteris preserved and it is easier to reduce the casing dimensions,so that the excitation and propagation of higher order modesare reduced. (fig. 3(b)).

Figure 4 shows the measured and simulated widebandresponse of one of the filters. Agilent ADS has been usedto carry out the simulations.

0

10

E 20

R -30-

z -40

50-

60,

-70Source 5

-80

M23\ / M34 8 10 12 14 16 18 20Frequency (GHz)

Fig. 4. Measured and simulated filter response (11.7(prototype). The specified passband is indicated.

(a) Resonator and coupling topology (solid line: main coupling; dashedline: cross coupling).

(b) Microstrip layout.

Fig. 3. Parallel coupled line filter with capacitive gap cross-coupling andhigh impedance input lines.

The design process starts from a conventional five-polebandpass filter, formed by A/4 sections of coupled transmis-sion lines. This original filter can be obtained using one of thewell known synthesis methods [2], [3]. Then, the topologyis modified so that the capacitive cross-coupling M24 isintroduced, by bending half the structure (fig. 3(b)) [4]. Thespacings between lines (included the capacitive gap) are thenoptimized. The resonator lengths must be also optimized, inorder to adjust their resonance frequencies, and to compensate

Simulated_ Measured

22 24 26

D 12.75 GHz band

B. Broadband Branch-Line HybridA conventional two-arm branch-line hybrid suffers from a

lack of operating bandwidth and flat amplitude characteristics.This problem makes it unsuitable when relative bandwidthsof 10% are required. The more evident solution to this problemis the use of a branch-line with three or more arms, but dueto the high working frequencies, inconvenient line impedancesand lengths are needed.The solution adopted to broadband the response of a two-

arm branch-line 3 dB coupler is described in [6] and [7]. Itconsists of designing the hybrid formed by A/4 transmissionline sections, having a defined mismatch at the center fre-quency. Then, two A/4-length cascaded matching sections inseries at each port leads to reflection loss minima and isolationmaxima at two frequencies separated from the central one. Theresult is an hybrid with equiripple transmission and couplingcharacteristics in an arbitrary bandwidth, that can be morethan 40%. Moreover, it is possible to replace each two A/4cascaded sections with a single high impedance A/2 (fig. 5)without significant performance degradation.

33rd European Microwave Conference - Munich 2003

TABLE IFILTER SPECIFICATIONS WITH SPLITTED PASSBAND.

PassbandRejection > 35 dBRejection > 50 dB

Prototype 110.70-11.80 GHz

12.8 GHz < f> 14 GHz14 GHz < f < 20 GHz

Prototype 211.70-12.75 GHz

12.8 GHz < f> 14 GHz14 GHz < f < 20.00 GHz

spectively (including the 3 dB attenuation due to the powerdistribution to each output). Finally, fig. 11 shows the input-to-input isolation, higher than 25 dB for both prototypes.

Fig. 5. Microstrip layout of a broadband branch-line hybrid. -100

III. CONSTRUCTION OF PROTOTYPES AND EXPERIMENTALRESULTS

Two prototypes corresponding to each passband have beenfabricated on RT/Duroid 6002 substrate with a relative dielec-tric constant c, = 2.94 and a height of 0.254 mm. The coppermetallization thickness is t = 35 um. The devices interfacesare SMA-3.5 mm. connectors, with a characteristic impedanceof 50 Q.

Fig. 6 shows the complete microstrip layout of the 10.70-12.80 GHz prototype, corresponding to the block diagram infig. 2. The internal section of the casing is also represented.The casing size has been reduced as possible in order todifficult the propagation of modes associated to the waveguidestructure that is formed by the enclosing walls. The criticaldimension is the separation between the lateral walls, that fixesthe cut frequency of the TEO1 rectangular waveguide mode,easily excited by the microstrip structure. The value of thiscut frequency is chosen high enough in order to not degradethe attenuation of the filter in the specified stopband.The final outer dimensions of the prototypes are approxi-

mately 75 x 25 mm.

m-20

° -30

D -40

O -500

a)a -60 j-70

-808 10 12 14 16 18

Frequency (GHz)20 22 24

Fig. 7. Measured wideband response (10.70 11.80 GHz band prototype).Attenuation and passband specification masks are included.

0

-10:

: 20 -

06 30

Dr -40-

c-50-

a)60-

-70

10 12 14 16 18Frequency (GHz)

20 22 24

Fig. 6. Complete layout of the device (10.7 11.8 GHz band). The casingsection is shown.

All measurements were carried out using an Agilent Tech-nologies E8364A Network Analyzer. Since this is a two portmeasurement system, the two additional ports are loaded withbroadband 50 Q loads. Figures 7 and 8 show the measuredwide-band response of each prototype. The passband andattenuation specification masks are included. It can be seen

that these specifications are achieved. The attenuation is higherthan 50 dB for frequencies up to 20 and 21.5 GHz, respec-

tively. Figures 9 and 10 show the corresponding passbandinsertion losses. Those are better than 5.2 and 5.4 dB, re-

Fig. 8. Measured wideband response (11.70 12.75 GHz band prototype).Attenuation and passband specification masks are included.

IV. CONCLUSION

Two microstrip input sections for a Ku-band IMUX, withfiltering and power distribution capabilities, have been de-signed, constructed and measured. Each section is formedby two identical filters and a 3 dB hybrid. The channels,centered at 11.25 and 12.225 GHz, have relative bandwidthsof 9.8% and 8.6%, respectively. Due to restrictive selectivityrequirements, filter responses with a transmission zero at the

33rd European Microwave Conference - Munich 2003

-80 "g

8

31L7

10.75 11 11.25 11.5Frequency (GHz)

Measured passband insertion loss (10.70

11.75 12

11.80 GHz band

upper stopband have been implemented. The devices show lowinsertion loss (2.2 and 2.4 dB), good wideband attenuationcharacteristics and high isolation. A meticulous design of theplanar structure and the casing geometry allows the devicesto be competitive with analogous waveguide structures, at alower cost, dimensions and volume.

ACKNOWLEDGEMENT

This work has been conducted by the Grupo de Microondasy Radar (Dpto. SSR, ETSI Telecomunicacion, UPM, Spain)under contract with Alcatel Espacio, Spain, (patent pending).It has also been partially financed by the Projects TIC2002-02657 and TIC2002-04569-C02-CO1 of the Spanish Ministryof Science and Technology.

REFERENCES

[1] R. J. Cameron and J. D. Rhodes, 'Asymmetric realizations for dual-modebandpass filters," IEEE Trans. Microwave Theory Tech., vol. MTT-29,no. 9, pp. 51-58, Jan. 1981.

[2] S. B. Cohn, "Parallel-coupled transmission-line-resonator filters," IRETrans. Microwave Theory Tech., vol. MTT-6, pp. 223-231, Apr. 1958.

[3] G. Matthaei, L. Young, and M. T. Jones, Microwave Filters, ImpedanceMatching Networks and Coupling Structures. New York: McGraw-HillBook Company, 1964.

[4] J.-S. Hong and M. J. Lancaster, "Transmission line filters with advancedfiltering characteristics," in Proc. 2000 IEEE MTT-S International Mi-crowave Symp., vol. 1, 2000, pp. 319-322.

[5] --, Microstripfilters for RF/microwave applications. New York: JohnWiley & Sons, Inc., 2001.

[6] J. V. Ashforth, "Design equations to realise a broadband hybrid ring or atwo-branch guide coupler of any coupling coefficient," Electronic Letters,vol. 24, pp. 1276-1277, Sept. 1988.

[7] B. Mayer and R. Knochel, "Branchline-couplers with improved designflexibility and broad bandwidth," in Proc. 1990 IEEE MIT-S InternationalMicrowave Symp., vol. 1, 1990, pp. 391-394.

11.75 12 12.25 12.5 12.75 13Frequency (GHz)

Measured passband insertion loss (11.70 12.75 GHz band

10.70-11.80 GHz prototype11.70-12.75 GHz prototype

t'-'

;f'|

8 10 12 14 16 18Frequency (GHz)

20 22 24

Fig. 11. Measured port-to-port isolation (both prototypes).

33rd European Microwave Conference - Munich 2003

-4.4

-4.6

-4.8

0-j -5.2 -

ft

0

a)'n -5.4

-5.6

-5.8

-610.5

Fig. 9.prototype).

-4.4

-4.6

-4.8

m

0j -5.20

a),) -5.4

-5.6

-5.8

-611.5

Fig. 10.prototype).

0r

-10

-20

- 30

.o -40

50

-60

-70

-80