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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 56, NO. 10, OCTOBER 2008 2293

Multiband Electromagnetic-Bandgap Structuresfor Applications in Small Form-Factor

Multichip Module PackagesTelesphor Kamgaing, Senior Member, IEEE, and Omar M. Ramahi, Senior Member, IEEE

Abstract—The design and implementation of package-levelelectromagnetic-bandgap (EBG) structures is presented. Byusing spiral-based inductance-enhanced electromagnetic-bandgapstructures (IE-EBGs), the relative periodicity for achievingbandgap at extremely low frequencies is substantially reducedin comparison to traditional EBGs. Using both full-wave elec-tromagnetic simulation and experimental characterization, it isdemonstrated that this type of structure can exhibit multiplebandgaps, which are individually tunable through variation of theinductance per unit area and/or the unit cell periodicity. Samplestructures with dimensions compatible with today’s micropro-cessor packaging technology are designed and fabricated in amultilayer organic flip-chip ball-grid array package substrate.These package-embedded IE-EBGs have unit cell dimensions lessthan 750 m and exhibit the first forbidden bandgaps for electro-magnetic wave propagation below 10 GHz, which is a frequencyband of interest for commercial wireless communication systems.

Index Terms—Artificial magnetic conductor, electromag-netic-bandgap (EBG) structures, multichip packaging, multilayerorganic package substrate, signal isolation.

I. INTRODUCTION

T HE tremendous potential of electromagnetic-bandgap(EBG) structures as circuit elements in filters, waveg-

uides, and antennas has made them a great attraction in themicrowave engineering community. While early research inthis area is mainly focused on optical frequencies, the intro-duction of innovative structures such as the high-impedanceelectromagnetic surface [1], where the bandgap behavior ofthe structures relies primarily on their capacitive and induc-tive loading and less on the periodicity, has paved the wayto designing EBG structures at microwave frequencies. Mostrecently, the use of EBG and other artificial electromagneticconductors for mitigating switching noise in high-speed circuitshas been introduced [2]–[4]. Additional research [5] has alsoshown that an EBG structure can be used for signal isolationin RF/analog mixed-signal systems. The solution demonstratedin [4] and similar research, however, mainly addresses an EBG

Manuscript received January 1, 2008, revised May 27, 2008. First publishedSeptember 23, 2008; current version published October 8, 2008.

T. Kamgaing is with Components Research, Intel Corporation, Chandler, AZ85226 USA (e-mail: [email protected]).

O. M. Ramahi is with the Electrical and Computer Engineering Department,University of Waterloo, Waterloo, ON, Canada N2L 3G1 (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2008.2003525

that can only be implemented for printed circuit board (PCB)modules [6]–[25], where the dimensions are relatively large.

In [10], the concept of spiral-based EBG structures wasmainly introduced for switching noise mitigation and it wasshown that enhancement of the EBG inductance can be achievedby using an additional layer of the multilayer PCB exclusivelyfor routing the inductor. It has also been demonstrated numer-ically [11] that the spiral can be patterned at the same level asthe patch to enhance the inductance per unit area, and hence,shift the location of the bandgap towards lower frequencies. In[17], single- and two-turn spiral EBGs were introduced as partof a systematic EBG design approach. This work, however,only focused on analyzing the first or primary bandgap anddid not address the existence of higher bandgaps. From thecost standpoint, the inductance enhancement approach has thesignificant advantage that it does not require new technologydevelopment or inclusion of exotic materials such as high-per-mittivity or high-permeability materials as required by otherapproaches reported in the literature [23], [24].

With the continuous integration of wireless communicationsystems and the associated demand for form-factor reduction, asubstantial downscaling of the EBG structures to package leveldimensions is necessary before this technology can be used insmall form-factor antennas and for signal isolation in highly in-tegrated RF modules. According to the industry trends for mul-tiradio integration, such a technology will have the most use insmall form-factor devices such as the ultramobile personal com-puter (UMPC), where multimode radios including voice, data,and location have to coexist.

In this paper, EBG structures are analyzed in a parallel-platewaveguide environment, which is representative for the powerdelivery network of a wireless communication multichipmodule as the one represented in Fig. 1. Besides mitigatingnoise propagation between the main chips of the system,the main purpose of this study is twofold, i.e., introducing amultiband EBG that can be used in place of a single widebandstructure while occupying relatively smaller real estate anddemonstrating the feasibility of this novel EBG at the packagelevel without inclusion of any special materials.

This paper is organized as follows. In Section II, we brieflyreview the concept of EBG structures and some techniques thathave been used to enable operation at very low frequencies. InSection III, we introduce and discuss the concept of multibandinductance-enhanced EBG (IE-EBG). Section IV addressesthe experimental verification of the IE-EBG concept throughpackage-level implementation.

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2294 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 56, NO. 10, OCTOBER 2008

Fig. 1. Illustrative view of noise propagation in multichip-module packagecontaining analog, digital, and mixed signal circuits from [30].

Fig. 2. Traditional mushroom EBG: top view, equivalent electrical model, andcross-sectional view.

II. OVERVIEW AND THEORY OF EBG STRUCTURES

EBG structures are typically realized by periodically loadinga grounded dielectric material with periodic metallic structures.Fig. 2 is an illustration of the high-impedance structure (HIS),also known as the mushroom EBG, originally introduced in[1]. It consists of metallic patches connected to the ground ofthe nonconductive dielectric substrate using vertical vias. Anytwo adjacent cells of this structure can be modeled as a parallel

resonator, whereby , the capacitance per unit area, rep-resents the fringing capacitance between adjacent patches and

, the inductance per unit area, represents the inductive path(patch–via–ground–via–patch) from one patch to the next. Asurface current induced on the patch will flow through the in-ductive path at low frequencies and through the capacitive pathat high frequencies. This means that the mushroom EBG allowsthe propagation of TM waves at low frequencies and the prop-agation of TE waves at high frequencies. Similar to res-onators, there exists a transition region, also known as the EBG,

Fig. 3. Capacitively enhanced EBG [1].

where the EBG does not support the propagation of any electro-magnetic waves. The EBG is primarily characterized throughthe center frequency of the bandgap given by

(1)

Considering a 2-D wave propagation in the structure, each maindirection can be modeled as a cascade of an identical parallel

resonator, whose bandwidth is given by

(2)

whereby is the characteristic impedance of the high-impedance surface and is the impedance of free space.

Other figures-of-merit include the fractional bandwidth, aswell the upper and lower edges of the bandwidth, which canbe determined using the method described in [25].

From (1), it can be seen that low-frequency EBGs require anincrease in capacitance per unit area or in inductance per unitarea. Inductance increase can be obtained by using longer viasor high-permeability material. Similarly, a capacitance increasecan be achieved by using high-permittivity material or by sub-stantially reducing the spacing between the adjacent patches. Tocircumvent the minimum trace spacing defined by given tech-nology nodes, multilayer patches, as illustrated in Fig. 3, can beused, whereby an overlap between the patches enables capaci-tance density in the order of that of regular parallel-plate capaci-tors. Other EBG structures, such as the uniplanar compact pho-tonic bandgap (UC-PBG) [26] and the alternating-impedanceEBG (AI-EBG) [27] have been demonstrated and do not usevertical vias. This, however, results in very large periodicitiesfor low-frequency applications [26], [27]. Furthermore, planarEBG structures can potentially lead to degradation in the signalintegrity, as was demonstrated recently in [30].

III. TOPOLOGY AND CONCEPT OF SPIRAL-BASED IE-EBG

A. Topology

Fig. 4 shows the top and cross-sectional views of the unit cellof the spiral-based IE-EBG. It is obtained by replacing the patchof the mushroom EBG cell [1] with a spiral inductor that has avarying metallization linewidth. The spiral inductor has one endconnected to a via such that the effective inductance of the viais increased. While full-wave analysis is necessary for accurate


KAMGAING AND RAMAHI: MULTIBAND EBG STRUCTURES FOR APPLICATIONS IN SMALL FORM-FACTOR MULTICHIP MODULE PACKAGES 2295

Fig. 4. Isometric and cross-sectional views of the IE-EBG unit cell using aspiral inductor as patch. Reduced Brillouin zone is illustrated.

determination of the inductor performance, an estimate of theinductance enhancement due to the spiral can be obtained fromthe thin-film inductance formulas reported in [28], which doesnot take into account any potential interaction of the spiral withsubstrate or potential substrate embedded grounds. In this ap-proach, the total self-inductance of the spiral is first determinedas the sum of the self-inductances of the individual segments, asillustrated in (3), whereby , , and represent width, thick-ness, and length of trace segment , respectively,

(3)

Second, the mutual inductance between any two parallel seg-ments is defined by (4). In this equation, and representthe length overlap and separation between trace segments I andI, respectively,

(4)

The total mutual inductance is the obtained by summing upthe individual mutual inductance pieces

(5)

The overall inductance of the spiral is then obtained as thesum of self-inductance and mutual inductance

(6)

For a defined technology or substrate stackup, the surfaceimpedance can be controlled by varying the geometrical param-eters of the inductor, which are the inner diameter, number ofturns, linewidth, and line spacing. With reference to (2) and alsoin analogy to the resonator, increasing the inductance perunit area instead of capacitance per unit area leads to a largerfractional bandwidth since the later is proportional to .

B. Simulation and Discussion of Results

To validate the concept of the spiral-based IE-EBG, we firstconsider implementation with PCB dimensions, where the pe-riod or unit cell dimension is 10 mm. Limited rows of EBG cellsare integrated inside a parallel-plate waveguide to evaluate its

Fig. 5. Modeled dispersion diagram of an IE-EBG in parallel-plate waveguideenvironment. Cell dimension is 10 mm � 10 mm. Normalized wavenumber isindicated for each direction of propagation.

Fig. 6. Insertion loss of a 10 cm � 10 cm parallel-plate waveguide with em-bedded IE-EBG. Period is 10 mm and distance between the ports is 0.3 mm.

impact on the waveguide resonant modes. Using Ansoft Cor-porations’ commercially available electromagnetic High Fre-quency Structure Simulator (HFSS) [29], we evaluate both thedispersion diagram, where we solved only the first three eigen-modes, and the transmission parameter between two desig-nated points of the finite array of 10 10 cells.

Fig. 5 shows the full – – – dispersion diagram of thestructure, computed on the boundary of the reduced Brillouinzone. It represents the propagating frequencies as a function ofboth the wavenumber and direction of wave propagation. Thisanalysis provides the behavior of waves propagating tangen-tially to the surface with the – and – segments repre-senting waves propagating orthogonally to the surface and the

– segment representing waves propagating in the diagonaldirection of the surface. Two bandgaps are exhibited, one from1.2 to 2 GHz, and the other from 3.42 to 3.75 GHz.

Fig. 6 shows the insertion loss versus frequency for thesame structure. The 20-dB bandgaps are exactly the same asthose predicted by the dispersion diagram. In addition, there



Fig. 7. Dispersion diagram illustrating the effect of varying the gap size on thelocation and width of the bandgap. Solid line is for g = 0:6mm and dotted lineis for g = 1:2 mm.

is a global minimum in the insertion loss curve at 2.6 GHz. Adirect comparison with the dispersion diagram indicates thatthis frequency point does not represent a narrow stopband, butit is rather the frequency point where the second mode starts totransition from a predominantly TE wave to a predominantlyTM wave. At this frequency point, the second mode is a pureTEM mode, as it intercepts the light line in the medium.

The first mode is a TM mode, which starts as a forward propa-gating TEM mode at very low frequency and low wavenumber,and changes into a forward propagating TM surface wave. Atvery high wave numbers, a decrease in the group velocity ofthis mode indicates that it is propagating backward. The secondmode is a hybrid mode, which starts as a TE wave at a verylow wavenumber and behaves like a TM wave at a very highwavenumber. The third mode is a pure TE wave.

C. Bandgap Tuning by Varying the Gapwidth

The first of two methods considered in controlling thebandgap(s) of the EBG in a parallel-plate environment consistsof varying the separation or gap between adjacent patches.In doing this, all other parameters are fixed. Fig. 7 shows thedispersion diagram ( – segment only) of the unit cell, wherethe patch separation has been varied from 0.6 to 1.2 mm. Thisis done by increasing the width of the outer loop of the spiralinductor. The upper band or second bandgap moves towardslower frequencies with continuous decrease in the bandwidth.When the gap reaches a certain value, as indicated by the solidline, the second bandgap disappears. On the other hand, it canbe seen that the lower or first bandgap remains unaffected.

When the gap between patches is very large, a secondbandgap is generated. As the size of the gap decreases, the thirdmode moves faster towards lower frequencies and at aboutthe same rate across the entire spectrum (wavenumbers). Thesecond mode also moves towards lower frequency. This shift ismore pronounced at higher frequencies. Since the second modeshifts much more slowly, the second bandgap will eventuallymerge into the first band due to the simultaneous presence ofboth capacitive and magnetic coupling between the adjacentcells.

Fig. 8. Effect of number of turns on the fundamental stopband of the IE-EBGin parallel-plate waveguide environment.

D. Bandgap Tuning by Varying the Number of Turnsof the Spiral Inductor

The second method considered for tuning the bandgaps con-sists of varying the number of turns of the spiral inductor usedas the IE-EBG patch. In this case, we consider a unit cell of aperiod of 5 mm. The substrate is lossless with a relative dielec-tric constant of 4.4. The EBG height and the separation betweenthe two plates remain 1.54 mm. The gap between the patches is0.6 mm and the via diameter is 0.4 mm. The main electricalperformance attributes are all derived from the – section ofthe dispersion diagram. The number of turns , ( ) of the spiralinductor was varied form 1 to 2.5. The properties of the ob-tained bandgaps are summarized in Fig. 8 along with those of thestandard mushroom structure represented by . The stan-dard EBG exhibits only one bandgap, which is located at muchhigher frequencies than the fundamental bandgap of structureswith an IE-EBG. As the number of turns of the spiral inductorincreases, the surface inductance increases very fast, leading tothe lowering of the center frequency. At the same time there isa decrease in the absolute bandwidth of the forbidden bandgap.A potential reason for the absolute bandwidth is the redistribu-tion of the electrical charges along the edges of the spiral in-ductors, which is the same electrical phenomenon that explainsself-resonance of the inductors. It is also important to note thatas the inductance goes beyond a certain critical value, the secondbandgap will also disappear. This specific case is illustrated inFig. 9.

IV. PACKAGE LEVEL IMPLEMENTATION OF EBGs AND

ELECTRICAL CHARACTERIZATION RESULTS

A. Design and Fabrication

In order to evaluate the electrical performance of theIE-EBGs for low-frequency applications, several structureswith periodicities ranging from 400 to 750 m have beendesigned and fabricated on the multilayer organic flip-chipball-grid array (FCBGA) substrate described in [32]. Theperformance of these structures is evaluated by inserting them



Fig. 9. Dispersion diagram illustrating the disappearance of the secondbandgap due to appropriate increase in inductance.

Fig. 10. Representative cross section of multilayer organic package substrate:(a) with and (b) without embedded EBG unit cell and (c) illustrative side viewwith testing pads.

in a parallel-plate waveguide environment, as illustrated by thecross-sectional representation of Fig. 10.

Fig. 11 shows the top view of the fabricated EBGs, wherebyall cells have a unit cells of periodicity m in both

- and - direction. EBG_1 and EBG_2 are traditional mush-room EBGs and have unit cell size of 750 m and patch to patchgap of 100 and 300 m, respectively. EBG_3 and EBG_4 usespiral patches with the same periodicity and gap between thepatches as their mushroom counterpart EBG_1. For simplicity,the inductor trace has a uniform width for all turns. The onlykey difference between EBG_3 and EBG_4 resides in the in-ductance density, which, in this case, is controlled by varying thenumber of turns and trace width of the spiral patches. EBG_3uses a two-turn spiral patch with a starting inner diameter of200 m, a uniform trace width of 100 m, and a spacing of26 m between adjacent turns. EBG_4 uses a three-turn spiralpatch with 200- m starting inner diameter and trace-to-tracespacing of 26 m. The associated trace width is 50 m.

B. Measurement and Discussion of Results

The electrical performances of the fabricated structures weremeasured in the form of two-port scattering ( ) parameters

Fig. 11. Snapshots of the top view of fabricated EBG structures.

using a 50-GHz performance network analyzer (PNA) andon-wafer ground–signal–ground (GSG) probes, as illustrated inFig. 10(c). The two test ports are separated by 15 EBG cells. Inaddition, the separation between the EBG columns, where thetest port is located, is m, as illustrated in Fig. 10(c).A standard short-open-load-through (SOLT) calibration wasused. Fig. 12 shows the transmission coefficient of the refer-ence parallel-plate waveguide in the absence of the IE-EBG.It can be seen that several resonant modes are excited. The

-parameters of the traditional EBG_1 is plotted in thesame plot. The traditional EBG does not exhibit any stopband



Fig. 12. Measured electrical performance of parallel-plate waveguide with andwithout traditional EBG (period = 750 �m, spacing = 100 �m).

Fig. 13. Modeled electrical performance of traditional EBG in parallel-platewaveguide environment.

behavior below 50 GHz, which is the maximum frequency thatcan be measured with the PNA.

In Fig. 13, we show electrical modeling results of the tradi-tional EBG. Two different via lengths are investigated. In thecase of short vias, the separation between the underlying groundplane and the EBG patches is 30 m and the separation betweenthe EBG patches and the signal plane is 75 m. In the case of thelong via, the via length is increased to 75 m. It can be noted thatthe EBG with the short via exhibits a bandgap above 60 GHz,whereas the structure with long vias exhibits a bandgap startingabove 40 GHz.

Fig. 14 shows the modeled and measured electrical per-formance of the two-turn spiral IE-EBG. This structure ex-hibits a total of four bandgaps below 100 GHz, including twobelow 50 GHz. The lower first fundamental bandgap is around13 GHz.

Fig. 15 shows the measured electrical performance of thethree-turn spiral IE-EBG. This EBG exhibits six bandgapsbelow 100 GHz including three below 50 GHz. The first twobandgaps are located at 9 and 27 GHz, respectively.

Fig. 14. Modeled and measured electrical performance of IE-EBG using two-turn spiral as patch (period = 750 �m, spacing = 100 �m).

Fig. 15. Measured electrical performance of IE-EBG using three-turn spiral aspatch (period = 750 �m, spacing = 100 �m).

V. CONCLUSION

Spiral-based EBG structures have been investigated usingelectromagnetic modeling and experimental characterization.It was shown that this type of EBG exhibits multiple forbiddenbandgaps, whereby the first bandgap is determined by thedimension of the unit cell. The second bandgap was shown tobe primarily dependent on the capacitive and inductive loadingof the unit cell and can be tuned independently of the primarybandgap. Package-level structures, exhibiting bandgaps in the9- and 27-GHz frequency ranges, have been designed with lessthan 750- m periodicity and implemented in a standard micro-processor multilayer organic packaging substrate. The realizedfeature indicates that this technology is an enabler for applyingEBGs into portable wireless communication systems includingwireless local area networks (WLANs) and ultra-wideband(UWB). While the smallest frequency demonstrated in thisstudy is 9 GHz, it is expected that a mere optimization of thespiral geometry could lead to operation at lower frequencieswithout any change in the package material or processing. SuchEBG structures will then be suitable for compact antennasand noise isolation for future technology platforms requiringmultimode radios.



ACKNOWLEDGMENT

The authors would like to thank all members of the RadioFrequency Packaging Integration Working Group, Intel Corpo-ration, Chandler, AZ, for various technical discussions. The au-thors would also like to acknowledge L. Wojewoda, Assemblyand Test Technology Development, Intel Corporation, for elec-trical measurement support.

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Telesphor Kamgaing (S’00–M’04–SM’05) re-ceived the Diplom.-Ingenieur degree in electricalengineering from the Darmstadt University of Tech-nology, Darmstadt, Germany, in 1997, and both theM.S. and Ph.D. degrees in electrical engineering fromthe University of Maryland at College Park, in 2003.

In 1999, he was a Guest Researcher with theNational Institute for Standards and Technology(NIST), Gaithersburg, MD, where he was involvedin the modeling and applications of silicon–carbidedevices. From 2000 to 2004, he was with the Digital

DNA Laboratories, Motorola Inc., Tempe, AZ, where he was involved in theresearch and development of silicon integrated passives and RF modules forwireless communication. Since 2004, he has been with the Intel Corporation,Chandler, AZ, where he has held several technical leadership positions in-cluding Research and Development Technology and Development Mangerresponsible for the electrical analysis of RF and low density interconnectpackaging. Most recently, he has been a Staff Research Scientist with a mainfocus on radio coexistence on ultra-small form-factor platforms and packaging



development for millimeter-wave applications. He has authored or coauthoredover 50 technical papers in refereed journals and conference proceedings. Heholds one U.S. patent with over 15 patents pending. His research interestsinclude CPU power delivery and EBG structures for various applications.

Dr. Kamgaing is a senior member of the IEEE Microwave Theory and Tech-nique Society and the IEEE Components and Packaging Manufacturing Tech-nology Society. He was the recipient of numerous awards including a govern-ment scholarship, several Best Student Awards and conference travel grants.

Omar M. Ramahi (S’86–M’90–SM’00) receivedthe B.S. degrees in mathematics and electrical andcomputer engineering (summa cum laude) fromOregon State University, Corvallis, in 1984, andthe M.S. and Ph.D. degrees in electrical and com-puter engineering from the University of Illinois atUrbana-Champaign, in 1986 and 1990, respectively.

From 1990 to 1993, he was a Visiting Fellow withthe University of Illinois at Urbana-Champaign.From 1993 to 2000, he was with the Digital Equip-ment Corporation (now Hewlett-Packard), where he

was a member of the Alpha Server Product Development Group. In 2000, hejoined the faculty of the James Clark School of Engineering, University ofMaryland at College Park, as an Assistant Professor, and later as a TenuredAssociate Professor. At the University of Maryland at College Park, he wasalso a faculty member of the CALCE Electronic Products and Systems Center.He is currently a Professor and the Natural Sciences and Engineering ResearchCouncil of Canada (NSERC)/Research in Motion (RIM) Industrial ResearchAssociate Chair with the Electrical and Computer Engineering Department,University of Waterloo, Waterloo, ON, Canada. He holds cross-appointmentswith the Physics Department and Astronomy and Mechanical EngineeringDepartment, University of Waterloo. He was instrumental in the developmentof computational techniques to solve a wide range of electromagnetic radiationproblems in the fields of antennas, high-speed devices and circuits, andelectromagnetic interference (EMI)/electromagnetic compatibility (EMC). Heis a consultant to several companies. He was a cofounder of EMS-PLUS LLCand Applied Electromagnetic Technology LLC. His research interests includeexperimental and computational EMI/EMC studies, high-speed devices andinterconnects, biomedical applications of electromagnetics, novel optimizationtechniques, and interdisciplinary studies linking electromagnetic applicationto novel materials. He has authored or coauthored over 170 journal and con-ference papers. He coauthored EMI/EMC Computational Modeling Handbook(Springer-Verlag, 2001, 2nd ed.).


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