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IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY, VOL. 4, NO. 1, JANUARY 2014 75

0.34-THz Wireless Link Based on High-OrderModulation for Future Wireless Local

Area Network ApplicationsCheng Wang, Bin Lu, Changxing Lin, Qi Chen, Li Miao, Xianjin Deng, and Jian Zhang

Abstract—This paper presents a 0.34-THz wireless link forfuture wireless local area networks (WLANs), which is based onhigh order 16-quadrature amplitude modulation (16QAM). Thesystem adopts super heterodyne transceivers and parallel digitalsignal-processing techniques. The 0.34-THz transceiver consists ofa 0.34-THz subharmonic mixer, a 0.34-THz waveguide H-ladderbandpass filter, and a 0.17-THz multiplier chain. Two 0.34-THzCassegrain antennas with 48.4-dBi gain have been developedto extend the transmission distance. Based on a 32-way par-allel signal processing, we have successfully realized the 3-Gb/s,16QAM real-time modulator and demodulator. The measureddata indicate that the lowest bit error rate of the 0.34-THz, 3-Gb/sdata link is 1.784 10 over a 50.0-m line-of-sight channel. Themaximum received energy per bit to noise power spectral densityratio (E N ) is 23.8 dB, while the output power of transmitteris 17.5 dBm and the noise temperature of receiver is 5227 K. Inaddition, this paper presents a 0.34-THz WLAN prototype basedon IEEE 802.11b/g protocol. TheWLAN prototype, which consistsof an access point and two terminal nodes, achieves a transmissiondata rate of 6.536 Mb/s over 1.15 m by using rectangular hornantennas.

Index Terms—Millimeter-wave communication, quadratureamplitude modulation (QAM), wireless local area network(WLAN).

I. INTRODUCTION

T HE demand for high-data-rate wireless communicationis increasing synchronously with the development of

military and consumer electronics applications. The analog car-rier of digital data link is extending from the microwave bandto millimeter and terahertz bands. Terahertz communicationis attractive in high-speed satellite data link, ultrashort-rangeinterface of portable device, instantaneous or intermittentlink, and wireless local area networks (WLANs). In the lastdecade, the academia has been striving towards optimizedelectronic and photonic devices, effective system architecturesand high-performance protocols for THz communication.Significant progress has been achieved in recent publications.

Manuscript received April 28, 2013; revised July 04, 2013, August 22, 2013,and October 24, 2013; accepted November 15, 2013. Date of publication Jan-uary 02, 2014; date of current version January 17, 2014. This work was sup-ported by the Terahertz Research Center, Institute of Electronic Engineering,China Academy of Engineering Physics (CAEP-IEE).The authors are with the Terahertz Research Center, Institute of Electronic

Engineering, China Academy of Engineering Physics, Mian Yang, 621900,China (e-mail: [email protected]).Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TTHZ.2013.2293119

Based on unitraveling carrier photodiode (UTC-PD) and InPhigh-electron mobility transistors (InP HEMTs), respectively,NTT Microsystem Integration Laboratories in Japan has de-veloped two 120-GHz wireless links [1], whose data rate is10 Gb/s and maximum transmission distance is 5.8 km [2].NTT recently realized a 300-GHz 24-Gb/s communicationlink over laboratory distance also by UTC-PD [3]. Based on a50-nm InP metamorphic high-electron mobility transistor (InPmHEMT), a 220-GHz transceiver has been developed by theFraunhofer Institute for Applied Solid-State Physics (IAF) inGermany [4], which realized 10-m transmissions of 25-Gb/son–off-keying (OOK) signal and 14-Mb/s 256 quadratureamplitude modulation (QAM) signal [5]. Xi’an University hasdeveloped a 135-GHz link with 10-Gb/s data rate based ona 0.13- m MOSFET switch ASK modulator [6]. Bell Labshas transmitted 2.5-Gb/s signal in the 625-GHz band usingduobinary baseband modulation [7].Most THz communication prototypes give up high-order

modulation due to challenges on implementation, includingcomplexity and speed limitation. ASK or OOK schemes seemto be more reasonable. They can achieve tens of Gb/s easilythrough analog modulators. However, in this paper, we proposea system framework based on super heterodyne Schottkybarrier diodes transceiver and parallel digital signal processing(DSP) techniques for 0.34-THz 16QAM transmission. It bringsthe advantages of: 1) higher spectrum efficiency [8]; 2) thepotential of higher output power from vacuum and solid-stateamplifier [9]–[12]; and 3) higher channel distortion tolerancefrom digital equalization algorithms. A 0.34-THz wireless linkhas been developed according to this framework. This linkhas successfully transmitted a 3-Gb/s, 16QAM signal over a50.0-m line-of-sight (LOS) channel with 17.5 dBm outputpower and 5227 K received noise temperature. The maximumenergy per bit to noise power spectral density ratio E N is23.8 dB. The minimum bit error rate (BER) is 1.784 10 .The minimum E N is 13.8 dB when 10 .Short-range communication is essential for exploiting the

terahertz spectrum resources. An overview article paid muchattention to short-range high speed communication above0.3 THz [13]. Furthermore, theoretical and experimental re-searches on propagation of THz wave in room environmenthave already been performed [14], [15]. IEEE 802.15 wirelesspersonal area network (WPAN) workgroup constituted IEEE802.15 THz interest group (IG-THz) to promote high-speedprotocols for THz communication. In this paper, we havedeveloped a 0.34-THz WLAN prototype based on the IEEE

2156-342X © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

76 IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY, VOL. 4, NO. 1, JANUARY 2014

Fig. 1. Framework of the 0.34-THz wireless link.

802.11b/g protocol to validate the character of future THzWLAN. It has realized 6.536-Mb/s real-time data transmissionover 1.15 m, among one access point (AP) and two terminalnodes. To the best of the authors’ knowledge, this is the firstTHz WLAN prototype.

II. SYSTEM ARCHITECTURE OF HIGH-SPEED WIRELESSDATA LINK AND WLAN

A. 3-Gb/s 16QAM Wireless Link

Fig. 1 presents the framework of the 0.34-THz 3-Gb/s wire-less link. The main reason for using a heterodyne transceiverrather than a zero-IF transceiver in the current system is thatthe latter usually has inferior performance. For instance, flickernoise in zero-IF transceiver will interfere with the basebandsignal more significantly when compared with the heterodynetransceiver. It is also difficult to eliminate the dc offset andsecond-order (IM2) and third-order inter-modulations (IM3) inthe zero-IF transceiver.16QAM modulation could improve the spectrum efficiency

and reduce the occupied bandwidth. It brings many advantages,given as follows.1) It enlarges the channel capacity. Frequency-division mul-tiplexing (FDM) is commonly adopted to accommodatemultiple users in a WLAN system. A 10-Gb/s, 16QAMsubband with 3.6-GHz physical bandwidth could sup-port a 100-Gb/s WLAN system (ten subbands) withinthe 40-GHz bandwidth. However, a system adoptingASK modulation whose bandwidth of a 10-Gb/s signalis 17 GHz can only support a 23-Gb/s WLAN within thesame 40-GHz bandwidth.

2) Compared with other high-order modulation schemes,the 16QAM scheme has higher spectrum efficiency than8PSK, but similar E N versus BER performance.

Fig. 2. Receiving front end of the 0.34-THz wireless link.

3) Semiconductor devices with narrow relative bandwidthand higher performance are more accessible and econom-ical. In addition, the measured phase noise in Section V-Aindicates that phase noise will not be a significant factor toinhibit the 16QAM demodulation in the 0.34-THz band.However, the lower output power, imperfect signal quality(spectrum purity or phase noise), and higher receivingnoise still make the realization of 16QAM scheme in the0.34-THz band a remarkable challenge.

As presented in Fig. 1, the 3-Gb/s modulator based ona field-programmable gate array (FPGA) and 3-Gs/s dig-ital–analog converter (DAC) modulates the primal binarysignal. The bandwidth of modulated signal could be calculatedusing

(1)

where is the data rate, is the rolloff factor, and is the mod-ulation order. The rolloff factor is set to 0.4 in current modula-tion. The occupied bandwidth of themodulated 3-Gb/s, 16QAMsignal is 1.05 GHz. The center frequency is 750MHz. The mod-ulated signal is then converted to inter-frequency (IF,15.15 GHz) by the first-stage converter and finally reaches the0.34-THz band by the THz transmitter. The ultimate band ofoperation is 339.425–340.475 GHz 339.95 GHz .Fig. 2 shows the 0.34-THz receiving front end. The trans-

mitter and receiver depend on the subharmonic mixer (SHM)to perform frequency mixing, which is pumped by the -band8 multiplier chain. When the SHM is used as an upconverter

in the transmitter, the maximum output power is 14.4 dBm,as shown in Fig. 3. The output power is relatively low, but stillsufficient for our communication experiment. The variation ofoutput power is 2.3 dB in the 335–345-GHz band. The 1-dBcompression point in 340.0 GHz is 16.6 dBm. The outputpower of the modulated 3-Gb/s signal at the 1-dB compressionpoint is 17.5 dBm. A solid-state power amplifier (SSPA) canfurther push the power level to 10 dBm in the future [10].Compared with fundamental mixer, the frequency of local os-

cillating frequency (LO) for the SHM is only half of the inputTHz wave, which eases the difficulty in generating sufficientLO power. The -band 8 multiplier chain consists of threedoublers, which operate in the 40–45-GHz, 80–90-GHz, and

WANG et al.: HIGH-ORDER MODULATION FOR FUTURE WLAN APPLICATIONS 77

Fig. 3. Output power and downconversion gain of 0.34-THz transceiver.

160–180-GHz bands, respectively. We utilize medium poweramplifiers (MPAs) in the 40–45-GHz and 80–90-GHz bandsbetween multipliers to provide 20 dBm of driving power. Themultiplier chain is driven by a 20.3-GHz dielectric resonanceoscillator (DRO) and provides an output of 9.39 dBm in 162.4GHz. We use a -band variable attenuator to adjust the LOpower for optimizing the conversion loss of SHM. A rectan-gular waveguide H-ladder bandpass filter is connected betweenthe SHM and antenna to reject the low sideband of the upcon-verted signal. A high directional Cassegrain antenna is finallyconnected in front of the transmitter and receiver.The received THz signal is also detected by the SHM. The

conversion gain of the SHM is shown in Fig. 3, which is12.25 dB to 13.75 dB within 335–345 GHz. The corre-

sponding IF range is 10.2–20.2 GHz. The SHM has perfectnoise temperature at room temperature, but also brings signifi-cant conversion loss. The SSB conversion loss shown in Fig. 3is approximately 3 dB higher than published data of similarSHMs. The explanation is that common SHMs usually workin low IF ( 100 MHz) and their conversion loss is measuredthrough black body radiation source. In current systems, theSHM works in high IF frequency 15.15 GHz and ismeasured by the classical RF method. However, the measureddata under the same setting show no significant difference innoise temperature or conversion loss between the two kinds ofSHMs. Due to the high conversion loss of SHM, the IF coaxialcable and IF low-noise amplifier (LNA, 12-17 GHz,

30 dB, 1.6 dB) contribute remarkable noise tothe receiver. The total receiver noise temperature is calculatedby

(2)

where , , and are the noise temperature ofSHM, IF coaxial cable, and LNA, respectively. According tothe -factor measurement using a black-body radiation source,the total DSB noise temperature of the receiver is 5277 K. Thenoise temperature introduced by the IF coaxial cable1.0 dB and the LNA are, respectively, 688 K and 1490 K. TheSHM contributes 3099 K to the front-end.

TABLE IBUDGET OF 0.34-THZ WIRELESS COMMUNICATION LINK

a. The loss introduced by atmosphere is estimated to be 0.5 dB.

The IF is recovered to 750 MHz again by a first-stage con-verter in the receiving front end. The 3-Gb/s, 16QAM demodu-lator resembles the modulator in the hardware structure, whichsamples the IF with a 3-Gs/s analog–digital converter (ADC).The demodulator applies a 32-way parallel demodulation al-gorithm based on the frequency-domain implementation of thematched filter and timing phase correction. We proposed inno-vative timing synchronization, channel equalization, and carrierrecovery algorithms and then validated them in the hardwareplatform. Coding and decoding have been performed to furtherreduce the BER. Section IV provides details of the modulatorand demodulator.Table I shows the link budget. The loss introduced by at-

mosphere attenuation in the 0.34-THz band is estimated to be10 dB/km. As a result, the 50.0-m transmission will suffer from0.5 dB loss. The calculated E N is 25.4 dB, which exceedsthe criteria of general digital demodulation.

B. WLAN Based on IEEE 802.11b/g

Short-range WLAN or WPAN is promising for futureTHz communication. Utilizing sufficient spectrum resources,0.3-THz band could easily support 10 100 Gb/s WLAN. Theatmosphere attenuation is inconspicuous in room environment.The III–V compound semiconductor integrated circuits haveachieved 75-mW output power in the 220-GHz band [11].The output power of IEEE 802.11b/g standard WLAN is100 mW in 2.4 GHz. It is feasible to get comparable powerin the 0.34-THz band in the future. However, two backdropsare still inevitable: First, the IEEE 802.11b/g WLAN occu-pies bandwidth 100 MHz. Nevertheless, THz WLAN willoccupy bandwidth of 1–100 GHz, which requires 10–30 dBhigher output power using similar architecture. Second, theeffective antenna aperture decreases with the increase of carrierfrequency as in

(3)

where is the directivity of antenna and is the wavelength.It can be predicted that, when the carrier frequency increases


Fig. 4. Architecture of 0.34-THz WLAN prototype.

Fig. 5. Schematic of 0.34-THz WLAN node.

Fig. 6. 0.34-THz WLAN node and detail of the diplex transceiver.

from the -band to 0.34 THz, the effective aperture decreases40 dB. Therefore, people must use high-gain antennas in a THzWLAN. Mechanical tunable or phase-array antennas may bepossible candidates for this.Fig. 4 shows the architecture of the 0.34-THz WLAN proto-

type in this paper, which consists of one AP and two terminalnodes. The AP and nodes adopt 25-dBi wideband rectangularhorns with a 7–8 3-dB main lobe beam width. The nodesA and B exchange data by relaying AP rather than a directwireless connection. Fig. 5 gives the schematic of 0.34-THzWLAN node. Considering that high-speed WLAN protocolis unavailable and incompatible for the current prototype, theMAC layer and partial physical layer are established through acommercial IEEE 802.11b/g wireless module, which operatesin 2.4–2.485 GHz. The 0.34-THz front-end is designed towork at diplex mode. A duplexer splits the RF signal from thecommercial WLAN module to the 0.34-THz transceiver. Theup-converting and receiving SHMs are pumped by LO signalfrom the same 20.3-GHz DRO. The isolation between upwardand downward links is provided by the directivity of the twoparallel 25-dBi horn antennas. Fig. 6 is the integrated 0.34-THzWLAN node and detail of the diplex transceiver. According totheoretical calculation, the transceiver could achieve E N of15.9 dB in 3.5-m distance.

Fig. 7. Circuit of the 0.34-THz SHM based on SBDs.

III. CRITICAL COMPONENTS FOR 0.34-THZ TRANSCEIVER

A. 0.34-THz SHM

A 0.34-THz SHM acts as signal generator and detector inthe transceiver. The circuit of the SHM is shown in Fig. 7. TheSHM is based on quartz substrate microstrip line, which is em-bedded in a metal waveguide cavity. The thickness of the sub-strate is 50 m and the width is 250 m. The Au layer, with4- m thickness, acts as conductor of the strip line. The circuitconsists of two individual substrates. Anti-parallel Schottky bar-rier diodes (SBDs) provide current–voltage (I–V) nonlinearityfor the SHM, which are bonded on the main quartz substrate.The LO signal provided by a 8 multiplier chain is coupled tothe main substrate by a -band waveguide probe. The receivedRF signal is also coupled by a waveguide probe and transferredto diode. An RF rejection LPF prevents the propagation of RFsignal towards LO and IF ports. The IF signals are filtered by aseventh-order high–low impedance filter in the auxiliary quartzsubstrate. A 18- m gold wire connects the main and auxiliarysubstrates. The pattern of gold wire is carefully optimized tointroduce sufficient inductance to isolate the LO and IF signal.Another 18- m gold wire is soldered near the RF probe toprovide dc grounding for SBDs. The adopted circuit structurebrings the advantage of broad RF and IF bandwidth.The optimization of the SHM is based on a precise 3-Dmodel

of Schottky barrier diodes (SBDs). The parameters of SBDs areas follows: total capacitance 7.5 to 10.5 fF; forwardturn-on voltage ; serial resistance ;ideality factor ; saturated current 1.12610 A. The I–V nonlinearity of the SBDs is determined by theSchottky junction. The peripheral semiconductor structures, in-cluding the cathode pad, the anode finger, the cathode pad, GaAssubstrate, and internal layers, bring parasitic parameters. Thesehave prominent influence on high frequency performance. The3-D model combines a nonlinear time-domain lumped diodemodel to analog I–V behavior and a 3-D frequency-domain elec-tromagnetic model based on finite-element method to analog RFbehavior. We have validated this 3-D model in a -band SHM[16]. The difference between simulation and measurement iswithin 0.5 dB.Fig. 8 shows the picture of the 0.34-THz SHM, optimized to

work in 10–20 GHz IF frequency. It also gives the measured


Fig. 8. 0.34-THz SHM and measured SSB conversion loss 168 GHz .

single side band (SSB) conversion loss. The data is takenfrom back-to-back measurement, rather than the thermal noisemethod. In back-to-back measurement, we connect RF portsof two SHMs in the opposite direction. One SHM works as upconverting and the other as down converting. The conversionloss of the two SHM is assumed to be the same. The SSBconversion loss could then be calculated through the total lossbetween two SHMs. This method reflects the performance ofSHM more directly from the viewpoint of the final application.However, the standing wave of the RF port is rough, whichbrings the ripple shown in Fig. 8.When the SHM is pumped by LO in 162.5, 168, and 173 GHz,

respectively, the lowest SSB conversion loss is 9.67, 9.78, and10.15 dB, respectively. The 3-dB IF bandwidth is 31, 41, and41 GHz. The difference between simulation and measurement iswithin 2 dB. Besides the ripple, the difference incorporateserrors in simulation, fabrication and measurement. When theSHM is used in 10–20 GHz IF band, the SSB conversion lossincreases to 12–13 dB. The measured lowest double side band(DSB) noise temperature of the SHM by the Y factor method is900–1100 K at the 1.0-GHz IF point.

B. 0.17-THz Multiplier Based on a Schottky Varactor

The -band 8 multiplier chain provides the LO signal topump the SHM,which consists of three doublers and twoMPAs.The final stage is a 0.17-THz doubler based on Schottky var-actors. Compared with varistors, varactors have better capa-bility of enduring higher driving power and better conversionefficiency.The 0.17-THz doubler and the measured gain compression of

the final multiplier chain are shown in Fig. 9. The multiplier isbased on 127- m quartz substrate microstrip line. The varactorchip combines four junctions in parallel. The gold band, witha width of 100 m, is bonded between the varactor chip andthe copper cavity to realize electrical grounding. It also helpsto disperse the heat from diodes. The input driving signal is80–90 GHz, 100 mW, with an output range of 160–180 GHz.The measured output power is 9.39 dBm at 162.4 GHz. Thecorresponding conversion efficiency is 8.69%.The gain slope of the multiplier chain is steep. When the

variation of primal driving signal is 3 dB, the variation of finaloutput power achieves 10 dB. It leads to difficulty in preciselytuning the optimal LO power level for the SHM. For instance,the temperature of the devices will increase successively whenthe dc bias is turned on. It introduces slight variation in thedriving signal, but approximate 1 dB variation in the final

Fig. 9. The 160–180-GHz doubler based on Schottky varactor and measuredgain compress of 8 multiplier chain in 162.4 GHz.

Fig. 10. 0.34-THz rectangular waveguide H-ladder bandpass filter and mea-sured -parameter.

LO. This leads the LO level deviating from the optimal value.Ultimately, the issue is solved by introducing a -band variableattenuator connected between the -band doubler and -bandMPA. The initial section is then driven to saturated status, whichis more stable. The LO level is then turned by the -band at-tenuator. The -band attenuator also helps to ameliorate thestanding wave and uniformity of the chain.

C. 0.34-THz Low-Insertion-Loss Bandpass Filter

Conventional terahertz instruments adopt quasi-optical meshfilters or photo crystal grids. The advantages are low insertionloss and flexibility in system tuning. However, due to the simpleresonant structure, the stopband rejection of mesh filter is im-perfect. The volumes of these filters are bulky and seeminglyunable to be integrated compactly with the front end. We de-veloped a 0.34-THz H-ladder bandpass filter based on a metalrectangular waveguide, as shown in Fig. 10.The filter adopts classical fourth-order H-ladder bandpass

structure. The conventional choice for the resonators in filter isTE101 mode. However, it brings bad tolerance on the errors ofmechanical fabrication. Instead, this filter adopts the resonatorsin TE102 mode, which could ease the difficulty. To compensatefor the high dispersion of the transmission line in Terahertzfrequency, we use the following equation is used to slightlyshift the center frequency of designed filter:

(4)

where is wavelength of desired center frequency and andare the wavelengths in upper and lower edges of pass band.

The filter is fabricated by high speed CNC mechanical milling.The entire cavity is split to up and down cavities along the center


Fig. 11. 0.34-THz Cassegrain Antenna and the measured pattern.

of the rectangular waveguide. The precision of fabrication ismeasured to be 2.5 m. Subsequently, they are connectedthrough the pinpointing of metal pins with a gap of 10 m.The simulated data shows that the center frequency is

341.4 GHz; the 3-dB bandwidth is 19.2 GHz; the minimumin-band return loss is 18.7 dB; the rejection at 325 GHz is betterthan 30.2 dB. The data measured by a vector network analyzer(VNA) indicates that the filter is centered at 343.4 GHz; the3-dB bandwidth is 18.6 GHz; the lowest insertion loss is1.41 dB; the minimum in-band return loss is 11.43 dB; below327.1 GHz (4.75% offset from the center frequency), thestopband rejection is better than 30 dB.The difference between the simulation and measurement is

mainly caused by errors in milling and gilding. It also should beindicated that the -parameter is measured with a 325–500-GHzVNA frequency extension. The interface of this instrument is aWR2.2 flange, which causes small mismatch with this filter’sWR2.8 flange. Practical results will be better than presenteddata.

D. 0.34-THz High-Gain Cassegrain Antenna

A high-directivity antenna is helpful for far field transmis-sion. Horn antennas are prevailing in Terahertz range, but high-directional horn antennas suffer from higher conductor loss.The Cassegrain antenna has been proven to be effective in the0.14-THz band [17]. A 0.34-THz Cassegrain antenna has beendeveloped in this paper as shown in Fig. 11. The physical aper-ture of initial feed horn is 3.0 mm 2.5 mm. The diameter ofsecond reflector is 19 mm. The diameter of the main paraboloidreflector is 160 mm.With the increase of working frequency, theprecision of fabrication and roughness of paraboloid surface arecritical to the antenna. Optical interference has been utilized tomonitor the fabrication errors and polish the paraboloid.In this application, the Cassegrain antenna is assembled with

the 0.34-THz front-end and posited on a tripod, which wouldprovide pinpointing precision of . The measuredantenna pattern is also shown in Fig. 11. It should be indicatedthat standard antenna is unavailable in 0.34-THz. The measuredantenna gain is calculated from the measured link loss. In thesetting of measurement, two 0.34-THz Cassegrain antennaswere used to transmit and receive the signal in a distance of50.0 m. The transmitter, whose output power has been cali-brated by an Erickson PM-4 power meter, provides a single toneat 340.0 GHz to the transmitting antenna. A receiver whoseconversion loss has been calibrated in the laboratory detects the

received signal from receiving antenna. The main lobes of thetwo antennas pinpoint each other precisely by careful tuningthe direction of tripod and monitoring the received signal.Subsequently, the transmission loss between waveguide flangesof two antennas is measured. We calculate the total antennagain by subtracting the transmission loss with the loss of freespace channel propagation and the atmosphere attenuation. Wethen average the total antenna gain to get single antenna gain.Following this strategy, the measured main lobe gain of the twoCassegrain antennas is 48.4 dBi in 340.0-GHz point. The 3-dBbeam width is 0.39 . The rejection between the main lobe andother lobes is better than 20.0 dB. The second lobe is 0.62away from the main lobe.

IV. 3-Gb/s MODULATOR AND DEMODULATOR PLATFORM

A. 3-Gb/s 16QAM Parallel Modulator

Considering the speed of digital devices (FPGA, ADC, DAC,and DSP), the complexity of algorithms and the feasibility ofextension in future, 3-Gb/s modulation based on digital pro-cessing is incompatible with classical serial processing architec-ture. In the current system, a 3-Gb/s modulator adopts 32-wayparallel 16QAM modulation architecture. Digital IF modula-tion, high-speed shaping filtering, and quadrature up convertingstructure are integrated in processing algorithms. The algorithmstructure of a 3-Gb/s modulator is shown in Fig. 12(a).The procedure of 3-Gb/s modulation could be described

as follows. First, the serial binary bits of 2.8 Gb/s are con-verted to 32-way parallel data flow with 87.5-Mb/s single-waydata rate through serial-to-parallel converter. Second, parallelReed–Solomon RS(255,239) coding is performed on the par-allel data flow. The efficiency of coding is 93.3%. Every 239bits are coded to 255 bits. The output coded flow is 3 Gb/s.Third, the coded flow is mapped to 32-way parallel I/Q dataflow with a 93.75 Mbit/s single way data rate and then filteredby parallel shaping filter with four times the sample rate. Third,shaped data will experience digital up converting. Eventually,the modulated signal will be sampled by 3-Gs/s high-speedDAC. The final IF output signal is centered at 750 MHz, withan occupied bandwidth of 1.05 GHz.High-speed parallel shaping filters and digital quadrature up

converting are the critical techniques in this architecture. In thehigh-speed parallel shaping filter algorithm, the shaping filter isbased on a square-root raised cosine filter. It can be decomposedto four multi-phase slave filters. The slave filters are further de-composed through a parallel FIR filtering algorithm, based onan iterative short convolution. In the digital quadrature up con-verting, parallel implementation is performed through the de-sign of parallel low-speed NCO. The hardware architecture ofthe 3-Gb/s modulator is shown in Fig. 13(a), which includes thepower module, clock module, DAC, FPGA, and peripheral in-terface. The photograph of the modulator is shown in Fig. 14(a).

B. 3-Gb/s 16QAM Parallel Demodulator

Compared with modulation, digital demodulation confrontsmore challenges. The distortion caused by atmosphere (dust,water vapor, molecule absorption, and atmosphere onflow).Also, the nonlinearity of the transmitter (gain compression of


Fig. 12. Algorithm structure of the 3-Gb/s 16QAM (a) modulator and (b) demodulator.

Fig. 13. Hardware architecture and photograph of the 3-Gb/s 16QAM (a) modulator and (b) demodulator.

Fig. 14. Photographs of 3-Gb/s 16QAM (a) modulator and (b) demodulator.

SHM, phase noise of LO, IM3, and group delay) will intro-duce phase and amplitude imbalance to the signal. Moreover,although the signal is transmitted through the 50.0-m LOS

channel, the received is relatively low due to the addi-tion of Gaussian white noise. As a result, we have proposed a32-way parallel demodulator based on the frequency-domainimplementation of a matched filter and timing phase correction,as described in [18].The architecture of demodulator is shown in Fig. 12(b). The

received 3-Gb/s IF signal ( 750 Msymbol/s,located at 750 MHz) will be sampled by a 3-Gs/s ADC with10-b resolution and then written to 32-way parallel FIFOsat 93.75-Mb/s single-way data rate. A mixer-free down con-verter and overlap operation are then performed. In this step,we realize timing frequency offset correction through indexcontrolled read operation of the FIFOs. Subsequently, afterthe 64-point digital Fourier transition (DFT), the data will bemultiplied by the DFT of the 33-point square-root raised cosine


(SRRC) matched filter (MF) and then multiplied by a rotator tocarry out timing phase correction (TPC). In the next step, weperform a 64-point inverse digital Fourier transition (IDFT).The demodulator transfers the middle 32 point outputs of IDFTto O&M timing error detector (TED). The timing synchroniza-tion algorithm in current architecture adopts a dual feedbackstructure. After the timing error is calculated by O&M TED, thetiming phase offset is then fed back to realize TPC. The timingfrequency offset is fed back to control the reading index ofFIFOs, based on a delete–keep algorithm. Channel equalizationis indispensible to overcome amplitude and phase distortion. Aparallel adaptive blind equalization module PCMA will processthe eight parallel symbols from O&M in the next step. Thismodule is based on a relaxed look-ahead pipelined paralleladaptive CMA equalization algorithm. The demodulator thenperforms a new phase and frequency detector (PFD)-basedparallel DFPLL structure for carrier synchronization. Eightparallel symbols are sent to the DFPLL for carrier recovery.I/Q outputs are then sampled to determine the constellationand data bits. Finally, the demodulator performs RS (255,239)decoding. Parallel-to-serial converter outputs 2.8-Gb/s serialdata flow.Systematic simulation indicates that the BER will be lower

than , while the is better than 14.3 dB withoutdata coding. The hardware architecture of the 3-Gb/s demod-ulator is shown in Fig. 13(b). The photograph is shown inFig. 14(b). The demodulator consists of one ADC10D1500chip and two V6LX240TFF1156 FPGAs. It integrates a PCI-ehigh-speed interface on broad to transfer binary data betweendemodulator and personal computer. We calculate the constel-lation of the demodulated data by the algorithm on broad. Thestatistic of BER is performed from peripheral instruments.

V. MEASURED LINK PERFORMANCE

A. 50.0-m Far-Field Transmission

The 0.34-THz transmission experiment is shown in Fig. 15.The transmission distance is 50.0 m, measured by a laserrange finder. The 0.34-THz transmitter outputs a 340.0-GHzsingle-tone signal, given by an 800-MHz IF single-tonesignal, which will be transmitted and finally detected by thereceiver. Fig. 16 shows the contrast of phase noise betweenthe primal 800-MHz IF single-tone signal and the received800-MHz IF single-tone signal. For the primal signal, therespective phase noise in 100-Hz, 1-kHz, 10-kHz, 100-kHz,and 1-MHz offset frequency are 106.2, 117.5, 120.8,118.3, and 125.4 dBc/Hz. For the received signal, the

phase noise in the above offset frequency are 53.6, 68.0,76.3, 77.1, and 93.7 dBc/Hz. The degradation of phase

noise is 31.7 52.6 dB. The increase of phase noise is partiallyaccredited to the frequency multiplying in transceiver, whichwill raise phase noise by . is the index ofmultiplying, which equals 16 in the current system. Thus, thephase noise would raise 24.1 dB by multiplying. Other parts ofthe increase in phase noise may be accredited to the phase noiseof the LO power chain and first-stage IF converter. In addition,

Fig. 15. Photograph of the 0.34-THz, 50-m, 3-Gb/s transmission experiment.

Fig. 16. Measured phase noise descending of a 800-MHz single toneGHz transmitted through 0.34-THz communication system.

we observe spurs in 117.8-Hz, 166.3-Hz, and 4.69-kHz offset,which have not been found in the primal signal.To evaluate the influence of phase noise, we calculate the

BER, determined by the phase noise of the received signal andbased on the phase-power spectral density (PPSD) [19].PPSD is gotten from single sideband power density spectrum

. The relationship between the two quantities is givenby

(5)

Using linear approximation, we get the rms phase variancefrom (6). The rms phase variance is calculated based on

the received phase noise in Fig. 16.We divide the received phasenoise curve to linear approximation with five straight lines. Thesymbols is the start and stop frequency of each straightline. The symbol represents the slope of straight line in phasenoise spectrum as

(6)

The calculated is 4.172 . In the 16QAM scheme, theallowed phase error is 16.9 , as shown in Fig. 17. The BER


Fig. 17. Allowed phase error of 16QAM scheme.

Fig. 18. Spectrum of primal and received 3-Gb/s 16QAM IF signal after50.0-m transmission (RF is centered at 339.95 GHz).

introduced by phase noise could be obtained based on theprobability of Gaussian distribution, as in

(7)

The calculation tells us that the BER introduced by phase noiseis , regardless of other distortions from RF channeland digital signal processing.After being corrected by decoding, the BER introduced by

phase noise will decrease significantly. The corrected BER isshown in Fig. 21. The original BER is between to

and is determined by complex factors.The primal 3-Gb/s IF spectrum and received 3-Gb/s IF spec-

trum after 50.0-m transmission are shown in Fig. 18. Theof primal signal from the 3-Gb/s modulator is better than 40 dB.The measured 99.99% power bandwidth is 1.051 GHz. The gaincompression and IM3 raise accessory shoulder in the receivedspectrum, which is 20 dBc lower than the main signal.Fig. 19 gives the measured relationship among the ,

the received and transmitted RF power. The maximum receivedis 23.8 dB with the output power of 17.5 dBm. The

corresponding received power is 51.4 dBm. The minimum re-ceived is 9.8 dB under 32.4-dBm output power. Thecorresponding received power is 66.2 dBm. The data is ob-tained by changing the IF input power of the SHM in the trans-mitter. The integration bandwidth in measurement is 1.1 GHz.

Fig. 19. Relationship between , received and transmitted power.

Fig. 20. Measured Constellation under different .

The measured is 1.6 dB lower than the link budget inTable I, which is mainly caused by the error of antenna gain bypointing, the uncertainty of atmosphere attenuation, and the un-certainty of instrument.If the RF power of transmitter is set to relatively higher level,

the system holds better . However, the quality of signalwill decrease due to gain compression. If the RF power de-creases, the will be lower, but the system linearity willbe improved. Fig. 20 shows the demodulated constellation ofthe 16QAM signal under different , which helps us tointerpret this relationship. When the transmitter works in max-imum output power 17.5 dBm, the correspondingreceived 23.8 dB. The constellation tends to con-verge to the center, which brings with it remarkable error bits

. As the decreases to 20.6 dBby reducing the output power of transmitter, the constellationconverges to the four phases and four amplitudes (16 points).The interference between nearby points is inconspicuous. Thecorresponding BER is . When the falls to12.9 dB, the constellation will present acute dispersion and therelative BER degrades to .The measured BER under different is shown

in Fig. 21. The lowest BER is while18.6 dB. When the is lower than 16.8 dB,

the BER will increase as the decreasing of by 0.826 dBper decade. At the lowest 9.8 dB, the BER reaches

. Generally speaking, a communication system withis sufficient in pragmatic application. The

minimum required is 13.8 dB, while .A cascaded MMIC amplifier will help us to achieve higher


Fig. 21. Measured BER under varied .

Fig. 22. Photograph of the 0.34-THz WLAN experiment.

with tolerable distortion in the future. Only rudimentaryRS (255,239) coding is integrated in current system; a morecomplex interleaved code will improve the performance of thesystem.

B. 0.34-THz WLAN Based on IEEE 802.11

Fig. 22 shows the 0.34-THz WLAN experiment. The0.34-THz WLAN AP is put in the opposite side of the0.34-THz WLAN nodes A and B. As a result, the main lobe ofthe rectangular horn antennas of the AP could cover the lobeof horns in nodes. Two PCs are connected to nodes A and B.The AP could identify the wireless nodes with IEEE 802.11b/gprotocol and act as an agent to relay the data between nodes Aand B. The system operation is stable. In the experiment, a datafile is transmitted from the computer of node A to that of nodeB by relaying of AP. The measured distance of transmissionis 1.15 m. The transmission data rate is 817 kByte/s (or 6.536Mbit/s), within 30 min. The BER of the WLAN prototype hasnot been measured. Because the protocol of the IEEE 802.11b/ghas a strong correction mechanism to detect error bits and cor-rect them, through coding and repeating transmission, whichensures no error bits, eventually.

VI. CONCLUSION

In this paper, a 0.34-THz wireless data link is presented. Thewireless data link is based on a super heterodyne transceiver and16QAM parallel digital modulation. A 0.34-THz transceiver

TABLE IISUMMARY OF RECENTLY PUBLISHED MMW/THZ COMMUNICATION LINKS

consisted by SHM, multiplier chain, band pass filter and highdirectivity antenna, is described. The algorithm and hardwareplatform of a 3-Gb/s 16QAM modulator and demodulator arealso elaborated. The experimental data indicates that the linkhas achieved a lowest BER of as transmitting a3-Gb/s signal over 50.0 m. Compared with the other publishedTHz communication research in Table II, the presented worksuccessfully realizes high-order modulation above 0.3 THz.Both the transmission distance and spectrum efficiency arebetter than those achieved in other works.Furthermore, a 0.34-THz WLAN prototype is realized. De-

spite the fact that the physical layer of the prototype (trans-ceiver, modulation and coding) has been completed, the mediaaccess and control (MAC) layer and high-speed communicationprotocol are still unexplored. We hope to develop a 0.34-THzWLAN that will combine both low-frequency wireless area net-work and high-speed THz data link in the future.

ACKNOWLEDGMENT

The authors would like to thank C. Zhou, J. Yao, W. Su,S. Xiao, X. Kang, P. Chen, B. Cui, and J. Jiang of CAEP-IEEfor their support of this work.

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Cheng Wang was born in Suining, China, on March8, 1987. He received the B.S. degree in engineeringphysics from Tsinghua University, Beijing, China,in 2008, and the M.S. degree in radio physics fromChina Academy of Engineering Physics, Mianyang,China, in 2011.He joined the Institute of Electronic Engineering,

China Academy of Engineering Physics, Mianyang,China, in 2011. His current research involvesmillimeter-wave//terahertz communication system,mixers and multipliers based on Schottky diodes,

and terahertz passive waveguide components.

Bin Luwas born in Chongqing, China, on August 18,1985. He received the B.S. degree in electronic en-gineering from Fudan University, Shanghai, China,in 2004, and the M.S. degree in communication andinformation system from China Academy of Engi-neering Physics, Mianyang, China, in 2011.He has been an Engineer since 2011 with the

Institute of Electronic Engineering (IEE), ChinaAcademy of Engineering Physics (CAEP), Mi-anyang, China, where he is engaged in researchon terahertz communication and radar systems,

millimeter-wave passive filters and digital processing algorithms.

Changxing Lin was born in Chongqing, China, onJanuary 7, 1986. He received the B.S. degree in engi-neering physics and Ph.D. degree in nuclear scienceand technology from Tsinghua University, Beijing,China, in 2007 and 2012, respectively.He was a Research Assistant from June 2009 to

December 2009 with the European Organization forNuclear Research (CERN). He joined the Instituteof Electronic Engineering, China Academy of Engi-neering Physics, Mianyang, China, in 2012. His cur-rent research involves algorithm and implementation

of high-speed demodulation for communication and terahertz wireless local areanetwork.

Qi Chen was born in Chongqing, China, onNovember 3, 1981. He received the B.S. degree inremote sensing techniques and instrument and M.S.degree in electromagnetic and microwave techniquesfrom Xidian University, Xian, China, in 2003, and2007, respectively. He is currently working towardthe Ph.D. degree in radio physics at the ChinaAcademy of Engineering Physics, Mianyang, China.He joined the Institute of Electronic Engineering,

China Academy of Engineering Physics, Mianyang,China in 2007. His research interests include mmW/

Terahertz antennas, photo crystal and meta-materials.

LiMiaowas born inMianyang, China, on September27, 1986. She received the B.S. and M.S. degreesin electromagnetic and microwave techniques fromSouthwest Jiao Tong University, Chengdu, China, in2009, and 2012, respectively.She joined the Institute of Electronic Engineering,

China Academy of Engineering Physics, Mianyang,China, in 2012. Her research includes terahertz non-linear circuits and passive components.

Xianjin Deng was born in Anyue, China, on June11, 1973. He received the B.S. degree in electronicengineering from Xidian University, Xian, China, in1998, and the M.S. degree in electronic science andtechniques fromUniversity of Electronic Science andTechnology of China, Chengdu, China, in 2006.He joined the Institute of Electronic Engineering,

China Academy of Engineering Physics, Mianyang,China, in 2003. His current research involves mil-limeter-wave (mm-wave)/terahertz communicationsystem, mm-wave solid-state power combining, and

microwave active circuits.

Jian Zhang was born in Sichuan, China, onNovember 27, 1968. He received the B.S. degree inelectronic techniques from the National Universityof Defense Technology, Changsha, China, in 1989,the M.S. degree in communication engineeringfrom the China Academy of Engineering Physics,Mianyang, China, in 1994, and the Ph.D. degree inelectrical engineering from Chongqing University,Chongqing, China, in 1998.He joined the Institute of Electronic Engineering,

China Academy of Engineering Physics, Mianyang,China, in 1989. his current research involves electronic systems, wireless com-munication, and terahertz science and technology.

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