mesh patch antenna report
TRANSCRIPT
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CHAPTER 1
Introduction
1.1 Background Vehicles are becoming mobile electronic communication systems, part of a wider telematics
network with applications at microwave and millimeter wave frequencies. Many low frequency
antennas below 1 GHz are printed on glass screens in the motor industry to reduce costs, hide the
antennas and protect them from vandalism. Microstrip patches are widely used as cheap,
conformal antennas for a wide variety of higher frequency applications and so there is currently
much interest in printing such antennas on, or within, the glass areas of vehicles for intelligent
transport and telematics systems. References [1], [2] reported on the performance of patch
antennas fixed directly to glass which formed a superstrate. Mounting antennas within the glass
offers the prospect of reducing costs but presents production problems such as thermal distortion
of the glass during processing and feeding the signal to the embedded antenna. In addition it is
not possible to print a solid conductor area on glass if it exceeds a few millimeters across as the
metal area reflects heat and distorts the glass during the shaping/lamination process. In that case
the metal must be meshed.It is not the intention for this seminar to look solely at glass based
applications but to present some interesting general properties of meshed antennas that are not
reported in the literature. Discussion on square antennas where the patch and ground plane were
meshed in various combinations and their effects of the varying line widths and line density on
gain, cross-polarization, resonant frequency and bandwidth. The antennas were fed using coaxial
probes but coplanar feeds can also be used.
1.2 Relevance The topic is related to Antenna designing which is also a part of our curricular. Here, traditional
patch antennas are described, later on which are meshed to form a antenna which is transparent
The transparency of the antenna can be varied but it adds to a significant change in the
parameters of the antenna.
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1.3 Literature Survey 1.3.1 Dipole Antenna A dipole antenna is a radio antenna that can be made of a simple wire, with a center-fed driven
element. It consists of two metal conductors of rod or wire, oriented parallel and collinear with
each other (in line with each other), with a small space between them. The radio frequency
voltage is applied to the antenna at the center, between the two conductors. These antennas are
the simplest practical antennas from a theoretical point of view. They are used alone as antennas,
notably in traditional "rabbit ears" television antennas, and as the driven element in many other
types of antennas, such as the Yagi. Dipole antennas were invented by German physicist
Heinrich Hertz around 1886 in his pioneering experiments with radio waves. Dipoles have an
radiation pattern, shaped like a toroid (doughnut) symmetrical about the axis of the dipole. The
radiation is maximum at right angles to the dipole, dropping off to zero on the antenna's axis.
The theoretical maximum gain of a Hertzian dipole is 10 log 1.5 or 1.76 dBi. The maximum
theoretical gain of a λ/2-dipole is 10 log 1.64 or 2.15 dBi.
Fig: 1.1 Electric field and magnetic field radiated by the Dipole.
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Fig:1.2 Radiation Pattern of a Diploe Antenna
1.3.2 Fractal Antenna Benoit Mandelbrot, the pioneer of classifying this geometry, first coined the term ‘fractal’ in
1975 from the Latin word derived from the Latin fractus meaning broken, uneven. Any of
various extremely irregular curves or shape that repeat themselves at any scale on which they are
examined. A fractal antenna is an antenna that uses a fractal, self-similar design to maximize the
length, or increase the perimeter (on inside sections or the outer structure), of material that can
receive or transmit electromagnetic radiation within a given total surface area or volume.
Such fractal antennas are also referred to as multilevel and space filling curves, but the key
aspect lies in their repetition over two or more scale sizes, or "iterations". For this reason, fractal
antennas are very compact, multiband or wideband, and have useful applications in cellular
telephone and microwave communications. A good example of a fractal antenna as a space
filling curve is in the form of a shrunken fractal helix. Here, each line of copper is just a small
fraction of a wavelength. A fractal antenna's response differs markedly from traditional antenna
designs, in that it is capable of operating with good-to-excellent performance at many different
frequencies simultaneously. Normally standard antennas have to be "cut" for the frequency for
which they are to be used—and thus the standard antennas only work well at that frequency. This
makes the fractal antenna an excellent design for wideband and multiband applications. For
example Mandelbrot discusses the basics of fractal theory as applied to the characteristics of a
coastline. The length of a coastline depends on the size of the measuring yardstick. As the
yardstick we use to measure every turn and detail decreases in length, the coastline perimeter
increases exponentially. As the view of a coastline is brought closer, we discover that within the
coastline there lie miniature bays and peninsulas. As we examine the coastline on a rescaled
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map, we discover that each of the bays and peninsulas contain sub-bays and sub-peninsulas.
There is a self-similar trait observed as we look at the coastline at various resolutions. The
number of microscopic structures begins to approach infinity. In fact, because of the large
number of irregularities, the physical length of a coastline is virtually infinite.
Fig: 1.3 Coast Line
Fig.1.3 represents an imaginary coastline. The grey lines are rulers being used to measure
the length of the coastline (L). These rulers are of the length S. Using the first ruler we see that it
L = 2 * S. When we decrease the length of S the number of times that S is used increases. What
these rulers illustrate is that as the size of the measuring device becomes smaller the accuracy of
the measurements becomes more and more accurate. From this fact we can assume that
eventually we will be able to get an exact measurement of the coastline. This statement is false.
As we decrease the size of the measuring device the length that we have to measure becomes
greater. We can see this by zooming in on the coastline. As we get closer and closer we will
notice that it looks very similar to how it looked from a greater distance away. Only now we are
much closer. This observation shows the self-similarity of the coastline. Therefore as we
decrease the size of the measuring device the length of the coastline will increase without limit,
thus showing us its fractal nature.
1.4 Motivation With the growing number of telematics systems used in cars there is an increased need for the
integration of antennas into the structure. Microstrip patch antennas which are lightweight and
low in cost can be integrated into different parts of the car body. Integrating the patch antenna
into the windscreen poses practical problems:
It is not possible to screen print large solid metal areas (e.g. a patch at 1.5 GHz) because this
would distort the windscreen during the heating process by which it is formed.
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This problem is overcome by replacing the solid metal areas of the patch by a mesh structure
which gives the additional advantage that the antenna gains a degree of transparency. Meshing
antenna although increases its transparency; it also changes its resonant frequency, Gain and
other parameters severely. Hence there is need to briefly discuss the effects meshing on the
above parameters.
1.5 Scope The Meshed antenna designed can be used in vehicular applications. As the Antenna is
transparent it can be used for some of the more complex applications, which includes embedding
these antennas on the solar panel.
1.6 Organization of Report Chapter 2 starts with the theory of Microstrip Patch antennas, it illustrates what are the
microstrip antennas how do they radiate, the different shapes available for microstrip patch
antennas. Chapter 3 explains different feeding methods for microstrip antennas along with
appropriate illustrations along with their equivalent circuit diagrams. Chapter 4 explains the
designing of microstrip antennas later on the designed antenna will be meshed to form Mesh
Patch Antenna, also we will discuss the effects on Gain, resonant frequency, by changing
meshing parameters such as line spacing, line width etc.
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CHAPTER 2
Microstrip Antenna
2.1 Introduction Microstrip antennas received considerable attention starting in the 1970s, although
the idea of a microstrip antenna can be traced to 1953 and a patent in 1955. Microstrip antennas,
as shown in Figure 2.1 consist of a very thin (t << λ0, where λ0 is the free-space wavelength)
metallic strip (patch) placed a small fraction of a wavelength (h << λ0, usually 0.003 λ0 ≤ h ≤
0.05 λ0) above a ground plane. The microstrip patch is designed so its pattern maximum is
normal to the patch (broadside radiator). This is accomplished by properly choosing the mode
(field configuration) of excitation beneath the patch. End-fire radiation can also be accomplished
by judicious mode selection. For a rectangular patch, the length L of the element is usually
λ0/3 < L < λ0/2. The strip (patch) and the ground plane are separated by a dielectric sheet
(referred to as the substrate), as shown in Figure 2.1. There are numerous substrates that can be
used for the design of microstrip antennas, and their dielectric constants are usually in the range
of 2.2 ≤ εr ≤ 12. The ones that are most desirable for good antenna performance are thick
substrates whose dielectric constant is in the lower end of the range because they provide better
efficiency, larger bandwidth, loosely bound fields for radiation into space, but at the expense of
larger element size. Thin substrates with higher dielectric constants are desirable for microwave
circuitry because they require tightly bound fields to minimize undesired radiation and coupling,
and lead to smaller element sizes; however, because of their greater losses, they are less efficient
and have relatively smaller bandwidths. Since microstrip antennas are often integrated with other
microwave circuitry, a compromise has to be reached between good antenna performance and
circuit design. Often microstrip antennas are also referred to as patch antennas. The radiating
elements and the feed lines are usually photo-etched on the dielectric substrate. The radiating
patch may be square, rectangular, thin strip (dipole), circular, elliptical, triangular, or any other
configuration.
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a) Microstrip Antenna
b) Side View of Microstrip Antenna
Fig: 2.1. Microstrip Antenna and its Side view
These and others are illustrated in Figure 2.2. Square, rectangular, dipole (strip), and circular are
the most common because of ease of analysis and fabrication, and their attractive radiation
characteristics, especially low cross-polarization radiation. Microstrip dipoles are attractive
because they inherently possess a large bandwidth and occupy less space, which makes them
attractive for arrays. Linear and circular polarizations can be achieved with either single elements
or arrays of microstrip antennas. Arrays of microstrip elements, with single or multiple feeds,
may also be used to introduce scanning capabilities and achieve greater directivities.
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Fig: 2.2. Shapes of Microstrip patch Elements
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CHAPTER 3
Feeding Methods There are many configurations that can be used to feed microstrip antennas. The four most
popular feeding methods are the microstrip line, coaxial probe, aperture coupling, and proximity
coupling. These are displayed in Figure 3.1. One set of equivalent circuits for each one of these
is shown in Figure 3.2. The microstrip feed line is also a conducting strip, usually of much
smaller width compared to the patch. The microstrip-line feed is easy to fabricate, simple to
match by controlling the inset position and rather simple to model. However as the substrate
thickness increases, surface waves and spurious feed radiation increase, which for practical
designs limit the bandwidth (typically 2–5%). Coaxial-line feeds, where the inner conductor of
the coax is attached to the radiation patch while the outer conductor is connected to the ground
plane, are also widely used. The coaxial probe feed is also easy to fabricate and match, and it has
low spurious radiation. However, it also has narrow bandwidth and it is more difficult to model,
especially for thick substrates (h > 0.02 λ0). Both the microstrip feed line and the probe possess
inherent asymmetries which generate higher order modes which produce cross-polarized
radiation. To overcome some of these problems, non-contacting aperture-coupling feeds, as
shown in Figures 3.1 (c ,d) have been introduced.
(a) Microstrip Line feed (b) Probe Feed
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(c) Aperture Coupled Feed
(d) Proximity Feed Coupled
Fig: 3.1 Typical Feeds for microstrip Antenna
The aperture coupling of Figure 14.3(c) is the most difficult of all four to fabricate and it also has
narrow bandwidth. However, it is somewhat easier to model and has moderate spurious
radiation. The aperture coupling consists of two substrates separated by a ground plane. On the
bottom side of the lower substrate there is a microstrip feed line whose energy is coupled to the
patch through a slot on the ground plane separating the two substrates. This arrangement allows
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(a) Microstrip Line (b) Probe
(c) Aperture-coupled (d) Proximity Coupled
Fig: 3.2 Equivalent Circuits for Typical Feeds
independent optimization of the feed mechanism and the radiating element. Typically a high
dielectric material is used for the bottom substrate, and thick low dielectric constant material for
the top substrate. The ground plane between the substrates also isolates the feed from the
radiating element and minimizes interference of spurious radiation for pattern formation and
polarization purity. For this design, the substrate electrical parameters, feed line width, and slot
size and position can be used to optimize the design. Typically matching is performed by
controlling the width of the feed line and the length of the slot. The coupling through the slot can
be modelled using the theory of Bethe, which is also used to account for coupling through a
small aperture in a conducting plane. Of the four feeds described here, the proximity coupling
has the largest bandwidth (as high as 13 percent), is somewhat easy to model and has low
spurious radiation. However its fabrication is somewhat more difficult. The length of the feeding
stub and the width-to-line ratio of the patch can be used to control the match .
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CHAPTER 4 Transmission line Model
The rectangular patch is by far the most widely used configuration. It is very easy to analyze
using both the transmission-line and cavity models, which are most accurate for thin substrates.
The transmission-line model is the easiest of all but it yields the least accurate results and it lacks
the versatility. However, it does shed some physical insight. the cavity model, a rectangular
microstrip antenna can be represented as an array of two radiating narrow apertures (slots), each
of width W and height h, separated by a distance L. Basically the transmission-line model
represents the microstrip antenna by two slots, separated by a low-impedance Zc transmission
line of length L.
(a) Microstrip Line
(b) Electric field Lines
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(c) Effective Dielectric Constant
Fig: 4.1 Microstrip line, its electric field lines, and effective dielectric constant geometry.
4.1 Fringing Effects Because the dimensions of the patch are finite along the length and width, the fields at the edges
of the patch undergo fringing. This is illustrated along the length in Figures 2.1(a,b) for the two
radiating slots of the microstrip antenna. The same applies along the width. The amount of
fringing is a function of the dimensions of the patch and the height of the substrate. For the
principal E-plane (xy-plane) fringing is a function of the ratio of the length of the patch L to the
height h of the substrate (L/h) and the dielectric constant εr of the substrate. Since for microstrip
antennas L/h>> 1, fringing is reduced; however, it must be taken into account because it
influences the resonant frequency of the antenna. The same applies for the width. For a
microstrip line shown in Figure 4.1(a), typical electric field lines are shown in Figure 4.1(b).
This is a nonhomogeneous line of two dielectrics; typically the substrate and air. As can be seen,
most of the electric field lines reside in the substrate and parts of some lines exist in air. As
W/h >> 1 and εr >> 1, the electric field lines concentrate mostly in the substrate. Fringing in
this case makes the microstrip line look wider electrically compared to its physical dimensions.
Since some of the waves travel in the substrate and some in air, an effective dielectric constant
εreff is introduced to account for fringing and the wave propagation in the line. To introduce the
effective dielectric constant, let us assume that the center conductor of the microstrip line with its
original dimensions and height above the ground plane is embedded into one dielectric, as shown
in Figure 4.1(c). The effective dielectric constant is defined as the dielectric constant of the
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uniform dielectric material so that the line of Figure 4.1(c) has identical electrical characteristics,
particularly propagation constant, as the actual line of Figure 4.1(a). For a line with air above the
substrate, the effective dielectric constant has values in the range of 1 < εreff < εr. For most
applications where the dielectric constant of the substrate is much greater than unity (εr >> 1),
the value of εr will be closer to the value of the actual dielectric constant εr of the substrate. The
effective dielectric constant is also a function of frequency. As the frequency of operation
increases, most of the electric field lines concentrate in the substrate. Therefore the microstrip
line behaves more like a homogeneous line of one dielectric (only the substrate), and the
effective dielectric constant approaches the value of the dielectric constant of the substrate.
Typical variations, as a function of frequency, of the effective dielectric constant for a microstrip
line with three different substrates are shown in Figure 14.6.
For low frequencies the effective dielectric constant is essentially constant. At intermediate
frequencies its values begin to monotonically increase and eventually approach the values of the
dielectric constant of the substrate. The initial values (at low frequencies) of the effective
dielectric constant are referred to as the static values, and they are given by,
W/h >> 1
ε𝑟𝑒𝑓𝑓 = 𝜀𝑟+12
+ 𝜀𝑟− 12
+ �1 + 12 ℎ𝑤�−0.5
(4.1)
4.2 Effective Length, Resonant Frequency, and Effective Width Because of the fringing effects, electrically the patch of the microstrip antenna looks greater than
its physical dimensions. For the principal E-plane (xy-plane), this is demonstrated in Figure 14.7
where the dimensions of the patch along its length have been extended on each end by a distance
3L, which is a function of the effective dielectric constant εreff and the width-to-height ratio
(W/h). A very popular and practical approximate relation for the normalized extension of the
length is
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(a) Top View (b) Side View
Fig: 4.2 Physical and effective lengths of Rectangular microstrip antenna
∆𝐿ℎ
= 0.412 �𝜀𝑒𝑓𝑓+0.3� �𝑊ℎ+0.264�
�𝜀𝑒𝑓𝑓−0.3� �𝑊ℎ+0.8� (4.2)
Since the length of the patch has been extended by 3L on ach side, the effective length of the
patch is now (L = λ/2 for dominant TM010 mode with no fringing)
Leff = L + 2∆L (4.3)
For the dominant TM010 mode, the resonant frequency of the microstrip antenna is a function of
its length. Usually it is given by ,
(𝑓𝑟)010 = 12𝐿√𝜀𝑟 �𝜇𝑜𝜀𝑜
= 𝑣𝑜2𝐿 √𝜀𝑟
(4.4)
where 𝑣𝑜 is the speed of light in free space. Since (14-4) does not account for fringing, it must be
modified to include edge effects and should be computed using ,
(𝑓𝑟𝑐)010 = 𝑞 𝑣𝑜2𝐿 √𝜀𝑟
(4.5)
Where ,
𝑞 = (𝑓𝑟𝑐)010(𝑓𝑟)010
(4.6)
The q factor is referred to as the fringe factor (length reduction factor). As the substrate height
increases, fringing also increases and leads to larger separations between the radiating edges and
lower resonant frequencies.
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4.3 Design Based on the simplified formulation that has been described, a design procedure is outlined
which leads to practical designs of rectangular microstrip antennas. The procedure assumes that
the specified information includes the dielectric constant of the substrate (𝜀𝑟) , the resonant
frequency (𝑓𝑟 ), and the height of the substrate h. The procedure is as follows:
Specify: εr , 𝑓𝑟 (in Hz), and h.
Determine: W,L
Design procedure:
1. For an efficient radiator, a practical width that leads to good radiation efficiencies
Is :
W = 12fr�µ0ε0
� 2εr+1
= 𝑣02fr
� 2εr+1
(4.7)
Where, 𝑣0 is the free space velocity of light.
Determine the effective dielectric constant of the microstrip antenna using, ε𝑟𝑒𝑓𝑓
Once W is found, determine the extension of the length ∆L using
The actual length of the patch can now be determined by solving for L, or
𝐿 = 12𝑓𝑟�𝜀𝑟𝑒𝑓𝑓�𝜇0𝜀0
− 2∆𝐿 (4.8)
Calculate for εreff and ∆L/h using the equations (4.1) and (4.2) respectively.
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Fig: 4.3 Circular and square mesh patches with their solid equivalents
The microstrip patch antenna is further meshed as shown in the figure. We are concentrating on
the designing of the Square Patch antenna. Square mesh patches have an overall dimension of ‘a’
with square holes of side ‘c’ spaced apart. If is the number of holes in one direction then the
amount of metal in the meshed patch compared to the solid patch is given by 𝑚 = 1 − 𝑛2𝑐2
𝑎2 and
the number of lines per wavelength by λ by λ𝑐+𝑑
. A reference patch, a conventional solid metal
patch and ground plane printed on RT Duroid substrate with εr = 2.33, was used in all cases for
comparing the measured parameters to provide a consistent benchmark.
4.4 Meshed Patch Over the solid ground plane. A series of meshed patches were manufactured with different line widths and line densities to
establish the basic properties of gain, cross-polarization and resonant frequency. The input
impedance is higher for the meshed patch and so the feed point is closer to the centre. The
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meshed patches were placed over solid ground planes at this stage. The first measurements
examined the effects of changing the line width and the line spacing on gain and cross-
(a) (b)
Fig: 4.4 Current distribution on meshed patch over solid ground plane
Polarization. The measured results for five samples are plotted in Fig. 4.5,where it can be seen
that the gain improves as the line width increases and the spacing decreases, i.e. as the area of
metal increases over the patch. On the other hand thin, widely spaced lines have better cross-
polarization. There is, therefore, a trade-off between gain and cross-polarization for a given
geometry. More work is needed to understand the effect of the meshing parameters on the
bandwidth. In general the bandwidth remained at about 1% for this patch study but variations up
to 0.3% were noted. The resonant frequency reduces as the percentage of metal decreases as
shown in Fig. 4.6, e.g. a meshed patch with side a=65mm, c= 2.5mm , d=0.7mm resonates at
1.37 GHz (52% metal) while the same standard patch antenna unmeshed has a resonance at 1.48
GHz. Hence for a given patch size the resonant frequency goes down as the number of mesh
lines is reduced resulting in a smaller antenna at a given frequency. The relationship was not
linear as the frequency of resonance reduces more quickly when the metal percentage falls below
60% as seen in Fig. 4.6. The effects noted in Figs. 4.5 and 4.6 were investigated further using the
simulation package Momentum. Fig. 4.4 shows the current distribution computed over the
meshed patch in two forms, Fig. 4.4(a) shows the magnitude while Fig. 4.4(b) shows the vector
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Fig: 4.5 Effect of line width and line spacing on measured normalized gain and cross-polarization. (a) Line spacing (b) line width. ——— gain change - - - - -- - cross-polar level.
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Fig: 4.6 Measured resonant frequency of mesh square patch as a percentage of metal content
compared to a solid metal patch.
The feed point is clearly visible. Fig. 4.4(a) shows that current is distributed on each of the
vertical lines of the mesh uniformly whereas for a conventional patch the current density is high
only at the edges of the patch. Note also the high current magnitudes at the edges of each mesh
line in Fig. 4.4(a). These excited mesh lines are closely coupled and the resulting radiation
pattern is similar to that measured for a standard patch. The loss in gain noted in Fig. 4.5 is
mainly accounted for by the conductor losses due to the high currents at the edge of each mesh
line. The current vector diagram in Fig. 4.4(b) shows that the currents flowing from the top to the
bottom of the patch flow into the horizontal conductor lines as well at the junctions with the
vertical lines. The consequence of this is that the current paths are longer and hence the meshed
patch radiates at a lower frequency than a standard patch. Therefore thicker mesh lines give rise
to a lower resonant frequency than thin ones.
4.5 Meshed ground plane The ground planes used in this study were about 2.5 times the size of the patch, resulting in some
radiation diffracted to the rear. An experimental study investigated meshing the ground plane in
a similar way to that of the patches, thus creating a more optically transparent antenna. A square
mesh structure was used for the rectangular patches operating in the fundamental mode. It should
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be noted that using a standard patch over a meshed ground offered no significant benefits.
A number of effects were observed when the ground plane was meshed and combined with a
meshed patch. Meshing the ground plane improved the bandwidth which increased typically
from 0.6% to 1.6% for 25% metallization while the resonant frequency reduced further to 1.21
GHz. Hence the resonant frequency of the standard patch at 1.48 GHz was reduced to 1.21 GHz
for the fully meshed patch, a reduction of 32%.The radiation patterns were most affected as
shown in Fig. 6. The most notable change was in the back radiation which increases inversely
with the density of the mesh. This is because the ground plane effectively leaks radiation through
the mesh, the more holes in the mesh the greater the leakage. The meshing also improves the
cross polarization levels in the forward direction by about 5 dB.
Fig: 4.7 Measured radiation patterns in H plane for rectangular meshed patch with solid and meshed ground planes. Solid ground plane; - - - - Meshed ground plane.
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CHAPTER 5
Applications and Future Challenges
It is widely used in the vehicular data communication application, also implemented in the places
where you want to hide the antenna.
Autonomous communications systems often involve the use of separate solar cells and antennas,
which necessitate a compromise in the utilization of the limited surface area available. These
separate items may be combined thus saving valuable 'real estate', provided that the antennas and
solar cells are compatible. One method for achieving this is to integrate the two kinds of device
on the same element i.e. solar cells are intimately combined with printed antennas, providing a
new device called SOLANT.
Mesh Patch antennas can also be integrated into a car wind screen.
Future Challenges is to increase the transparency of the antenna while maintaining the radiation
properties of the antenna.
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CHAPTER 6
Conclusions
This seminar explains the effects of meshing a Patch antenna and the ground plane. The radiation
patterns are not significantly affected by meshing the patch alone, keeping a solid ground plane,
but the gain suffers by up to 3 dB when compared to a standard patch. In general, as the meshing
gets denser, the gain of the antenna also increases. Bandwidth is a function of meshing space,
bandwidth increases with increase in mesh spacing. However, gain of the antenna reduces with
the increase in the mesh spacing. Meshing the patch lowered the resonant frequency by up to
20% and on meshing the ground plane as well as the patch radiation leaks through the mesh
increasing the radiated fields in the reverse direction dependent on the mesh density. The
resonant frequency drops further and a reduction of 32% was measured. The meshed patch offers
a complex trade-off between parameters but gives opportunities for improving the bandwidth
and reducing the cross polarization and the antenna dimensions at the expense of the gain.
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1905, 1994.
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