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Hindawi Publishing CorporationInternational Journal of Microwave Science and TechnologyVolume 2010, Article ID 980957, 8 pagesdoi:10.1155/2010/980957

Research Article

Noise-Cancelling CMOS Active Inductor and Its Application inRF Band-Pass Filter Design

Santosh Vema Krishnamurthy, Kamal El-Sankary, and Ezz El-Masry

Sexton Campus, Dalhousie University, 1360 Barrington Street, Halifax, Canada NS B3J 1Z1

Correspondence should be addressed to Kamal El-Sankary, [email protected]

Received 6 August 2009; Revised 21 October 2009; Accepted 4 January 2010

Academic Editor: Liang-Hung Lu

Copyright © 2010 Santosh Vema Krishnamurthy et al. This is an open access article distributed under the Creative CommonsAttribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work isproperly cited.

A CMOS active inductor with thermal noise cancelling is proposed. The noise of the transistor in the feed-forward stage of theproposed architecture is cancelled by using a feedback stage with a degeneration resistor to reduce the noise contribution to theinput. Simulation results using 90 nm CMOS process show that noise reduction by 80% has been achieved. The maximum resonantfrequency and the quality factor obtained are 3.8 GHz and 405, respectively. An RF band-pass filter has been designed based onthe proposed noise cancelling active inductor. Tuned at 3.46 GHz, the filter features total power consumption of 1.4 mW, low noisefigure of 5 dB, and IIP3 of −10.29 dBm.

1. Introduction

Inductors, either passive or simulated by active devices arekey components in Radio Frequency Integrated Circuits(RFIC) analog building blocks such as filters [1], oscillators[2], phase shifters [3], and low-noise amplifiers (LNA) [4].As integrated circuit technologies are progressing, the usageof passive inductors is degrading due to their large chip area,low-quality factor, and less tunability. Mainly the inductoris the major chip area consuming building block [5]; thehigher the inductance required, the higher the chip area.While tunability can be implemented using passive inductorand low-resistance switches, continuous tuning is not easilyachieved. Due to these disadvantages, the concept of activeinductors is becoming more attractive.

Many proposed active inductor architectures using MOStransistors [1–5] can be found in the literature. Highernoise, nonlinearity and power consumption are the majordisadvantages of active inductors due to the fact that thesecircuits are realized using active devices which have highernoise. Thermal noise cancelling (NC) has been successfullyimplemented in LNA circuits by cancelling the noise of theirinput transistors [6, 7]. In this paper, NC is applied toactive inductor architecture. First the transistor contributingmost of the noise of the active inductor is identified in

the forward path of Gyrator-C structure, and its noisecontribution is cancelled using the feedback path while thenoise contribution of the latter to the input is degenerated bythe inclusion of shielding resistor in the feedback. NC allowsactive inductors to be used in applications such as front-endactive RF filter in receivers and also to design low-phase noiseoscillators.

The paper is organised as follows. In Section 2, the NCactive inductor is presented, as well as the principle of NCand simulation results. In Section 3, a differential second-order RF band-pass filter based on the proposed NC activeinductor and its performance are demonstrated. Finally, theconclusion is given in Section 4.

2. NC Active Inductor Design

In the active inductor circuit shown in Figure 1(a), M2 isidentified as the transistor with the highest noise contribu-tion.

2.1. Noise Contribution of the Transistor M2. The circuitused to implement NC in active inductor is the Gyrator-C structure proposed in [1] and shown in Figure 1(a). Inorder to apply NC, we studied the noise contribution of each

2 International Journal of Microwave Science and Technology

VDD

I1

3

In,M2

M1 M2 Va1

M3

IbZin

2

I2 In,M2

(a)

02× 109

1

2

3

4

2× 108

Frequency (Hz)

5

6

109 1010108

Node 370

0.5

0.25

0.75

1

1.25

1.5 Node 2

Ou

tpu

tim

peda

nce

toga

inra

tios

atn

ode

2an

d3

(b)

Figure 1: (a) Active Inductor Circuit without NC. (b) The ratio of output impedance to the respective gain at nodes 2 and 3.

transistor and identified the transistor that has the highestpercentage of noise to the total input referred noise. Usuallyin LNA, noise cancelling is applied to the input transistor.This is not the case of active inductors, and due to the back-to-back Gm structure a careful study is needed to identify thetransistor that contributes the highest noise at the frequencyof interest.

The noise current of M2 is modelled as the currentsource In,M2 , between nodes 2 and 3 in Figure 1(a). For betterunderstanding, we have shown the noise ofM2 as two currentsources, one flowing out of node 3 and other flowing intonode 2 [8]. From noise analysis and simulation results, M2

is found to be contributing more than 25% to the inputnoise at high frequency. The reason behind this high noisecontribution is that the noise of M2 will be amplified by thegains from the nodes 2 and 3 when referred to the input.These gains depend on the impedance at nodes 2 and 3divided by the respective gains calculated from input node1 to the nodes 2 and 3. As shown in Figure 1(b), the outputimpedance to the gain ratios go higher as frequency increases.Thus the noise current of M2 multiplied by these two ratiosto be referred to the input goes higher as frequency increases.

The input referred noise from nodes 2 and 3 can becalculated by breaking the feedback from drain of M3 to theinput. The input referred noise contribution from node 3 iscalculated as

In,M2refnode3 =In,M2node3Ro3CL

AI3CL

= In,M2node3

Ro3OL/(1 + AI3OLβ

)

AI3OL/(1 + AI3OLβ

)

= In,M2node3Ro3OL

AI3OL.

(1)

Hence,

In,M2refnode3 = In,M2node3

C2s +(gm1 + gm2 + 1/ro2

)

gm1gm2c1s + gm1gm2. (2)

Similarly, we can obtain the input referred noise from node 2as

In,M2refnode2=In,M2node2

(C2s + ro2rob

(gm1 + gm2 + 1/ro2

))

gm1(1+c1s)(1+rob/ro2 +robc3s+

(gm1 +gm2

)rob) ,

(3)

where In,M2refnode2,3 are the input referred noise currents of M2

from nodes 2 and 3, respectively. In,M2node2,3 are noise currentsof M2 at nodes 2 and 3, Ro3CL is the closed loop resistance atnode 3, Ro3OL is the open loop resistance at node 3, AI3CL isthe closed loop transimpedance gain from node 1 to node 3for the shunt-shunt feedback circuit in Figure 1(a), AI3OL isthe open loop transimpedance gain from node 1 to node 3,and β = gm3 is the feedback factor.

Equations (2) and (3) reveal that the noise contributionof M2 goes higher at high frequencies. This can be under-stood from the fact that the Gyrator structure in Figure 1(a)has shunt-shunt feedback structure and the capability of thisfeedback topology to lower the output impedances at nodes2 and 3 is reduced at higher frequencies due to the feedbackforward gain reduction. That leads to higher contribution ofthe noise sources In,M2node2,3 when multiplied by these outputimpedances at nodes 2 and 3 to be referred to the input atnode 1.

2.2. NC Principle and Simulation Results. Figure 2 illustratesthe modification introduced for cancelling the noise of M2

(Rf is neglected at this time and nodes 1 and 4 are shorted).The noise current of M2, which is modelled as the currentsource In,M2, flows into node 2 but out of node 3 as shown

International Journal of Microwave Science and Technology 3

VDD

I1

3

M3 M4

4

2nd stage

In,M2

M2M1 VaIb In,t

Vin 1

2

I2 In,M2 Rf

Figure 2: Principle of noise cancelling design.

0

0.2

I n,M

2/In

,ou

t

0.4

1

0.6

21.5

Frequency (GHz)

2.5 3 3.5 4 4.5 5 5.5 6 6.5

With NCWithout NC

Figure 3: Simulated In,M2/In,out versus frequency with and withoutNC.

in Figure 2. This creates two fully correlated noise voltagesat nodes 2 and 3 with opposite phases. These two voltagesare converted into current by M3 and M4, respectively. Byproperly matching the AC currents of M3 and M4 aroundthe resonant frequency, the noise contributed by M2 can becancelled at the input.

Conversely, the signal voltages at nodes 2 and 3 are inphase, resulting in constructive addition at node 1. Thecondition for complete noise cancelling of M2 can be derivedas

In, input = In, M2ZO3gm3 − In, M2ZO2gm4 = 0 (4)

=⇒ gm2ro2 + gm1ro2 + C2ro2s

s2C2c3ro2 + sc3ro2(gm2 + gm1

)+ ro2

(gm2 + gm1

)+ gm1

gm3

= ro2robro2robgm1 + ro2robgm2 + sC2ro2rob + ro2 + rob

gm4,

(5)

where ZO2,3 are the total impedance at the nodes 2and 3, ro3,4,b are the output resistances of transistorsM3,4,b, and C1,2,3 are the total capacitances at nodes 1, 2, and3.

0

10

20

pA/s

qrt

(Hz)

30

40

Frequency (Hz)

With M4 ON

With M4 OFF

1010108 2× 108 2× 109109

60

50

Figure 4: Simulated input referred noise with and without NC.

0

Rel

ativ

eco

ntr

ibu

tion

(%)

50

121.5

Frequency (GHz)

2.5 3 3.5 4 4.5 5

With Rf

Without Rf

Figure 5: Simulated noise contribution with and without Rf .

VDD

I1

3

M1 M2 Va1

M3 M4

4

IbZin

2

I2 Rf

Figure 6: Simplified active inductor.

Zin

+GM1

−GM2C

(a)

Zin

RP

Leq

Rs

CP

(b)

Figure 7: (a) Gyrator-C realization of active inductor (b) equivalentpassive model.


V2

gm1∗(V1 −V2)Cgs1

g2 C2 gm1∗V2

V4V1V3

g1 C1

g f

gm3∗V3 gm4

∗V2

g4 C4 g3 C3

Figure 8: Small signal model of the simplified active inductor in Figure 1.

For simplicity, we have ignored few terms in the aboveNC condition in (5). Figure 3 compares the noise contribu-tion of M2 with and without NC to the total output noiseafter the above NC condition is matched relatively at thefrequencies of interest by properly choosing the size andbiasing current of M3 and M4. We can see from Figure 3 thatthe noise contribution from M2 is reduced at all frequenciesand it is almost contributing 0% around the resonancefrequency.

To further confirm the effect of noise cancelling, the casewith M4 being turned OFF is simulated while maintainingthe same power consumption by the following steps: (1)connecting the gate of M4 to VDD; (2) Va is adjustedaccordingly to increase the bias current ofM2; (3) the biasingcurrent of M3 is adjusted to give the same power dissipationas M4 is turned ON. The simulated total input referred noisecurrent when M4 is turned ON/OFF is shown in Figure 4. Itshould be noticed from this graph that the noise has beenreduced by 80% when M4 is ON (with NC).

Even after cancelling the noise of M2, we can still furthersuppress the noise of the second stage in Figure 2. We can seethat the entire noise of the second stage is directly added tothe input noise as is the case in every back-to-back Gyratoractive inductor. To further optimize the active inductor forlower noise, the resistor Rf is added to degenerate the noisein,t of the entire 2nd stage as shown in Figure 2. Even thoughRf will introduce noise to the input but the benefit of itsinclusion on the noise reduction overshadows the effect ofits added noise. The noise contributed by the second stageto the input while Rf is included can be calculated as below.Figure 2 can be considered as shunt-shunt feedback structurewhereM1–4 are considered to be in feed-forward loop and Rf

in the feedback loop.The noise contribution of the second stage can be

obtained as

In,input = In,tZo4CL

AI4CL

= In,tZo4OL/

(1 + AI4OLβ

)

AI4OL/(1 + AI4OLβ

)

= In,tZo4OL

AI4OL,

(6)

In,input =In,t gm2ro2

Rf A1(gm2gm3ro2 + gm4

) ≈ In,t

R f A1gm3, (7)

where In,t is the total noise current of the entire 2nd stage,Zo4CL, AI4CL are the closed loop impedance and gain at node4, Zo4OL, AI4OL are the open loop impedance and gain at node4 and A1 is the gain contributed from M1 and M2. From (7),it is evident that increasing Rf shields the input node of theactive inductor from the noise contribution of the secondstage formed byM3,M4, and Ib. Figure 5 shows the simulatednoise contribution of the second stage when compared withand without Rf . Figure 5 shows the noise contribution ofsecond stage to the total input referred noise. Here we cansee that by adding Rf , the noise contribution of 2nd stagehas been reduced to 15% from 45%.

2.3. Active Inductor Parameters. In this section, the differentparameters of the proposed active inductor are derived. Thesimplified active inductor is shown in Figure 6. The Gyrator-C principle on which active inductor is based on is depictedin Figure 7(a) and its equivalent passive model is shownin Figure 7(b). The differential pair M1 and M2 forms thepositive transconductance Gm1 with input node at 1 andoutput nodes at 2 and 3. The transistor pairM3 andM4 formsthe negative transconductance −Gm2 with input nodes at 2and 3 and output node at 1. These two Gm blocks convertthe parasitic capacitors at nodes 2 and 3 to an equivalentinductor appearing at node 1. The ideal current sources inthe active inductor are implemented using simple currentmirror.

The equivalent Zin can be obtained from the smallsignal analysis circuit in Figure 8. For better intuition, a fewparasitic parameters have been ignored. The approximateexpression for Zin is as follows:

Zin(s) ≈ s/C1 + g3/C1C3

s2 + sA + B, (8)

where A denotes (gm4gm1/GmC1)(g4g f /(g4 + g f )) + g1/C1 +g3/C3 + g1g3C2/GmC1C3 − ω2(C2/Gm)) and B denotes((gm1gm2gm3 +gm1gm4g3)/GmC1C3)(g f /(g4 +g f ), and gm1–4 are


−1252× 109

−100−75−50−25

2× 108

Frequency (Hz)

025

109 1010108

(Zin phase)

5075

100

60

40

80

100

120

160

140

(Zin magnitude)

(deg

)(d

B)

Figure 9: Simulated frequency response.

−3

−2

−1Imag

(E9)

0

1

Real (E9)

2.5

Pole-zero analysis

−2.5 0−10 −5−7.5−12.5−15

3

2

Pole = “1 : Rf = 3.75 K”;

Pole = “0 : Rf = 0”;Pole = “3 : Rf = 11.25 K”;

Pole = “4 : Rf = 15 K”;Pole = “2 : Rf = 7.5 K”;

Figure 10: Pole-zero plot for variation of Rf .

the transconductances of transistors M1–4, C1–3, g1–4 are thetotal parasitic capacitances and conductances at nodes 1–4,g f is the conductance of feedback resistor,Gm = gm1+gm2+g3,and the cubic term in the denominator is substituted to−ω2s, because the effect of negative real pole arising dueto cubic term is of second order [1] when compared todominant conjugate poles.

From (8), Zin is equivalent to the input impedance ofthe RLC passive network shown in Figure 7(b). The passiveequivalent elements are obtained as

Rs =Gmg3

(gm1gm2gm3 + gm1gm4g3

)(g f /(g4 + g f

)) , (9)

where Rs is the series resistor with Leq and the parallelparasitic capacitor Cp are given by

Leq =GmC3

((g f + g4

)/g f)

gm1gm2gm3 + gm1gm4g3,

Cp = C1.

(10)

The resonant frequency is given by

ω0 =√√√√

gm1gm2gm3 + gm1gm4g3

GmC1C3

((g f + g4

)/g f) ,

f0 = 1

2π√LeqC1

.

(11)

The quality factor at the resonant frequency is obtained as

Q0 =√

B

A. (12)

The simulated frequency response of the active inductoris shown in Figure 9. The simulations are carried out in90 nm STM CMOS process. The transistor sizes (W/L in μm)are M1 (6/0.18), M2 (6/0.18), M3 (11.5/0.18), M4 (2.3/0.18),I1 = 280μA, Ib = I2 = 400μA, Va = 0.8 V, and VDD =1.2 V. The active inductor resonates at centre frequency off0 = 3.8 GHz with a quality factor of Q0 = 405. Thepole-zero analysis performed using Spectre simulator showsthat all the poles reside on the left side of the jω axis andconsequently the active inductor is stable. Also varying thefeedback resistor Rf (from 0 to 30 kΩ) and gm1 and gm2 (byvarying the active inductor biasing currents I2 and Ib from300 μA to 500 μA) was applied to confirm the stability ofthe active inductor in case of tuning its center frequencyand quality factor. Figure 10 shows the zero-pole plot for theactive inductor transfer function while changing Rf from 0to 15 kΩ. Moreover, transient analysis has been performed toconfirm the stability.

Table 1 compares the performance of the proposed activeinductor with the previously published active inductors [1, 9,10]. Comparison of performances proves that this inductorhas a good quality factor, wide inductive bandwidth, low-power consumption of 1.2 mW, and very less noise whenintegrated over 500 MHz bandwidth.

3. RF Band-Pass Filter and Simulation Results

Figure 11 shows the proposed pseudo-differential RF Band-pass filter implemented using proposed NC active inductor.The input stage is designed by the common-gate transistorMin which is biased by Iin and Rin. A source follower stage isused as an output buffer for output matching and to reducethe loading effect.

The effect of NC is high lightened by comparing theperformance of the two band-pass filters. One designed usingthe active inductor with NC and other using the activeinductor without NC (the transistor M4 is turned OFF asmentioned above). In Figure 12, we can see that when the


VDD VDD

IinI1

34M3M4

Ib Va M2 M11

2

Rf

Vout−Vout+

MoutMout

Vin+ Vin−

Rs

M1 M2 IbVa

3M4M3

4

I1

1

Iin

Vin

RoutRout

Rin Rin

Vc R f

2

I2MinMinVcI2

Figure 11: RF Band-pass filter based on NC Active inductor.

−30

Gain with NC Gain without NC

Same gain

−20

2× 109

Frequency (Hz)

3× 109 4× 109 1010109

−10

0

10

20

30

Gai

n(d

B)

(a)

−30

Gain without NC Gain with NC

−20

2× 109

Frequency (Hz)

Same quality factor

3× 109 1010109

−10

0

10

20

30

40

4× 109

50

Gai

n(d

B)

(b)

0

NF with NC NF without NC

5

2× 109

Frequency (Hz)

4× 109

Noise figure

3× 109 1010109

10

15

20

25

30

NF

(dB

)

(c)

5

NF without NC NF with NC

4× 109

10

2× 109

Frequency (Hz)

Noise figure

3× 109 1010109

15

20

25

30

NF

(dB

)

(d)

Figure 12: NF and gain comparison for band-pass filter with and without NC active inductor.


4

NF (actual)NF (mismatch)NF (mismatch)

5

2× 109

Frequency (Hz)

3× 109 1010109

6

7

8

9

10

11

12

13N

F(d

B)

(a)

−5

Gain (actual)Gain (mismatch)Gain (mismatch)

0

2× 109

Frequency (Hz)

3× 109 1010109

5

10

15

20

25

30

35

Gai

n(d

B)

(b)

Figure 13: NF and frequency response with process variation.

gain and centre frequency of the proposed band-pass filterare same, the NF without and with NC are 17 dB and 5 dB,respectively. And also we can see that when the 2 filters aretuned to have same quality factor and centre frequency, theNF without and with NC are 17 dB and 5.7 dB, respectively.

Figure 13 shows the filter NF and frequency responseunder extreme cases of combined process and mismatchvariations. We can see that the NF has changed by almost0.75 dB around the centre frequency while the gain of thefilter has been changed without much variation in the centrefrequency. The gain of the filter can be retrieved by tuning Qusing the biasing current I2 in Figure 11.

The linearity of the band-pass filter was examined byplotting the third-order intercept point (IIP3) using periodicsteady state simulation as shown in Figure 14. The conditionsfor simulating the linearity are center frequency f0 =3.467 GHz and the two signals injected were at frequencies of3.48 GHz and 3.49 GHz, respectively. While noise cancellingreduces the noise floor, the harmonic distortion is notreduced as a result of NC conditions, which leads to almostconstant dynamic range for the filter with and withoutNC. However, the harmonic distortion can be cancelled byselecting different conditions other than the one used for NC,therefore, simultaneous cancellation of noise and distortionis hampered [13]. Despite the fact that this filter presents anacceptable linearity, the latter can be improved by increasingthe overdrive voltages of transistorsM1–4 at the cost of higherpower supply. Moreover, distortion cancelling techniques

−100

−80

−60

Y0

(dB

m)

−40

−20

prf (dBm)−10

Input referred IP3 = −10.2926

1st order frequency = 3.48 G3rd order frequency = 3.49 G

−15−20−25

Trace = “3rd order”; ipn curvesTrace = “1st order”; ipn curves

20

0

ep = −22

Figure 14: IIP3 simulation result.

such as the one presented in [14] can be adapted to cancelthe distortion of the proposed active inductor and that toimprove the overall linearity of the designed RF filter.

Table 2 compares the performance of the band-pass filterusing the proposed active inductor with previously publishedRF band-pass filters. Comparison of performances provesthat this filter features lower-power consumption of 1.4 mWand lower noise figure of 5 dB at the filter centre frequency.


Table 1: Comparison of active inductor performances.

Parameter Reference [1] Reference [9] Reference [10] This work

Technology 0.20 μm/1.8 V 0.13 μm/1.2 V 0.35 μm/1.5 V 90 nm/1.2 V

Inductive bandwidth n.a. 300 MHz–7.32 GHz 6.8 MHz–2.97 GHz 600 MHz–3.8 GHz

L (nH) 29 38–144 30.9 165–530

QLmax n.a. [email protected] GHz 434@1 GHz 120@3 GHz

QR 665 61 n.a. 405

Pdis (mW) 4.4 1 0.6 1.2

Noise 0.8 μV/√

Hz 69.31 μV∗ 91.3 μV 43.37nV∗

∗Calculated.

Table 2: Comparison of band-pass filter performances.

Parameter Reference [1] Reference [11] Reference [12] This work

Technology 0.20 μm 0.18 μm 0.18 μm 90 nm

Filter order 2 2 5 2

ω0 (GHz) 5.4 2.44 3.45∼3.6 3.46

Pdis (mW) 4.4 10.8 28 1.4

NF (dB)25.6 18 14 5

(at 5.7 GHz) (at 2.45 GHz) (at 3.6 GHz) (at 3.46 GHz)

IIP3 (dBm) −13.9 — −2.4 −10.29

4. Conclusion

A 3.8 GHz noise cancelling CMOS active inductor is pre-sented. The proposed active inductor has low noise due tonoise cancelling and resistive degeneration. The circuit haswide inductive bandwidth, high-resonant frequencies withhigh-quality factor. Simulation results are provided in 90 nm1.2 V STM CMOS process. A second-order band-pass filterhas been designed based on this proposed active inductor.The designed filter including the input and output bufferstages has low NF and low-power consumption which makesit suitable for front-end RF filtering applications.

References

[1] H. Xiao and R. Schaumann, “A 5.4 GHz high-Q tunableactive-inductor bandpass filter in standard digital CMOStechnology,” Analog Integrated Circuits and Signal Processing,vol. 51, no. 1, pp. 1–9, 2007.

[2] H. Xiao and R. Schaumann, “A low-voltage low-power CMOS5 GHz oscillator based on active inductors,” in Proceedingsof the 9th International Conference on Electronics Circuits andSystems, vol. 1, pp. 231–234, September 2002.

[3] M. Abdalla, G. V. Eleftheriades, and K. Phang, “A differential0.13 μm CMOS active inductor for high-frequency phaseshifters,” in Proceedings of the IEEE International Symposiumon Circuits and Systems, pp. 3341–3344, 2006.

[4] D. K. Shaeffer and T. H. Lee, “A 1.5 V 1.5 GHz CMOS low noiseamplifier,” IEEE Journal of Solid-State Circuits, vol. 32, no. 5,pp. 745–759, 1997.

[5] A. Pascht, J. Fischer, and M. Berroth, “A CMOS low noiseamplifier at 2.4 GHz with active Inductor load,” in Proceedingsof the IEEE Topical Meeting on Silicon Monolithic IntegratedCircuits in RF Systems, pp. 1–5, 2001.

[6] L. Chih-Fan and L. Shen-luan, “A broadband noise-cancellingCMOS LNA for 3.1–10.6 GHz UMB receivers,” IEEE Journal ofSolid-State Circuits, vol. 42, no. 2, pp. 329–339, 2007.

[7] B. Federico, A. M. K. Eric, and B. Nauta, “Wide-band CMOSlow-noise amplifier exploiting thermal noise cancelling,” IEEEJournal of Solid-State Circuits, vol. 39, no. 2, pp. 275–282, 2004.

[8] B. Razavi, Design of Analog CMOS Integrated Circuits, TataMcGraw-Hill, New Delhi, India, 2001.

[9] H. Ugur Uyanik and N. Tarim, “Compact low voltage high-Q CMOS active inductor suitable for RF applications,” AnalogIntegrated Circuits and Signal Processing, vol. 51, no. 3, pp. 191–194, 2007.

[10] A. Thanachayanont and S. S. Ngow, “Low voltage high-QVHF CMOS transistor-only active inductor,” in Proceedingsof the IEEE International Midwest Symposium on Circuits andSystems, vol. 3, pp. 552–555, 2002.

[11] Z. Gao, J. Ma, M. Yu, and Y. Ye, “A fully integrated CMOSactive bandpass filter for multiband RF front-ends,” IEEETransactions on Circuits and Systems II, vol. 55, no. 8, pp. 718–722, 2008.

[12] K.-H. Liang, C. C. Ho, C.-W. Kuo, and Y.-J. Chan, “CMOS RFband-pass filter design using the high quality active inductor,”IEICE Transactions on Electronics, vol. E88-C, no. 12, pp. 2372–2375, 2005.

[13] F. Bruccoleri, A. M. Klumperink Eric, and B. Nauta, WidebandLow Noise Amplifiers Exploiting Thermal Noise Cancellation,vol. 840 of The Springer International Series in Engineering andComputer Science, Springer, New York, NY, USA, 2005.

[14] W.-H. Chen, G. Liu, B. Zdravko, and A. M. Niknejad, “A highlylinear broadband CMOS LNA employing noise and distortioncancellation,” IEEE Journal of Solid-State Circuits, vol. 43, no.5, pp. 1164–1175, 2008.

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