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DesignCon 2014
Quantitative EMI Analysis of
Electrical connectors Using
Simulation Models
Michael Rowlands, Molex
Alpesh Bhobe, Cisco Systems
Patrick Casher, Molex
Xiao Li, Cisco Systems
Abstract
Radiated energy is a problem that grows in complexity as data rates increase. In
addition, EMI problems often show up late in the system verification process, near to
system product shipping deadlines. The solutions to these EMI problems are very costly
and difficult to implement. Hence it is always good to capture the potential EMI
problems via simulation and analysis during the product design phase, rather than during
EMC regulatory measurements towards the end of the product development. In addition,
simulation techniques for EMI are often complicated and time consuming, and also not
suited for broadband analysis.
This paper describes a method of using 3D field solver tools to analyze radiated
energy across a range of frequencies. A 3D field solver model is run and the s-
parameters are generated across a range of frequencies. The initial solve point is used to
generate quantitative results for radiated energy. Then just the initial solve is re-run at a
various frequencies, chosen based on interesting points from the s-parameter results. The
initial solve completes quickly so that multiple points can be used to generate radiated
energy results in a range of frequencies. The method is then used to analyze the EMI
performance from a few connector structures and compared to lab measurements.
Various features are then compared concerning their effect on EMI.
Author(s) Biography
Michael Rowlands is an Electrical Engineer in the Signal Integrity and Connector
Design group at Molex. He specializes in signal integrity at multi-gigahertz frequencies.
He received a Bachelor's and Master's degree in Electrical Engineering from MIT in
1998. Upon graduation, he worked four years as a signal integrity engineer at Teradyne
in Boston. He designed cable assemblies, circuit boards and interconnects for test
equipment up to 6 GHz. In 2002, he worked at a startup company in Illinois. The
company designed dispersion compensation microchips at 12.5 Gbps for fiber-optic
communications. He designed circuit boards to demonstrate and verify 12.5Gbps
performance and made algorithm improvements based on system modeling. He has
authored or co-authored and presented technical papers at ECTC, DesignCon, IMAPS,
IPC-APEX and PCB East. In 2005, as part of the Research and Development at Endicott
Interconnect Technologies years he designed and analyzed circuit boards, chip packages
and custom computing systems. Since 2009, he's worked at Molex designing next-
generation 25-40Gbps I/O and board-to-board connectors.
Alpesh U. Bhobe received his Ph.D. in Electrical Engineering from the University of
Colorado at Boulder, Colorado in 2003. He was a Post-Doc at NIST in Boulder, Colorado
from 2003-2005. While at the University of Colorado and at NIST his research interest
included the development of FDTD and FEM code for EM and Microwave applications.
Currently, he is working as a Manager in EMC Design Cisco Systems, San Jose, CA.
Patrick Casher received his BSEE and MSEE in Electrical Engineering from Illinois
Institute of Technology in 1991 and 2006 respectively. The concentration of study for
his graduate work was in the areas of Numerical Methods, Electromagnetics and High-
Speed Circuit design. He joined Molex Incorporated in 2001 and he is currently the
Signal Integrity Engineering Manager. His team is responsible for the development of
high-speed IO connectors and cages from a signal integrity and shielding effectiveness
standpoint. He holds a least a dozen patents in the fields of high-speed connector and
shield design.
Xiao Li is a master student of EMC lab of Missouri University of Science and
Technology (MS&T) where his study focuses on the analysis and measurement for EMC
and SI problems. He received his first master degree from Beijing Jiaotong University,
China. Prior to moving to the US in 2005, he was working as EMC research at the EMC
lab of the National Institute of Metrology in Beijing, China. Currently he is working at
CISCO as an intern in EMC team focusing on EMC simulation, troubleshooting and
measurement tool development.
Section I
Introduction
Electromagnetic Interference/ Electromagnetic Compatibility (EMI/EMC) is a
concern for any electronic system, especially at multi-Gbps data rates. EMI/EMC
problems are often found late in the build phase of a project which makes problems
expensive and difficult to fix. The goal of this study is to establish a manageable
simulation method for analyzing broadband EMI effects and to use it for quick EMI
analysis , with various sources and structures.
EMI from backplane connectors have been studied via simulations and
measurements before. However, a comprehensive method to analyze radiation up to 40
GHz does not exist it literature. In [1] the study focuses on radiation mechanism and
radiation affecting factors for a generic backplane connector up to 10 GHz. The paper
also discusses techniques to reduce radiated emissions. A common-mode current and a
near-field probe measurement technique were introduced in [2]. The method was applied
to evaluate the EM1 performance of an open-pin-field module-on-backplane connector
up to 3 GHz.
The outline of this paper is as follows. In Section II we introduce the simulation
methodology to analyze the EMI performance of backplane connections from 1 – 18
GHz. Section III explains in details the measurement setup used to analyze the Total
Radiated Power from connectors. The section also discusses the methodology used to
correlate simulation results to measurements. In section IV we present EMI radiation
analysis from different connectors types based on the proposed simulation method.
Section II
Simulation Method The electromagnetic simulation method to analyze radiation from PCB connectors
must be sufficiently accurate to replicate real life EMI issues. Additionally, in order to be
practical, the EMI simulation method must generate results in the order of hours or a few
days. The simulation method introduced in this paper begins by using a 3D full wave
solver to calculate EM fields & Total Radiated Power in the frequency range of 1 – 18
GHz. Then, using equation 1, the solver frequency sweep generates s-parameters that are
used to calculate all energy that doesn’t come out of simulation ports. The field solver
antenna parameters are then used to generate a ratio of dissipated- to-radiated energy.
This ratio is used along with the s-parameter data to calculate radiated energy and
dissipated energy at the field-solve frequency. The s-parameter data over frequency
shows resonance peaks and anti-peaks. These frequency points are noted. The initial
field solve is re-run at various frequencies, without sweeping frequencies. Specific
frequencies are chosen to include the peak resonance and anti-resonances noted in the
broadband s-parameter energy. Additional frequencies to solve fields are spaced out
through the relevant band, by choosing a near DC frequency, and then choosing regular
intervals. At each discrete frequency, the fields are solved and the antenna parameters
calculated. The final result is a set of curves vs. frequency for energy lost in the
structure, energy dissipated and energy radiated. The final results curves give a
quantitative look at energy radiated vs. energy dissipated. In a structure, using a Network
Analyzer, the energy going in and coming out can be readily measured by s-parameters.
The energy that doesn’t transmit, where does it go? Knowing the radiated vs. dissipated
energy ratio tells exactly where the energy goes.
In order to establish this broadband EMI simulation method, the simulation results must
be correlated to measured data. In this study, measurements are made using a
Reverberation chamber, a single-ended co-axial input, a horn probe and a loop probe as
shown in Fig. 1. In order to correlate simulation and measurement results, efforts are
made to change the measurement geometry to be more like the simulation, and vice
versa. The structures analyzed include typical connector configurations and typical circuit
board configurations. When energy goes into a structure, and doesn’t come out, it often
assumed the radiated energy is the lost energy, minus the typical differential insertion
loss of the structure. The sources analyzed include a pure differential signal, pure
common mode, skew sources which include common and differential energy, plus a
ground path with energy on the entire path. These experiments will compare radiated
energy in various structures, each with a range of sources. Conclusions are drawn from
the data, to identify what structures and sources cause the biggest radiated energy. Also,
the results are compared to the s-parameter data to see what, if any, correlation there
exists between s-parameters and radiated energy. The s-parameter to radiated energy
relation is analyzed to see if s-parameters are sufficient for understanding EMI, or if
detailed information on the 3D structure is needed for accurate EMI analysis.
The method to simulate radiated power, using the HFSS 3D field-solver, begins with the
mesh settings. The mesh on the rectangular radiation boundary box is set to a length
based limit of wavelength/12 for 40GHz in air. This forces sufficient mesh on the
boundary box to get accurate radiated power results. The first solution is done at a high
frequency and then the frequency sweep is completed to get broadband s-parameter
results. The s-parameter results, especially, return and insertion loss, are examined to
find any resonances in the structure. These resonance frequencies are noted so that
subsequent radiated power solution will include may frequency points near the
resonances.
After the simulation is complete, the antenna parameters are calculated and the accepted
and radiated power is recorded at the mesh solution frequency. A convenient way to
generate radiated power results, for multiple frequencies, is to use dependent solves.
After the first solution is complete, dependent solution points are generated at every
frequency of interest. The dependent solution points re-use the mesh from the first
solution, and are set to run one more adaptive pass. Analyzing all the dependent solves
overnight, the mesh solution antenna parameters are generated and plot to make a curve
of radiated power vs. frequency.
A check is done on the simulation results are generating using a 3D electro-magnetic
field solver. Here are the settings used in HFSS to produce accurate results:
HFSS 3D field solver settings to simulate radiated power:
- In order to generate sufficient accuracy, set the maximum mesh length on the
radiation boundary, to 1/12 of the wavelength at the mesh solution frequency.
- Run a full solve at a multi-GHz solution frequency, such as 40GHz, and complete
the simulation sweep. Look at differential and common mode loss and note any
resonance frequencies.
- Edit the waveport sources. A single-ended 1 mW source, (0.321623V across 50
ohms), with all other ports terminated to 50 ohms is convenient because it
matches the lab setup. In HFSS, “Incident voltage” and “Include Post Processing
Effects” must be turned on to match exactly with the lab 1 mW source.
- At the solution frequency, generate the accepted power and radiated power
antenna parameters using a far-field, infinite sphere setting.
- One way to get a curve of radiated power vs. frequency after solving the model
for the first time, is to disable the frequency sweep and add dependent solves at
each frequency of interest, which point to the initial solved mesh and run one
more adaptive mesh. Solving all these setups takes many hours and is typically
run overnight.
In order to verify the accuracy of the antenna parameters and understand the radiated
power calculation, a field calculator was used on the solved 3D model. The accepted
power from the far-field antenna parameters represents all the power that does not come
out of the ports of the model. Therefore, the accepted power equals the input source
power times (1 minus energy from all ports):
����������� = ���������� × [| ��|� +| ��|
� +⋯+| ��|�] 1
The radiated power equals all the power passing through the radiation boundary surface.
The field calculator is used to generate radiated power by integrating the pointing vector
on the surface of the radiation boundary.
���� =∮� ×�∗ ∙ � 2
The accepted power minus the radiated power equals all the power dissipated in the
model and not coming out of the ports of the model. This quantity is generated in the
field calculator by integrating the conductor loss on all the conductive surfaces of the
model, then adding all the dielectric loss in all the non-conductive objects in the model.
In [4-5] the antenna parameters are compared to the calculated results. These results
show good agreement between the antenna parameters and the results generated manually
with the field calculator. Henceforth, the antenna parameters will be used to quickly
generate radiated power and dissipated power in a model.
Table 1 Antenna Parameters Compared to Field Calculator Results
Frequency 30 GHz
Source Voltage 0.316228 V
Zo 50 ohms
Source power 0.001 Watts
Antenna incident power 0.001 Watts
1 – {power out of all ports} 0.0002922 Watts
Antenna accepted pwr 0.000298 Watts
Field calculator radiated rower 0.000223 Watts
Antenna radiated power 0.000221 Watts
Field calculator conductive loss 5.34E-05 Watts
Field calculator dielectric loss 2.65E-05 Watts
Field calculator total dissipated power 7.99E-05 Watts
Antenna dissipated power (acc – rad) 7.70E-05 Watts
A ground structure driven with a small source is used to simulate a system with some
noise of the ground path. The source impedance of that ground structure is set to a low
number, such as 0.1ohms and the source voltage is set to 14mV to create a source power
of 1mW/ 0dBm.
Section III
III. 1: Measurement Procedure In order to correlate simulation and measurement, both the simulation and the lab
measurement must be setup carefully, with the environments matching as closely as
possible. This is done with a simulation-measurement-simulation method. The
simulation is done first, noting the various settings of the simulation and getting familiar
with the performance results. Then the measurement is done, with an effort to replicate
the simulation settings. Inevitably, the measurement equipment cannot do exactly the
same things as the simulation. The exact measure settings are noted and then brought
back into the simulation. The final simulation is done to reproduce the measurement
results as closely as possible.
The measurements of radiated power are done using a reverberation chamber as shown in
Figure 1. The equipment used is ETS-Lundgren SmarTIMM. The 1-18GHz horn is used
as the receiver is the ~2 meters x 1 meter x 1 meter chamber. The network analyzer
makes the measurements, which are done by driving the device with a signal, and
measuring the radiated energy with the receiver antenna. Cables connect the network
analyzer to the device, through an amplifier to boost the signal to a level that’s easier to
measure. There’s a metal rod with fines in the chamber which stirs and reflects the
electromagnetic waves into numerous angles. The data recorded is the average and
maximum energy measured over all the stirring. The horn antenna receives energy
radiated from the device and cables connect that signal back to the network analyzer
which measures the received energy.
Figure 1: Reverberation Chamber Measurement Setup
In order to correlate measured data to simulated results, the different components
of the measurement path like the cables, the chamber, etc. must be calibrated. The cable
loss is measured by simply bypassing the amplifier and the chamber, thus connecting the
cables in a loop. The amplifier gain is generated by measuring the cable path with the
amplifier and subtracting the cable loss. The chamber calibration is show in Figure 2.
The horn-to-horn measurement is done in the chamber, and the cable loss is subtracted to
get the chamber calibration factor. The last piece of the puzzle is the device loss. For
instance, in a connector test board, there are a few centimeters of trace before the
connector, which adds a little bit of loss. The cable, amplifier, chamber calibration and
test board loss are added together to get the total calibration factor.
Figure 2: Reverberation Chamber Calibration Setup
Calibrating the test setup is critical to correlating measurement results to
simulation results. For the cable and amplifier, this is straightforward by connecting
cables in the right path and yields predictable results. The chamber calibration requires
careful setup of the antenna horns and calculations with the cable loss.
(a) (b)
Figure 3 (a) and (b) show the reverberation chamber calibration setup.
(a) (b)
Figure 3 (a) Cable feed-through into EMI reverberation chamber & (b) cable feed
through to horn antenna
The total calibration factor is added to the raw measurement, in order to correlate
simulation results to measurements. The simulated result is equal to a known source,
right at the device, with all the far-field radiation integrated at the radiation boundary of
the simulation. Thus the measurement + total calibration factor = simulated radiation
power.
III. 2: Results of Measurement to Correlation Effort
The calibration results are shown in Figure 4 includes cable loss, amplifier gain,
chamber calibration and total calibration curve. The gain curve is lower at low
frequencies because the amplifier is at its max output for most of the frequency range.
The amplifier nominal gain is 23dB, and has a maximum output power of about 12dBm.
Note that the gain curve and the cable loss mostly cancel each other, so the chamber
calibration is similar to the total calibration.
Figure 4: Measurement Calibration Data
During the measurements, there is access to two reverberation chambers. They
are identical in geometry and similar in antenna horns. The chamber calibration factor is
similar between the two reverberation chambers. There are some noticeable differences
below 6GHz. This result substantiates the concept that the chamber effects are linear,
passive, and repeatable across multiple chambers of the same geometry.
Combining the total calibration factor with a measurement, allows the direct
comparison of simulation results to measurement results. The correlation, between
simulation and measurement, is done with a surface mount connector, on a test board,
with edgecard mating. The measurement equipment is set to a 1 mW, single-ended
source. The simulation is also set to a 1 mW, single-ended source. Figure 6 shows the
simulation and measurement of radiated power, and shows good correlation overall. The
trend across the frequency range correlates well. The magnitude matches well at
frequencies greater than 10GHz. The measurement shows peaks much more sharply than
the simulation.
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Figure 5: Dual Chamber Calibration Factors
Figure 6: Total Radiated Power, Simulation vs. Measurement
Since most radiated energy problems are at 10GHz and higher, the measurement-to-
simulation correlation is sufficient to compare connectors and geometries in simulation
and see what features tend to decrease radiated power.
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Simulated Measured
Section IV
Results of Simulation Experiments
Simulations are completed for various configurations. Multiple connectors are
analyzed, and each connector is analyzed with a set of multiple sources. The connectors
analyzed are a right angle surface mount connector, a right-angle press-fit connector; a
mezzanine connector and a vertical surface mount connector. This is summarized in
Table 2 and Figure 7.
Table 2 Connector types simulated to generate total radiated power
Connector Type Ground structure
Right-angle surface mount Ground pins same as signal pins
Right-angle press-fit Ground pins wider than signal pins
Mezzanine press-fit Large ground pins, like a shield
Vertical surface mount Large ground pins, like a shield
Figure 7: Connector types simulated to generate total radiated power
The energy sources are in a differential configuration, common mode
configuration, single-ended, ground noise and “super common mode”. The ground noise
configuration is a voltage source driving the entire ground structure of the model. The
super common mode configuration is a voltage source that drives all metal in the model,
including the ground structure and the signals. In all the source configurations, the 3D
field solver model uses open waveports, which means that all metal, including ground is
driven as a terminal and solved as a signal. In the edit source setup, the grounds are set to
0V for the single-ended, differential and common mode configurations. For the ground
noise configuration, the signals are set to 0V and the ground structure is driven with a
voltage. In the super common mode configuration, both signal pins and the ground
structure are driven in phase with 1/3 of 1 mW each. The voltages are set so that the
source is 1 mW. The ground structure termination is set to 0.1 ohms to ground to
represent a low impedance path to ground. The signal pins are terminated with 50 ohms
to ground. The 1 mW normalized source configurations are noted in Table 3.
Table 3: Source configurations normalized to 1 mW
Source
Configuration
Signal 1 (V) Signal 1
phase
Signal 2 (V) Signal 2
phase
Ground (V)
Single-ended 0.316228 0 0 0 0
Differential 0.223607 0 0.223607 180 0
Common Mode 0.223607 0 0.223607 0 0
Ground noise 0 0 0 0 0.014142
Super Common
mode
0.181659 0 0.181659 0 0.008124
The purpose, of the multiple-source configurations, is to understand radiated
power as a function of sources in a system. A typical concern for EMI is noise on a
ground structure. Another typical EMI concern is common mode energy on a differential
path. Figure 8 shows total radiated power (TRP) results for the prototype right-angle
surface mount connector 7. The results in Figure 8 clearly show that given a constant
source power sinewave, common mode energy radiates more at frequencies below 8
GHz. Above 8 GHz, differential energy and common mode energy are similar. The
radiated power from a differential signal has a resonance peak at 10 GHz, while the
common mode radiated power is lower at that frequency. Results clearly show that
ground noise doesn’t radiate very efficiently per mW, compared to the signal structure.
With a 14 mV amplitude sinewave, on the ground structure at 10 GHz, the structure
radiates power at almost 2 orders of magnitude lower than a 0.316 V sinewave driven on
a signal pin.
In order to compare the radiated power of these various structures at a constant
voltage, a source of 0.1V amplitude is used in all configurations. The results are plotted
in Figure 9. Given the same source power, the differential mode radiated power is the
highest of all the configurations at 10 GHz. The common mode radiated power is highest
from 0 to 9 GHz. Above 11 GHz, the common mode and differential mode radiated
power is about the same.
Figure 8: Total Radiated Power vs. Frequency, All sources normalized to 1 mW,
prototype right-angle surface mount connector
Figure 9: Total Radiated Power vs. Frequency, All sources normalized to 0.1V,
prototype right-angle surface mount connector
When the configurations are driven with the same voltage, the relative merit of
the configurations changes significantly compared to constant source power. By voltage,
the ground structure radiates the most power. Due to low impedance, the ground
structure radiates much less than differential and common mode. The super common
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Frequency (GHz)Single-ended 1mW Differential 1mWCommon Mode 1mW Ground noise 1mWSuper common mode 1mW
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Single-ended 0.1V Differential 0.1V Common Mode 0.1V
Ground noise 0.1V Super comm 0.1V
mode configuration is just a super-position of the ground noise and common mode
configurations, so ground noise and common mode will be the focus of subsequent
analysis.
Let’s examine the radiated power results compared to s-parameter values. S-
parameter files for electronic structures are generally more available than radiated power
data. In Figure 10, it can be seen that taking 1 minus all energy coming out of the ports,
yields a curve that has the same shape as the radiated power of a single-ended source.
This makes sense because the s-parameter file contains single port data. Also, it makes
sense because all energy into the structure must go somewhere. Most of it goes out of the
ports, while the remainder (1 – all ports) is either radiated or dissipated. In the antenna
parameters, this remainder is the accepted power. The dissipated power is the dielectric
and conductor loss. The accepted power reduced by the dissipated power generates the
radiated power curve. Therefore, the radiated power from a single-ended source is 1-
{energy from all ports} shifted lower by the loss in the structure. Both the differential
source and radiation results and the common mode source and radiation results roughly
correlate by shape. The loss dips correlate to radiated energy peaks, but magnitudes do
not correlate well. S-parameters are good to estimate the trends of radiated power vs.
frequency, but are not good enough to estimate absolute magnitudes or whether a
component will pass or fail EMI tests.
Figure 10: Prototype Right-angle SMT Connector: Radiated Power vs. S-parameter
In a typical electronic system, the sources will not be 1 mW for all configurations,
nor will they be 0.1V on each metal pin. In addition, the sources will not be a single
frequency sinewave. For the purposes of a system analysis, the sinewaves are weighted
by a typical frequency spectrum for binary data. Differential, common mode and ground
noise uses a 25Gbps binary waveform spectrum. The single-ended source uses a binary
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Frequency (GHz)SDD21 Differential 1mWCommon Mode 1mW 1 - allportsSCC21 Single-ended 1mW, rt angle SMTSingle-ended dissipation
waveform spectrum at a data rate of 8Gbps. Differential signals are driven with
waveforms typically from 0.3V to 1V. The common mode signals are designed to be
small, for instance the IEEE 802.3ba specifies AC common mode output to 30mV max.
Ground noise and super common mode are designed to be very small in a system,
hopefully 10mV or less. Single-ended pins may be driven to 0.3V to 1.0V, similar to the
differential signals. However, they are typically lower data rates and slower rise time
than differential signals. Assume 0.6V amplitude, with 6dB loss on the differential path,
the result is 0.3V on the differential signal. Assume a 0.5V sinewave on the single-ended
path with a frequency range of 1 to 8GHz. Assume 30 mV common mode, plus -15dB
mode conversion of the 0.6V differential signal, which adds another 100mV for a
maximum common mode total of 0.13 V. While lab measurements occasionally show
large ground noise spikes of 100 mV peak amplitude or more, the high frequency
sinewave component is typically much smaller. Let’s use 25mV as the ground noise
sinewave amplitude. Figure 11 adjusts the source voltages to these estimates of typical
system values.
Figure 11: Analysis of radiated power from a prototype right-angle SMT connector in a
system, weighted by the frequency spectrum of binary data waveforms
For this estimate of signals in a system, it’s clear that large amplitude single-ended
sources generate the most radiated power at lower frequencies, up to about 4GHz,
assuming a fast, 8Gbps single-ended source. Differential signals dominate above 8GHz.
In no frequency range, does the 25mV ground noise generate the most radiated power.
However, it must be noted that all plots shown thus far are in an unshielded environment.
Most systems have a grounded shield or cage, around some components, to reduce
radiated power. The next experiment is to simulate a shield around a connector and
examine what happens to radiated power when voltage is driven onto that shield. The
signal pins inside the cage certainly radiate more per mW than the ground structure, but
they are shielded by the cage. In Figure 12, a shielded differential signal is analyzed and
compared with a noise source on the shield structure.
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Single-ended 0.5V, 8Gbps Differential 0.3V, 25Gbps
Common Mode 0.13V, 25Gbps Ground noise 0.025V, 25Gbps
Figure 12: SHIELDED SYSTEM: Analysis of radiated power in a system with shielding
around the connector, weighted by the frequency spectrum of binary data
It’s clear that for an unshielded case, a 25mV source on the ground structure is
much less significant than the driven signals, be they common mode, differential or
single-ended. However, with a shielded structure, all the driven signals are sufficiently
shielded that they have radiated energy that’s insignificant compared to a noisy shield.
The right-angle surface mount connector analyzed has a noticeable differential
resonance around 10GHz. This connector is a prototype and the final connector design
has significant resonance reduction, compared to the prototype. The final right-angle
SMT design with resonance improvement is analyzed and compared to the prototype
design (Figure 13). Notice that there’s a big improvement at 10GHz, since the resonance
is reduced.
Figure 14 compares the TRP for the 4 different connectors when each is driven
with a purely differential signal. At low frequency, there’s up to 20dB difference in the
radiation from one connector to another. At frequencies of 10GHz and above, the
differences are less pronounced.
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Single-ended 0.5V, 8Gbps Differential 0.3V, 25Gbps
Common Mode 0.13V, 25Gbps Ground noise 0.025V, 25Gbps
Figure 13: Total Radiated Power vs. frequency, estimated system sources, for prototype
vs. final design of right-angle SMT connector
Figure 14: Total Radiated power vs. Frequency among 4 connector types, with
differential 25Gbps sources
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Differential 0.3V, 25Gbps final Common Mode 0.13V, 25Gbps final
Differential 0.3V, 25Gbps proto Common Mode 0.13V, 25Gbps proto
final design SDD21 prototype SDD21
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Mezz, Diff 0.3V, 25Gbps Right-angle press-fit, Diff, 0.3V, 25Gbps
Vert SMT, Diff 0.3V, 25Gbps right-angle SMT, Diff 0.3V, 25Gbps
Figure 15 shows the TRP results for different connectors when driven with a
common mode signal. At low frequency, there’s up to 15dB difference in the radiation
from one connector to another. At frequencies of 10GHz and above, the differences are
less pronounced.
Figure 15: Total Radiated power vs. Frequency among 4 connector types, with common
mode 25Gbps sources
Using the selection of connector results, these results are compared to EMI spec
to estimate what radiated energy performance is good. With the datasets of total radiated
power for multiple connectors, in various configurations, the question remains; what will
pass EMI tests? This is a question that’s best answered by measurement to the spec, or
extensive analysis. Some estimates can be made to get a rough idea of EMI. There is an
FCC spec for EMI for enterprise (business/commercial) environments, which has a limit
of 54dBuV/m for electrical field, at 3 meters from a component, above 1GHz. Using
total radiated power, over the surface of a sphere with 3 meter radius, the average electric
field is calculated. In order to meet the 54dBuV/m spec, the estimate is that the total
radiated power (TRP) must be less than or equal to -24dBm. In Figure 16, the estimated
TRP target is plotted compared to the four analyzed connectors, in the unshielded case. It
is clear that at 25Gbps no unshielded connector meets the target.
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Mezz, Comm 0.13V, 25Gbps Right-angle pfit, Comm, 0.13V, 25Gbps
Vert SMT, Comm 0.13V, 25Gbps Right-angle SMT, Comm 0.13V, 25Gbps
Figure 16: Total Radiated Power, Estimated target to meet 54dBuV/m FCC spec vs.
Connector type @ 25Gbps
In the case with 12.5Gbps, the radiated energy (Fig. 17) is much closer to meeting
the -24dBm target. In fact, the Mezzanine connector meets the TRP target with both
common mode and differential mode energy sources.
Multiple drivers are also investigated. The cases analyzed include differential and
common mode sources, driven on all differential pairs in the model. The right-angle
press-fit connector is analyzed in this configuration, with all 8 pairs (16 signal pins)
driven with a source. The results are shown in Figure 18.
These results show that multi-pair sources add up to higher radiated power than a
single pair, as expected. More power in equals more power out. However, it seems that
the differential radiated power adds up more linearly than the common mode power. One
hypothesis is that the common mode sources generate some patterns that cancel out
radiated energy. Another hypothesis is that the pairs in the right angle pairs have various
lengths, each of which excites a different resonance. These differential resonances may
add up to create smooth, wide peaks in the radiated energy. The multi-pair source
configurations warrant further study.
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Mezz, Diff 0.3V, 25Gbps Right-angle press-fit, Diff, 0.3V, 25Gbps
Vert SMT, Diff 0.3V, 25Gbps Right-angle SMT, Diff 0.3V, 25Gbps
Mezz, Comm 0.13V, 25Gbps Right-angle pfit, Comm, 0.13V, 25Gbps
Vert SMT, Comm 0.13V, 25Gbps Right-angle SMT, Comm 0.13V, 25Gbps
TRP target for 54dBuV/m average E-field
Figure 17: Total Radiated Power, Estimated target to meet 54dBuV/m FCC spec vs.
Connector type, 12.5Gbps
Figure 18: Total Radiated Power vs. Frequency for Multi-pair sources vs. single-pair source
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Mezz, Diff 0.3V, 12.5Gbps Right-angle press-fit, Diff, 0.3V, 12.5Gbps
Vert SMT, Diff 0.3V, 12.5Gbps Right-angle SMT, Diff 0.3V, 12.5Gbps
Mezz, Comm 0.13V, 12.5Gbps Right-angle pfit, Comm, 0.13V, 12.5Gbps
Vert SMT, Comm 0.13V, 12.5Gbps Right-angle SMT, Comm 0.13V, 25Gbps
TRP target for 54dBuV/m average E-field
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Differential 8 pairs, 0.3V 25Gbps Differential 0.3V, 25Gbps
Common Mode 0.13V, 25Gbps Common Mode 8 pairs, 0.3V 25Gbps
Single-ended connector paths are analyzed surrounded by ground contacts on both
sides, compared to ground contacts on one side. (Figure 49) The data shows clearly that
the signal with grounds on both sides radiates less energy than with ground on one side.
This suggests that connects that more ground pins, even small structures, help confine the
radiated fields.
Figure 49: Snapshot and plot of radiated energy vs. ground pin configuration
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Single-ended 0.5V, 8Gbps grounds on both sides
Single-ended 0.5V, 8Gbps grounds on one side
SECTION V
Conclusion
Simulations are shown to produce accurate radiated energy results. Simulation
results correlate well with measurements if the both the simulation and the measurement
are setup accurately. In this investigation, the key factors for correlation are:
1. Match the simulation source to the measurement source
2. Generate sufficient mesh on the radiation boundary of the simulation
3. Calibrate the measurement carefully, including cable, horn, chamber and board
effects. The chamber loss is found to be the trickiest part of the calibration.
Simulation results across various connectors and multiple source configurations are
completed. These results show some differences in connector radiation at frequencies
below 10GHz, and at strong resonance frequencies. Above 10GHz, all connectors look
similar. At resonance frequencies, the connectors show higher a radiated energy peak,
corresponding to the signal source. For instance, a connector structure with a common
mode resonance shows a radiated energy peak when driven by a common mode signal,
but not when driven by a differential signal. At low frequencies, the connector with a
large, simple ground structure has the lowest radiated energy, while the connector with a
large, multi-pin, ground structure has the higher radiated energy. With sources at
25Gbps, the data suggest that all connectors analyzed probably need a shield to meet
FCC EMI specs. At 12.5Gbps, one connector has radiated energy that meets the
estimated TRP target. With many sources at multi-Gbps data rates, the simulation data
suggests it is not likely that an unshielded, open air connector will meet FCC EMI specs.
Using readily available S-parameters, one can generate an estimate of radiated power
by calculated all the energy that doesn’t exit the ports of the model. Then, by subtracting
a typical smooth curve of conductive and dissipative loss, the radiated power can be
estimated. This radiated power estimate is reasonable for sources that match the s-
parameter source, which in most cases is single-ended. It is not accurate to apply this
method to generate estimates of differential or common mode radiated energy from a
single-ended s-parameter file. Further study is needed to evaluate whether a single-ended
s-parameter file can be converted to a differential/common s-parameter file and then used
to generate reasonable estimates of differential and common mode radiated energy.
Results show that single-ended sources, at relatively high voltage amplitudes and at
data rate of a few Gbps, can be the dominant radiated energy source up to about 5GHz.
Differential and common mode sources can have similar radiated energy or significantly
different radiated energy, depending on the connector geometry. Simulations confirm
that tens of millivolts, on the outer shield structure, are likely to be the dominant radiated
energy sources in a system.
References
[1] Zhang Li ; Pingfang Yu ; Chua Chee-Parng, “The EMI characteristics of High
Speed backplane connector,”
[2] Xiaoning Ye, James L. Drewniak, Jim Nadolny and David M. Hakanson, “High-
Performance Inter-PCB Connectors: Analysis of EMI Characteristics,” IEEE
Trans. Electrogmagnetic Compatibility, vol. 44, pp. 165–174, 2002.
[3] Xiaoning Ye, Jim Nadolny, James L. Drewniak, Richard E. DuBroff, Thomas P.
VanDoren and Todd H. Hubing, “Experimental and FDTD study of the EMI
performance of an open-pin-field connector for modules-on-backplanes”, IEEE
Trans. Electromagnetic Compatibility, vol. 2, pp. 789-794, 2000.
[4] Xiaoxia Zhao, Angela Li, Hongmei Fan, Alpesh Bhobe, Kam Taunk, Jinghan Yu,
Philippe Sochoux, et al, “Validating EMC Simulation by Measurement in
Reverberation Chamber”, Proc. DesignCon, 2013
[5] Alpesh Bhobe, Mike Fogg, Steve Dunwoody, Rich Long, “Application of Full-
wave 3D Field Solvers to Predict EMI Behavior in SFP Cages”, Proc.
DesignCon, 2010