rf and microwave basics

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RF/MW Component and System Measurements in Practice

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On day-1, we’ll start with the basics. This will help us to refresh our fundamentals beforewe proceed to advanced measurement topics.

We’ll talk about various terms and definitions used in day-to-day work. It is veryimportant to clearly understand their meanings.

Next we’ll revise the transmission line theory. We’ll compare various types oftransmission lines available for use.

We’ll also present the signal flow graph theory, understand the usefulness of complex-looking Smith Chart, and discuss why do need S-parameters.


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we’ll talk about definitions and parameters specific to RF and microwave componentsand systems. We’ll discuss in detail the important terms like Gain Compression, NoiseFigure and Phase Noise.

Next, we’ll shift our focus to components which form the system building blocks. We’lltalk about both passive and active devices and understand their characteristics andusefulness in the entire system configuration.


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Reviewing the receiver system performance and block diagram. We’ll highlight variousassociated terms like signal-to-noise ratio, inter-modulation distortion, dynamic rangeetc.

Having discussed the basics, we’ll move on to RF/Microwave test instruments. Thesignal sources will first be discussed. Here we’ll explain the simple block diagram andvarious related specifications like spectral purity.


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As an example, Teflon has a dielectric constant equal to 2.0 and is non-magnetic. Therefore the propagation speed of an electromagnetic wave in Teflon is

v = c / sqrt(2 X 1) = 0.7 X c = 2.1 X 108 m/s.


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A significant advantage associated with the use of microwaves for communications is their large bandwidth.Presently, microwave communications are widely used for telephone networks, in television broadcast andseveral other communications applications by defence service, railways etc. Short wavelengths also simplifythe design and installation of high directivity antenna. Antenna directivity and gain depends on the ratio ofphysical antenna aperture to the wavelength of the signal transmitted.Small antenna size and the property of reflection of microwaves from metallic surfaces make it practical tooperate radar systems at these frequencies. There is a whole variety of radars : early-warning radar, missile-tracking radar, missile-guidance radar, fix-control radar, weather-detection radar, air-traffic control radar andeven radars to detect and control the speed of automobiles.Unlike lower radio frequencies, these waves are not reflected and practically not absorbed by the ionosphere.This property makes microwaves suitable for Radio Astronomy and Satellite communications.Microwaves exhibit another interesting feature Molecular, atomic, and nuclear systems display variousresonance phenomena when placed in microwaves. The resonance absorption and the absorption spectraprovide information on the molecular structure of the materials. Besides scientific research, absorption ofmicrowaves by molecular resonances is well suited for various industrial measurements.Just like any other form of energy, microwave energy can also be used for heating. Thermal effects producedby microwaves have a variety of industrial applications. Microwave ovens for cooking follow the principle ofdielectric heating.Microwave diathermy machines produce heat inside the muscles without heating the tissues and skin outside.Also, microwave drying machines are used in printing, textile, and paper industries.


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For dealing with small wavelengths, methods of circuit representation and analysis needto be modified. The phase difference caused by the interconnection between variouscomponents or different parts of a single component is not negligible. Therefore, analysesbased on Kirchhoff's laws and voltage-current concepts are not adequate to describe thecircuit behaviour at microwave frequencies.It is necessary to analyse the circuit or networks in terms of electric or magnetic fieldsassociated with it.Measurements at microwave frequencies are carried out in terms of field amplitudes,phase differences and powers carried by the waves.Several special measurement techniques have been developed for use at microwavefrequencies such as based on standing wave pattern.


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TRANSMISSION LINES

In this lesson you will be introduced (or re-introduced) to the concept and some basic theory of transmission lines. The mathematics will be kept to a minimum although we make no apology for the occasional formula. Your instructor will describe the terms as they arise.The concept of transmission lines is basic to the understanding of RF and microwave technology, although the concept is by no means limited to those high frequencies but is also applied at power line frequencies (50 or 60Hz.) for understanding the transmission of electrical power over long distances.


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Transmission Lines come in many forms, here are some well known types.


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An equivalent lumped-circuit analogy of a transmission line cab be developed. A shortlength of parallel-wire transmission line and its equivalent circuit are shown.

The circuit contains a series inductance with resistance and a shunt capacitance withconductance. It can be imagined that a transmission line consists of an infinite number ofinfinitely short lengths of this type of two-port network, cascaded.The differential voltage drop (dV) across this infinitesimally short section of line (dx) isthe following :

dV = (R . dx + jwL . dx) I

The current through this section of line is the following.dI = (G . dx + jwC . dx) V

Solving these equations gives the relationship of V to I on the transmission line. The ratioof V to I is known as the transmission line characteristic impedance (Zo).


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In high-quality transmission lines at high frequencies, jwL >> R and jwC >> G. Underthese conditions R and G are negligible.

Zo then reduces to this equation :Zo = sqrt (L/C) ohms

Note the following from this approximation :

-The characteristic impedance is a real number (not complex or imaginary)-Current and voltage are in phase for the wave traveling down the infinite line.-The transmission line is lossless (L and C only).


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The need for efficient transfer of RF power is one of the main reasons behind the use oftransmission lines. At low frequencies where the wavelength of the signals are muchlarger than the length of the circuit conductors, a simple wire is very useful for carryingpower. Current travels down the wire easily, and voltage and current are the same nomatter where we measure along the wire.At high frequencies however, the wavelength of signals of interest are comparable to ormuch smaller than the length of conductors. In this case, power transmission can best bethought of in terms of traveling waves. When the transmission line is terminated in itscharacteristic impedance Z0 (which is generally a pure resistance such as 50 or 75 W),maximum power is transferred to the load. When the termination is not Z0, the portion ofthe signal which is not absorbed by the load is reflected back toward the source. Thiscreates a condition where the voltage along the transmission line varies with position.We will examine the incident and reflected waves on a transmission line with differentload conditions in the next three slides.


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Since a transmission line terminated in its characteristic impedance results in maximumtransfer of power to the load, there is no reflected signal. This result is the same as if thetransmission line was infinitely long. If we were to look at the envelope of the RF signalversus distance along the transmission line, it would be constant (no standing-wavepattern). This is because there is energy flowing in one direction only.


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Next, let's terminate our line in a short circuit. Since purely reactive elements cannotdissipate any power, and there is nowhere else for the energy to go, a reflected wave islaunched back down the line toward the source. For Ohm's law to be satisfied (no voltageacross the short), this reflected wave must be equal in voltage magnitude to the incidentwave, and be 180o out of phase with it. This satisfies the condition that the total voltagemust equal zero at the plane of the short circuit. Our reflected and incident voltage (andcurrent) waves will be identical in magnitude but traveling in the opposite direction.Now let us leave our line open. This time, Ohm's law tells us that the open can supportno current. Therefore, our reflected current wave must be 180o out of phase with respectto the incident wave (the voltage wave will be in phase with the incident wave). Thisguarantees that current at the open will be zero. Again, our reflected and incident current(and voltage) waves will be identical in magnitude, but traveling in the oppositedirection. For both the short and open cases, a standing-wave pattern will be set up on thetransmission line. The valleys will be at zero and the peaks at twice the incident voltagelevel. The peaks and valleys of the short and open will be shifted in position along theline with respect to each other, in order to satisfy Ohm's law as described above.


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Finally, let's terminate our line with a 25 W resistor (an impedance between the fullreflection of an open or short circuit and the perfect termination of a 50 W load). Some(but not all) of our incident energy will be absorbed in the load, and some will bereflected back towards the source. We will find that our reflected voltage wave will havean amplitude 1/3 that of the incident wave, and that the two waves will be 180o out ofphase at the load. The phase relationship between the incident and reflected waves willchange as a function of distance along the transmission line from the load. The valleys ofthe standing-wave pattern will no longer go zero, and the peak will be less than that ofthe short/open case.


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The first term for reflected waves is reflection coefficient gamma (G). The magnitude portion ofgamma is called rho (r). Reflection coefficient is the ratio of the reflected signal voltage to theincident signal voltage. For example, a transmission line terminated in Zo will have all energytransferred to the load; hence Vrefl = 0 and r = 0. When ZL is not equal to Zo , some energy isreflected and r is greater than zero. When ZL = a short or open circuit, all energy is reflected and r= 1. The range of possible values for r is then zero to one.Since it is often very convenient to show reflection on a logarithmic display, the second way toconvey reflection is return loss. Return loss is expressed in terms of dB, and is a scalar quantity. Thedefinition for return loss includes a negative sign so that the return loss value is always a positivenumber (when measuring reflection on a network analyzer with a log magnitude format, ignoringthe minus sign gives the results in terms of return loss). Return loss can be thought of as the numberof dB that the reflected signal is below the incident signal. Return loss varies between infinity for aZo impedance and 0 dB for an open or short circuit.As we have already seen, two waves traveling in opposite directions on the same media cause a"standing wave". This condition can be measured in terms of the voltage standing wave ratio(VSWR or SWR for short), and is defined as the maximum value of the RF envelope over theminimum value of the envelope. This value can be computed as (1+r)/(1-r). VSWR can take onvalues between one and infinity.


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Standing wave ratio (SWR) is the ratio of the maximum voltage on the line to theminimum voltage on the line and can be used to find p, the scalar reflection coefficient.For this reason, SER is a popular term to describe mismatch.


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Slide 10RF transmission lines can be made in a variety of transmission media. Common examples are coaxial, waveguide, twisted pair, coplanar, stripline and microstrip. RF circuit design on printed-circuit boards (PCB) often use coplanar or microstrip transmission lines. The fundamental parameter of a transmission line is its characteristic impedance Zo. Zo describes the relationship between the voltage and current traveling waves, and is a function of the various dimensions of the transmission line and the dielectric constant (er) of the non-conducting material in the transmission line. For most RF systems, Zo is either 50 or 75 ohms.

For low-power situations (cable TV, for example) coaxial transmission lines are optimized for low loss, which works out to about 75 ohms (for coaxial transmission lines with air dielectric). For RF and microwave communication and radar applications, where high power is often encountered, coaxial transmission lines are designed to have a characteristic impedance of 50 ohms, a compromise between maximum power handling (occurring at 30 ohms) and minimum loss.


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In order to completely characterize an unknown linear two-port device, we must make measurements under various conditions and compute a set of parameters. These parameters can be used to completely describe the electrical behavior of our device (or network), even under source and load conditions other than when we made our measurements. For low-frequency characterization of devices, the three most commonly measured parameters are the H, Y and Z-parameters. All of these parameters require measuring the total voltage or current as a function of frequency at the input or output nodes (ports) of the device. Furthermore, we have to apply either open or short circuits as part of the measurement. Extending measurements of these parameters to high frequencies is not very practical.


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At high frequencies, it is very hard to measure total voltage and current at the device ports. One cannot simply connect a voltmeter or current probe and get accurate measurements due to the impedance of the probes themselves and the difficulty of placing the probes at the desired positions. In addition, active devices may oscillate or self-destruct with the connection of shorts and opens.

Clearly, some other way of characterizing high-frequency networks is needed that doesn't have these drawbacks. That is why scattering or S-parameters were developed. S-parameters have many advantages over the previously mentioned H, Y or Z-parameters. They relate to familiar measurements such as gain, loss, and reflection coefficient. They are defined in terms of voltage traveling waves, which are relatively easy to measure. S-parameters don't require connection of undesirable loads to the device under test. The measured S-parameters of multiple devices can be cascaded to predict overall system performance. If desired, H, Y, or Z-parameters can be derived from S-parameters. And very important for RF design, S-parameters are easily imported and used for circuit simulations in electronic-design automation (EDA) tools like Agilent's Advanced Design System (ADS). S-parameters are the shared language between simulation and measurement.

An N-port device has N2 S-parameters. So, a two-port device has four S-parameters. The numbering convention for S-parameters is that the first number following the "S" is the port where the signal emerges, and the

second number is the port where the signal is applied. So, S21 is a measure of the signal coming out port 2 relative to the RF stimulus entering port 1. When the numbers are the same (e.g., S11), it indicates a reflection measurement, as the input and output ports are the same. The incident terms (a1, a2) and output terms (b1, b2) represent voltage traveling waves.


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S11 and S21 are determined by measuring the magnitude and phase of the incident, reflected and transmitted voltage signals when the output is terminated in a perfect Zo (a load that equals the characteristic impedance of the test system). This condition guarantees that a2 is zero, since there is no reflection from an ideal load. S11 is equivalent to the input complex reflection coefficient or impedance of the DUT, and S21 is the forward complex transmission coefficient. Likewise, by placing the source at port 2 and terminating port 1 in a perfect load (making a1 zero), S22 and S12 measurements can be made. S22 is equivalent to the output complex reflection coefficient or output impedance of the DUT, and S12 is the reverse complex transmission coefficient.

The accuracy of S-parameter measurements depends greatly on how good a termination we apply to the load port (the port not being stimulated). Anything other than a perfect load will result in a1 or a2 not being zero (which violates the definition for S-parameters). When the DUT is connected to the test ports of a network analyzer and we don't account for imperfect test-port match, we have not done a very good job satisfying the condition of a perfect termination. For this reason, two-port error correction, which corrects for source and load match, is very important for accurate S-parameter measurements (two-port correction is covered in the calibration section).


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'Microwaves' is a descriptive term used to identify electromagnetic waves in the frequencyspectrum ranging approximately from 1GHz to 30GHz. This corresponds to wavelengthsfrom 30 cm to 1 cm.


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If the test signal were a pure sine wave, nonlinearities in a device would introduceharmonics in the device's output. The magnitude of the harmonics could be a measure ofthe amount of nonlinearity in the device..


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A popular measurement of non-linearity in amplifier is gain compression. Shownis a plot of an amplifier's output power versus input power at a single frequency.Gain at any power level is the slope of this curve. Notice that the amplifier has alinear region of operation where gain is constant and is independent of powerlevel. This is commonly called "small signal gain". As the input power isincreased to a level that causes the amplifier to saturate, gain non-linearlyincreases, causing the "large signal" response, showing the limitation in theamplifier's output power. For this session, gain compression is defined asmeasuring the output power when the gain has decreased by 1 dB, although ingeneral, an x dB compression can be found. This "1 dB compression point" is acommon measure of an amplifier's output capability.


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Another source of distortion in amplifiers is that caused by intermodulation products. When two or more input signals are applied to a non-linear amplifier, the output contains additional frequency components called intermodulation products. The output signal will contain frequency components at the two fundamental input frequencies f1, f2; harmonics at 2f1, 2f2, 3f1, 3f2; second order product at f1+ f2 and third order products at 2f1- f2, 2f2- f1. As shown the third order products are very close to the fundamentals and typically fall within the amplifiers bandwidth, producing distortion in the output.

Measurement of the third order intermodulation products is very important for in-band channel measurements, since the distortion components can fall within theactual communication channel of interest and typically cannot be filtered.


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Shown is a typical intermodulation distortion measurement using two signalgenerators and a spectrum analyzer. The signal generators outputs areconnected together through a coupler or combiner. The isolation of this devicemust be sufficient enough to isolate the RF signal generators from each other,reducing mismatch interactions of the two RF input signals. Another possibility toreduce the interaction of the RF input signals is to turn off the ALC functions ofboth signal generators. This reduces many amplitude variations caused by theintermodulation effects of the RF sources.

Shown is an intermodulation measurement made at a center frequency of 20MHz with the two RF inputs at 19.9 MHz and 20.1 MHz. The third order productsat 19.7 MHz and 20.3 MHz are approximately -40 dBc. The spectrum analyzermust have a resolution bandwidth narrow enough to resolve the individual signalsin this type of test.


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The third-order intercept, or TOI, is a common figure of merit for theintermodulation distortion produced by the amplifier under test. TOI derives itsname from the relationship that, on a logarithmic scale, the amplitude of the third-order products increases three times as fast as the amplitude of the fundamentalsignals. For example, if the fundamental signals increased 10 dB the third-orderproducts would increase 30 dB.

In theory, as the amplitude of the fundamental signals increases higher andhigher, there would exist a point where the amplitudes of the fundamental signalsand the third order products would be equal. This extrapolated point ofintersection is called the third-order intercept. This TOI point is never actuallyattained due to amplifier's gain compression effects, but it can be calculated fromlower level amplitude measurements according to, TOI = S + D/2, where Srepresents the amplitude of the fundamental signals in dBm, and D is the numberof dB that the third-order products are below the fundamentals.


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In this example, a perfect amplifier would add no noise, and the signal would be an amplified replica. However, in practice, this noise is broadband in nature, and can mask the wanted signal. The noise floor, as seen in a given bandwidth, limits the detection of weak signals.

Noise Limits the Processing of Weak Signals. A receiver designer recognizes the boundary conditions that limit the usable dynamic range and S/N. The noise floor limits the detection of weak signals, and the distortion products limit the upper level of detectable power.


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Noise follows the normal power transfer laws. A conjugate match is needed for an optimum noise power transfer from the source to the load. This is termed the available noise power. Nyquist arrived at the equation Pav. = KTB watts. 290K was formally adopted as the Standard Temperature for determining noise figure. This gives us the figure of -174dBm / Hz as the universal noise floor at that temperature. So, example of this relationship :Doubling the bandwidth will add 3dB to the available noise power. Also doubling the absolute temperature increases the Noise power by 3dB


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We are now going to talk about signal to noise ratios, and how wedefine Noise Figure.

In this example , we have an amplifier G=20dB. The left graphshows the I/p Signal and noise versus frequency, while the rightshows the Signal and noise after the 20dB gain. If the device isnoiseless, then the o/p noise floor will go up by its gain. IE from -100dBm to -80dBm. However, the amplifier is noisy. It adds noise,such that the noise floor is at -70dBm, and additional 10dB.

The signal to Noise Ratio has changed by 10dB. This degradationis the Noise Figure of the device.


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In this example, we have a satellite :effective radiated power of +55dBm,path loss, of +200dB, antenna with gain of 60dB.

The signal power to the receiver is -85dBm. The receiver sensitivity :kTB is at -174dBm /Hz the noise power in a 100 MHz bandwidth is10 log ( 10^8) of that ina 1Hz BW.the noise figure of the device is +5dB.

receiver noise floor is at -89dBm. If we wanted to double the link margin, then we could double the transmitter power. This would cost billions fo dollars in terms of increased payload and /or higher rated, more expensive components and more challanging engineering issues. Another way is to increase the gain of the receiver. This would cost millions in terms of size and mechanical engineering, and the debates over local environmental issues and planning permissions. While lowering the Noise Figure of the front end would be a fraction of this, and is the more attractive economically.


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Before we get to the formal definitions of phase noise, let's look at the difference between an ideal signal (a perfect oscillator). In the frequency domain, this signal is represented by a single spectral line.

In the real world however, there is always small, unwanted amplitude and phase fluctuations present on the signal. Notice that frequency fluctuations are actually an added term to the phase angle portion of the time domain equation. Because phase and frequency are related, you can discuss equivalently about unwanted frequency or phase fluctuations.

In the frequency domain, the signal is no longer a discrete spectral line. It is now represented by spread of spectral lines - both above and below the nominal signal frequency in the form of modulation sidebands due to random amplitude and phase fluctuations.

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In the ideal case, all of the power in a sine wave is concentrated at a single frequency. Random noise within the source will cause the power to be spread over a small range of frequencies. The spread is referred to as phase noise and is often mathematically modeled as random phase modulation. The units of phase noise are dBc/Hz: dB down from the carrier in a 1 Hz bandwidth. Phase noise is specified at a frequency offset from the CW output. For example, the phase noise of a CW source may be specified as: -97dBc/Hz @ 100 kHz offset from a CW frequency at 20 GHz.

Phase noise may be directly measured from the spectrum of a source. This method requires that the phase noise of the analyzer be much better (~10dB) than the phase noise of the source being tested. Often, the phase noise of a source is measured using test equipment that has been optimized for this purpose. Phase noise is generally displayed on a log-log axis. This enables both the close in phase noise (offsets < 1 kHz) and the far out phase noise (offsets > 10 kHz) to be easily examined on one plot.

The phase noise plot above was generated using the Agilent E4440A by using the phase noise measurement personality. The source is at 1 GHz.

Residual FM is a measure of the small amount of FM inherent in an CW output. Residual FM is specified within a bandwidth. Most sources typically specify residual FM per the CCITT specified bandwidth. The CCITT bandwidth starts at 300 Hz offset from the carrier frequency and stops at a 3 kHz offset. Within this band, all of the noise shown on the phase noise curve contributes to residual FM.


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Attenuation can be achieved in two ways. One, by using graphitized sand. When a microwave signal encounters the sand the current generated turns the signal energy into heat. This method is most useful for terminators. The other method is by placing a resistive rod or vane at the centre of the electric field. The electric field induces a current flow resulting in an ohmic power loss. The vane method is useful in variable attenuators.


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When a lumped element resonator is excited by an active device, the resultant signal voltage or current will have a resonant frequency fr = 1/2o sqrt(LC)


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Every oscillator must have a resonator to set the frequency of the oscillator and an activedevice to produce the instability. Once an unstable condition is produced by the activedevice, then the signal current and voltage will have a frequency which is set by theresonant frequency of the resonator.


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The range of impedance for reasonable mechanical ratios, for example if D/d varies from 1.5 to 10 the impedance for an air dielectric varies from 24.3 ohm to 138 ohms. So any mechanically easily made coax line is going to have a characteristic impedance around 25 to 100 ohms. For 50 ohms the ratio is 2.3.This has not answered the question; why 50ohms? It seems to be a compromise.The lowest attenuation occurs when the inner to outer diameter ratio is 3.6 which corresponds to an impedance of 77.5 ohms. On the other hand, because the electric field strength is highest near the surface of the inner conductor this field strength could cause breakdown if too high. By calculating between the voltage breakdown in air and the transmitted power in a reflectionless line gives a diameter ratio of 1.65, corresponding to 30 ohms.


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The 1.0 mm series adapters are designed to be used for the measurement of components with 50 ohm 1.0 mm connectors with a frequency range from dc to 110 GHz.

The 1.0 mm connector utilizes an air dielectric interface for the highest accuracy and repeatability. The coupling diameter and thread size maximize strenght, increase durability, and provide highly repeatable connections. The connectors are designed so that the outer conductors engage before the center conductors.


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Objective: To get basic understanding of how spec an's work, how to use them to their fullest potential, and how to make more effective for particular applications.

Agenda: Overview - to define spectrum analysis and introduce types of tests made with a spectrum analyzer

Theory of Operation - to learn about the hardware inside an analyzer, and how the components all work together.

Specifications - understanding the specificationss of the analyzer will help determine if a particular instrument will make the required measurements and the accuracy of the results.

Features - hardware and firmware features make the analyzer more effective for particular applications. We will discuss some of the more important and widely used features.

Summary -


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Welcome to our presentation - Network Analysis Fundamentals by Agilent Technologies.


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Here are some examples of the types of devices that you can test with network analyzers. They include both passive and active devices (and some that have attributes of both). Many of these devices need to be characterized for both linear and nonlinear behavior. It is not possible to completely characterize all of these devices with just one piece of test equipment.

The next slide shows a model covering the wide range of measurements necessary for complete linear and nonlinear characterization of devices. This model requires a variety of stimulus and response tools. It takes a large range of test equipment to accomplish all of the measurements shown on this chart. Some instruments are optimized for one test only (like bit-error rate), while others, like network analyzers, are much more general-purpose in nature. Network analyzers can measure both linear and nonlinear behavior of devices, although the measurement techniques are different (frequency versus power sweeps for example). This module focuses on swept-frequency and swept-power measurements made with network analyzers


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Here is a key to many of the abbreviations used above:Response84000 84000 series high-volume RFIC testerDed. Testers Dedicated (usually one-box) testersVSA Vector signal analyzerSA Spectrum analyzerVNA Vector network analyzerTG/SA Tracking generator/spectrum analyzerSNA Scalar network analyzerNF Mtr. Noise-figure meterImped. An. Impedance analyzer (LCR meter)Power Mtr. Power meterDet./Scope Diode detector/oscilloscope

MeasurementACP Adjacent channel powerAM-PM AM to PM conversionBER Bit-error rateCompr'n Gain compressionConstell. Constellation diagramEVM Error-vector magnitudeEye Eye diagramGD Group delayHarm. Dist. Harmonic distortionNF Noise figureRegrowth Spectral regrowthRtn Ls Return lossVSWR Voltage standing wave ratio


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Components are tested for a variety of reasons. Many components are used as "building blocks" in more complicated RF systems. For example, in most transceivers there are amplifiers to boost LO power to mixers, and filters to remove signal harmonics. Often, R&D engineers need to measure these components to verify their simulation models and their actual hardware prototypes. For component production, a manufacturer must measure the performance of their products so they can provide accurate specifications. This is essential so prospective customers will know how a particular component will behave in their application.

When used in communications systems to pass signals, designers want to ensure the component or circuit is not causing excessive signal distortion. This can be in the form of linear distortion where flat magnitude and linear phase shift versus frequency is not maintained over the bandwidth of interest, or in the form of nonlinear effects like intermodulation distortion.

Often it is most important to measure how reflective a component is, to ensure that it absorbs energy efficiently. Measuring antenna match is a good example.


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In many situations, magnitude-only data is sufficient for out needs. For example, we may only care about the gain of an amplifier or the stop-band rejection of a filter. However, as we will explore throughout this paper, measuring phase is a critical element of network analysis.

Complete characterization of devices and networks involves measurement of phase as well as magnitude. This is necessary for developing circuit models for simulation and to design matching circuits based on conjugate-matching techniques. Time-domain characterization requires magnitude and phase information to perform the inverse-Fourier transform. Finally, for best measurement accuracy, phase data is required to perform vector error correction.


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Now that we fully understand the relationship of electromagnetic waves, we must also recognize the terms used to describe them. Common network analyzer terminology has the incident wave measured with the R (for reference) receiver. The reflected wave is measured with the A receiver and the transmitted wave is measured with the B receiver. With amplitude and phase information of these three waves, we can quantify the reflection and transmission characteristics of our device under test (DUT). Some of the common measured terms are scalar in nature (the phase part is ignored or not measured), while others are vector (both magnitude and phase are measured). For example, return loss is a scalar measurement of reflection, while impedance results from a vector reflection measurement. Some, like group delay, are purely phase-related measurements. Ratioed reflection is often shown as A/R and ratioed transmission is often shown as B/R, relating to the measurement receivers used in the network analyzer


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Transmission coefficient T is defined as the transmitted voltage divided by the incident voltage. If |Vtrans| > |Vinc|, the DUT has gain, and if |Vtrans| < |Vinc|, the DUT exhibits attenuation or insertion loss. When insertion loss is expressed in dB, a negative sign is added in the definition so that the loss value is expressed as a positive number. The phase portion of the transmission coefficient is called insertion phase.

There is more to transmission than simple gain or loss. In communications systems, signals are time varying -- they occupy a given bandwidth and are made up of multiple frequency components. It is important then to know to what extent the DUT alters the makeup of the signal, thereby causing signal distortion. While we often think of distortion as only the result of nonlinear networks, we will see shortly that linear networks can also cause signal distortion.


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Before we explore linear signal distortion, lets review the differences between linear and nonlinear behavior. Devices that behave linearly only impose magnitude and phase changes on input signals. Any sinusoid appearing at the input will also appear at the output at the same frequency. No new signals are created. When a single sinusoid is passed through a linear network, we don't consider amplitude and phase changes as distortion. However, when a complex, time-varying signal is passed through a linear network, the amplitude and phase shifts can dramatically distort the time-domain waveform.

Non-linear devices can shift input signals in frequency (a mixer for example) and/or create new signals in the form of harmonics or intermodulation products. Many components that behave linearly under most signal conditions can exhibit nonlinear behavior if driven with a large enough input signal. This is true for both passive devices like filters and even connectors, and active devices like amplifiers


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Here is an example of a square wave (consisting of three sinusoids) applied to a bandpass filter. The filter imposes a non-uniform amplitude change to each frequency component. Even though no phase changes are introduced, the frequency components no longer sum to a square wave at the output. The square wave is now severely distorted, having become more sinusoidal in nature.


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Let's apply the same square wave to another filter. Here, the third harmonic undergoes a 180o phase shift, but the other components are not phase shifted. All the amplitudes of the three spectral components remain the same (filters which only affect the phase of signals are called allpass filters). The output is again distorted, appearing very impulsive this time.


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Another useful measure of phase distortion is group delay. Group delay is a measure of the transit time of a signal through the device under test, versus frequency. Group delay is calculated by differentiating the insertion-phase response of the DUT versus frequency. Another way to say this is that group delay is a measure of the slope of the transmission phase response. The linear portion of the phase response is converted to a constant value (representing the average signal-transit time) and deviations from linear phase are transformed into deviations from constant group delay. The variations in group delay cause signal distortion, just as deviations from linear phase cause distortion. Group delay is just another way to look at linear phase distortion.

When specifying or measuring group delay, it is important to quantify the aperture in which the measurement is made. The aperture is defined as the frequency delta used in the differentiation process (the denominator in the group-delay formula). As we widen the aperture, trace noise is reduced but less group-delay resolution is available (we are essentially averaging the phase response over a wider window). As we make the aperture more narrow, trace noise increases but we have more measurement resolution.


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Now that we have seen some of the measurements that are commonly done with network and spectrum analyzers, it might be helpful to review the main differences between these instruments. Although they often both contain tuned receivers operating over similar frequency ranges, they are optimized for very different measurement applications.

Network analyzers are used to measure components, devices, circuits, and sub-assemblies. They contain both a source and multiple receivers, and generally display ratioed amplitude and phase information (frequency or power sweeps). A network analyzer is always looking at a known signal (in terms of frequency), since it is a stimulus-response system. With network analyzers, it is harder to get an (accurate) trace on the display, but very easy to interpret the results. With vector-error correction, network analyzers provide much higher measurement accuracy than spectrum analyzers.

Spectrum analyzers are most often used to measure signal characteristics such as carrier level, sidebands, harmonics, phase noise, etc., on unknownsignals. They are most commonly configured as a single-channel receiver, without a source. Because of the flexibility needed to analyze signals, spectrum analyzers generally have a much wider range of IF bandwidths available than most network analyzers. Spectrum analyzers are often used with external sources for nonlinear stimulus/response testing. When combined with a tracking generator, spectrum analyzers can be used for

scalar component testing (magnitude versus frequency, but no phase measurements). With spectrum analyzers, it is easy to get a trace on the display, but interpreting the results can be much more difficult than with a network analyzer.


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Here is a generalized block diagram of a network analyzer, showing the major signal-processing sections. In order to measure the incident, reflected and transmitted signal, four sections are required:

• Source for stimulus• Signal-separation devices• Receivers that downconvert and detect the signals• Processor/display for calculating and reviewing the resultsWe will briefly examine each of these sections. More detailed information about the signal separation devices and receiver section are in the appendix.


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The signal source supplies the stimulus for our stimulus-response test system. We can either sweep the frequency of the source or sweep its power level. Traditionally, network analyzers used a separate source. These sources were either based on open-loop voltage-controlled oscillators (VCOs) which were cheaper, or more expensive synthesized sweepers which provided higher performance, especially for measuring narrowband devices. Excessive phase noise on open-loop VCOs degrades measurement accuracy considerably when measuring narrowband components over small frequency spans. Most network analyzers that Agilent sells today have integrated, synthesized sources, providing excellent frequency resolution and stability.


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The next major area we will cover is the signal separation block. The hardware used for this function is generally called the "test set". The test set can be a separate box or integrated within the network analyzer. There are two functions that our signal-separation hardware must provide. The first is to measure a portion of the incident signal to provide a reference for ratioing. This can be done with splitters or directional couplers. Splitters are usually resistive. They are non-directional devices (more on directionality later) and can be very broadband. The trade-off is that they usually have 6 dB or more of loss in each arm. Directional couplers have very low insertion loss (through the main arm) and good isolation and directivity. They are generally used in microwave network analyzers, but their inherent high-pass response makes them unusable below 40 MHz or so.

The second function of the signal-splitting hardware is to separate the incident (forward) and reflected (reverse) traveling waves at the input of our DUT. Again, couplers are ideal in that they are directional, have low loss, and high reverse isolation. However, due to the difficulty of making truly broadband couplers, bridges are often used instead. Bridges work down to DC, but have more loss, resulting in less signal power delivered to the DUT. See the appendix for a more complete description of how a directional bridge works.


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The next portion of the network analyzer we'll look at is the signal-detection block. There are two basic ways of providing signal detection in network analyzers. Diode detectors convert the RF signal level to a proportional DC level. If the stimulus signal is amplitude modulated, the diode strips the RF carrier from the modulation (this is called AC detection). Diode detection is inherently scalar, as phase information of the RF carrier is lost.

The tuned receiver uses a local oscillator (LO) to mix the RF down to a lower "intermediate" frequency (IF). The LO is either locked to the RF or the IF signal so that the receivers in the network analyzer are always tuned to the RF signal present at the input. The IF signal is bandpass filtered, which narrows the receiver bandwidth and greatly improves sensitivity and dynamic range. Modern analyzers use an analog-to-digital converter (ADC) and digital-signal processing (DSP) to extract magnitude and phase information from the IF signal. The tuned-receiver approach is used in vector network analyzers and spectrum analyzers.


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The last major block of hardware in the network analyzer is the display/processor section. This is where the reflection and transmission data is formatted in ways that make it easy to interpret the measurement results. Most network analyzers have similar features such as linear and logarithmic sweeps, linear and log formats, polar plots, Smith charts, etc. Other common features are trace markers, limit lines, and pass/fail testing. Many of Agilent's network analyzers have specialized measurement features tailored to a particular market or application. One example is the E5100A/B, which has features specific to crystal-resonator manufacturers.


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Let's look at the three basic sources of measurement error: systematic, random and drift.

Systematic errors are due to imperfections in the analyzer and test setup. They are repeatable (and therefore predictable), and are assumed to be time invariant. Systematic errors are characterized during the calibration process and mathematically removed during measurements.Random errors are unpredictable since they vary with time in a random fashion. Therefore, they cannot be removed by calibration. The main contributors to random error are instrument noise (source phase noise, sampler noise, IF noise).

Drift errors are due to the instrument or test-system performance changing after a calibration has been done. Drift is primarily caused by temperature variation and it can be removed by further calibration(s). The timeframe over which a calibration remains accurate is dependent on the rate of drift that the test system undergoes in the user's test environment. Providing a stable ambient temperature usually goes a long way towards minimizing drift.


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Shown here are the major systematic errors associated with network measurements. The errors relating to signal leakage are directivity and crosstalk. Errors related to signal reflections are source and load match. The final class of errors are related to frequency response of the receivers, and are called reflection and transmission tracking. The full two-port error model includes all six of these terms for the forward direction and the same six (with different data) in the reverse direction, for a total of twelve error terms. This is why we often refer to two-port calibration as twelve-term error correction


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The two main types of error correction that can be done are response (normalization) corrections and vector corrections. Response calibration is simple to perform, but only corrects for a few of the twelve possible systematic error terms (the tracking terms). Response calibration is essentially a normalized measurement where a reference trace is stored in memory, and subsequent measurement data is divided by this memory trace. A more advanced form of response calibration is open/short averaging for reflection measurements using broadband diode detectors. In this case, two traces are averaged together to derive the reference trace.

Vector-error correction requires an analyzer that can measure both magnitude and phase. It also requires measurements of more calibration standards. Vector-error correction can account for all the major sources of systematic error and can give very accurate measurements.

Note that a response calibration can be performed on a vector network analyzer, in which case we store a complex (vector) reference trace in memory, so that we can display normalized magnitude or phase data. This is not the same as vector-error correction however (and not as accurate), because we are not measuring and removing the individual systematic errors, all of which are complex or vector quantities.


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Vector-error correction is the process of characterizing systematic error terms by measuring known calibration standards, and then removing the effects of these errors from subsequent measurements.

One-port calibration is used for reflection measurements and can measure and remove three systematic error terms (directivity, source match, and reflection tracking). Full two-port calibration can be used for both reflection and transmission measurements, and all twelve systematic error terms are measured and removed. Two-port calibration usually requires twelve measurements on four known standards (short-open-load-through or SOLT). Some standards are measured multiple times (e.g., the through standard is usually measured four times). The standards themselves are defined in a cal-kit definition file, which is stored in the network analyzer. Agilent network analyzers contain all of the cal-kit definitions for our standard calibration kits. In order to make accurate measurements, the cal-kit definition MUST MATCH THE ACTUAL CALIBRATION KIT USED! If user-built calibration standards are used (during fixtured measurements for example), then the user must characterize the calibration standards and enter the information into a user cal-kit file. Sources of more information about this topic can be found in the appendix.


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A network analyzer can be used for uncorrected measurements, or with any one of a number of calibration choices, including response calibrations and one- or two-port vector calibrations. A summary of these calibrations is shown above. We will explore the measurement uncertainties associated with the various calibration types in this section.


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Let's look at some actual measurements done on a bandpass filter with different levels of error correction. The uncorrected trace shows considerable loss and ripple. In fact, the passband response varies about 1 dB around the filter's center frequency. Is the filter really this bad? No. What we are actually measuring is the sum of the filter's response and that of our test system.

Performing a normalization prior to the measurement of the filter removes the frequency response of the system (transmission tracking error) from the measurement. The loss that was removed was most likely caused by the test cables. After normalization, the frequency response of the filter still contains ripple caused by an interaction between the system's source and load match. This ripple even goes above the 0 dB reference line, indicating gain! However, we know that a passive device cannot amplify signals. This apparent anomaly is due to mismatch error.

The measurement shown after a two-port calibration is the most accurate of the three measurements shown. Using vector-error correction, the filter's passband response shows variation of about 0.1 dB around its center frequency. This increased level of measurement flatness will ensure minimum amplitude distortion, increase confidence in the filter's design, and ultimately increase manufacturing yields due to lower test-failure rates.


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Whenever possible, reflection calibrations should be done with a cal kit that matches the connector type of the DUT. If adapters need to be used to mate the calibrated test system to the DUT, the effect of these adapters on measurement accuracy can be very large. This error is often ignored, which may or may not be acceptable. As the slide shows, the adapter causes an error signal which can add or subtract with the desired reflection signal from the DUT. Worst-case effective directivity (in dB) is now:

-20 log (Corrected-coupler-directivity + radapters)

If the adapter has a SWR of say 1.5 (the less-expensive variety), the effective directivity of the coupler drops to around 14 dB worst case, even if the coupler itself had infinite directivity! In other words, with a perfect Zo load (rL= 0) on the output of the adapter; the reflected signal appearing at the coupled port would only be 14 dB less than the reflection from a short or open circuit. Stacking adapters compounds the problem, as is illustrated above. Consequently, it is very important to use quality adapters (or preferably, no adapters at all) in your measurement system, so system directivity is not excessively degraded. While error-correction can mitigate the effect of adapters on the test port, our system is more susceptible to drift with degraded raw (uncorrected) directivity.


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When performing a two-port calibration, we have some choices based on the type of calibration standards we want to use. So far, we have only discussed coaxial calibration techniques. Let's briefly look at TRL (through-reflect-line), a calibration technique that is especially useful for microwave, noncoaxial environments such as fixture, wafer probing, or waveguide. It is the second-most common type of two-port calibration, after SOLT. TRL solves for the same 12 error terms as the more common SOLT calibration, but uses a slightly different error model.The main advantage of TRL is that the calibration standards are relatively easy to make and define at microwave frequencies. This is a big benefit since is difficult to build good, noncoaxial, open and load standards at microwave frequencies. TRL uses a transmission line of known length and impedance as one standard. The only restriction is that the line needs to be significantly longer in electrical length than the through line, which typically is of zero length. TRL also requires a high-reflection standard (usually, a short or open) whose impedance does not have to be well characterized, but it must be electrically the same for both test ports.

For RF applications, the lengths of the transmission lines needed to cover down to low frequencies become impractical (too long). It is also difficult to make good TRL standards on printed-circuit boards, due to dielectric, line-dimension, and board-thickness variations. And, the typical TRL fixture tends to be more complicated and expensive, due do the need to accommodate

throughs of two different physical lengths.

There are two variations of TRL. True TRL calibration requires a 4-receiver network analyzer. The version for three-receiver analyzers is called TRL* ("TRL-star"). Other variations of this type of calibration (that share a common error model) are Line-Reflect-Line (LRL), Line-Reflect-Match (LRM), Thru-Reflect-Match (TRM), plus many others.


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Time-domain reflectometry (TDR) is a very useful tool that allows us to measure impedance versus distance. One good application for TDR is fixture design and the design of corresponding in-fixture calibration standards. We can distinguish between capacitive and inductive transitions, and see non-Zo transmission lines. TDR can help us determine the magnitude and position of reflections from transitions within the fixture, and we can measure the quality of the calibration standards. As long as we have enough spatial resolution, we can see the reflections of the connector launches independently from the reflections of the calibration standards. It is very easy to determine which transition is which, as the designer can place a probe on a transition and look for a large spike on the TDR trace.

With time-domain gating, we can isolate various sections of the fixture and see the effects in the frequency domain. For example, we can choose to look at just the connector launches (without interference from the reflections of the calibration standards), or just the calibration standards by themselves.

Another application for TDR is fault-location for coaxial cables in cellular and CATV installations. We can use TDR in these cases to precisely determine the location of cable faults such as crimps, poor connections, shorts, opens -- anything that causes a portion of the incident signal to be reflected.


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Time-domain transmission (TDT) is a similar tool which uses the transmission response instead of the reflection response. It is useful in analyzing signal timing in devices such as SAW filters. Gating is also useful in TDT applications. In the above example, a designer could look at the frequency response of the main surface wave without the effect of the leakage and triple-travel error signals.


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Let's briefly review how balanced devices work. Ideally, a balanced device only responds to or generates differential-mode signals, which are defined as two signals that are 180o out of phase with one another. These devices do not respond to or generate in-phase signals, which are called common-mode signals. In the top example of a balanced-to-single-ended amplifier, we see that the amplifier is responding the differential input, but there is no output when common-mode or in-phase signals are present at the input of the amplifier. The lower example shows a fully balanced amplifier, which is both differential inputs and outputs. Again, the amplifier only responds to the differential input signals, and does not produce an output in response to the common-mode input.One of the main reasons that balanced circuits are desirable is because external signals that are radiated from an RF emitter show up at the terminals of the device as common mode, and are therefore rejected by the device. These interfering signals may be from other RF circuitry or from the harmonics of digital clocks or data. Balanced circuits also reject noise on the electrical ground, since the noise appears in phase to both input terminals, making it a common-mode signal.


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We just discussed ideal balanced circuits. Real world devices on the other hand, don’t completely reject common mode noise or generate only differential signals. The example on the top shows that a balanced device will produce a small amount of common mode signal that rides on top of the differential signal output. This common mode signal is the result of differential to common mode conversion, and it is a source of electromagnetic interference. The bottom example shows what can happen when a common-mode signal is present at the input to the device. Common to differential mode conversion results in a differential signal at the output of the device, which will interfere with the desired differential output, which is not shown on the slide for simplicity. This mode conversion makes a circuit susceptible to electromagnetic interference.


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Now that you see how these balanced circuits operate, you can begin to understand why characterizing this behavior is very important to RF and digital designers. They need to characterize the differential to differential mode, which represents the desired mode of operation, as well as the undesired mode conversions. Another challenge is that differential devices and circuits often have input and output impedances other than 50 ohms. Characterizing these devices with a standard two-port network analyzer is difficult. And finally, there are other differential parameters that are also not measured with standard two-port network analyzers, such as common-mode rejection ratio or conjugate matching.


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Agilent’s solutions for balanced measurements address the challenges that R&D and digital designers face when characterizing differential devices. Data can be presented as mixed-mode S-parameters, which allows designers to clearly see the desired and undesired modes of operation. An optional time-domain package allows designers to view traditional TDR and TDT waveforms. Using a vector network analyzer-based solution offers excellent dynamic range and accuracy. The software provides many features needed for balanced measurements that are not provided in the firmware of traditional two-port network analyzers.


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Objective: To get basic understanding of how spec an's work, how to use them to their fullest potential, and how to make more effective for particular applications.

Agenda: Overview - to define spectrum analysis and introduce types of tests made with a spectrum analyzer

Theory of Operation - to learn about the hardware inside an analyzer, and how the components all work together.

Specifications - understanding the specificationss of the analyzer will help determine if a particular instrument will make the required measurements and the accuracy of the results.

Features - hardware and firmware features make the analyzer more effective for particular applications. We will discuss some of the more important and widely used features.

Summary -


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What is spectrum analysis? Very basically, it is the analysis of the signal by finding its energy or power as a function of frequency. The spectrum analyzer is the tool of spectrum analysis, it allows viewing and measurement of the way energy is distributed in frequency. This is called the frequency domain.By measuring signals in terms of their frequency domain components, insight into operation, performance specifications and troubleshooting are facilitated.

Understanding the important aspects of a spectrum analyzer will enable an operator make accurate measurements and to have confidence in the interpretation of the results.


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Some types of test:

There are several different tests or measurements that can be made with a spectrum analyzer. The most usual are to do with modulation, distortion, or noise.When measuring a modulated signal; modulation degree, sideband amplitude, modulation quality, and occupied bandwidth are the usual parameters.The measurement of distortion of a system or device are necessary to verify performance specifications include: intermodulation, harmonics, and spurious emissions.Not only is it important to understand the signal being transmitted, amplified, or filtered, but it is also very important to measure noise in the system ordevice in order to characterize its adverse effect on overall performance.

To comply with various national and international regulations, and for good engineering practice, the control of unwanted emissions is most easily understood with a spectrum analyzer. These are EMI, electromagnetic interference, measurements. Understanding the tests you need to make is critical for choosing the right measurement tool and getting the most out of it.

M6-118

Traditionally, when you want to look at an electrical signal, you use an oscilloscope to see how the signal varies with time. This is very important information, however, it doesn't give you the full picture. To fully understand the performance of your device/system, you will also want to analyze the signal(s) in the frequency-domain. This is a graphical representation of the signal's amplitude as a function of frequency The spectrum analyzer is to the frequency domain as the oscilloscope is to the time domain. (It is important to note that spectrum analyzers can also be used in the fixed-tune mode (zero span) to provide time-domain measurement capability much like that of an oscilloscope.)

The figure shows a signal in both the time and the frequency domains. In the time domain, all frequency components of the signal are summed together and displayed. In the frequency domain, complex signals (that is, signals composed of more than one frequency) are separated into their frequency components, and the level at each frequency is displayed.

Frequency domain measurements have several distinct advantages. For example, let's say you're looking at a signal on an oscilloscope that appears to be a pure sine wave. A pure sine wave has no harmonic distortion. If you look at the signal on a spectrum analyzer, you may find that your signal is actually made up of several frequencies. What was not discernible on theoscilloscope becomes very apparent on the spectrum analyzer.

Some systems are inherently frequency domain oriented. For example, many telecommunications systems use what is called Frequency Division Multiple Access (FDMA) or Frequency Division Multiplexing (FDM). In these systems, different users are assigned different frequencies for transmitting and receiving, such as with a cellular phone. Radio stations also use FDM, with each station in a given geographical area occupying a particular frequency band. These types of systems must be analyzed in the frequency domain in order to make sure that no one is interfering with users/radio stations on neighboring frequencies. We shall also see later howmeasuring with a frequency domain analyzer can greatly reduce the amount of noise present in the measurement because of its ability to narrow the measurement bandwidth.

From this view of the spectrum, measurements of frequency, power, harmonic content, modulation, spurs, and noise can easily be made. Given the capability to measure these quantities, we can determine total harmonic distortion, occupied bandwidth, signal stability, output power, intermodulation distortion, power bandwidth, carrier-to-noise ratio, and a host of other measurements, using just a spectrum analyzer.


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Some systems are inherently frequency domain oriented. For example, many telecommunications systems use what is called Frequency Division Multiple Access (FDMA) or Frequency Division Multiplexing (FDM). In these systems, different users are assigned different frequencies for transmitting and receiving, such as with a cellular phone. Radio stations also use FDM, with each station in a given geographical area occupying a particular frequency band. These types of systems must be analyzed in the frequency domain in order to make sure that no one is interfering with radio stations on neighboring frequencies.

From this view of the spectrum, measurements of frequency, power, harmonic content, modulation, spurs, and noise can easily be made. Given the capability to measure these quantities, we can determine total harmonic distortion, occupied bandwidth, signal stability, output power, intermodulation distortion, power bandwidth, carrier-to-noise ratio, and a host of other measurements, using just a spectrum analyzer


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The Fourier analyzer uses a time record of a signal, digitizes it using digital sampling, and then performs the mathematics required to convert it to the frequency domain, and display the resulting spectrum. It is as if the analyzer is looking at the entire frequency range at the same time using parallel filters.

With its real-time signal analysis capability, the Fourier analyzer is able to capture periodic as well as random and transient events. It also can provide significant speed improvement over the more traditional swept analyzer and can measure phase as well as magnitude. However it does have its limitations, particularly in the areas of frequency range, sensitivity, and dynamic range. We shall discuss what these terms are and why they are important in a later section.

As analog-to-digital converters (ADC) and digital signal processing (DSP) technologies advance, Fourier analyzers can operate at frequencies that are high enough to make them an important addition to the RF engineers toolbox. These analyzers can offer significant performance improvements over conventional spectrum analyzers, and will no doubt assume an increasingly important place as the analyzer of choice.


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The most common type of frequency-domain analyzer is the swept-tuned receiver.Very basically, these analyzers "sweep" across the frequency range of interest, displaying all the frequency components present. It works much like your AM radio, except that the dial controls the tuning on the radio, and the output is a speaker rather than a display.The present advantages of swept-tuned over Fourier analyzer are:

•wider frequency range,•larger dynamic range,•lower noise floor.

In most implementations of swept analyzers phase information is lost.

In the next part of this lesson, the term spectrum analyzer will refer only to the swept-tuned analyzer.


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Block Diagram.This is the basic block diagram for the swept-tuned spectrum analyzer. The analyzer will have a much more complicated block diagram than this but from the point of view of the functioning of the analyzer this diagram works well.Before considering how it all works together to create a display of frequency versus amplitude on the screen, the major functional components will be briefly discussed.Why is it important to understand the block diagram?To some extent, even for simple instruments, knowing how the instrument works can add confidence for the user and authority to the results. A spectrum analyzer is not a simple instrument and may be used for many applications and types of signals. Except for the most simple measurements with a spectrum analyzer some degree of expert knowledge is necessary for good measurement practice and interpretation of results. A knowledge of the block diagram will go a long way towards that expert knowledge.


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Input Attenuator:The RF input attenuator is used to adjust the level of the RF signal into the mixer. It is important to be able to do this in order to protect the mixer from large signals and to control the spurious free dynamic range of the spectrum analyzer. The RF attenuator may be set to zero dB, however this condition must be used with care, analyzer designers have used mechanical interlocks or special button sequences to avoid an operator accidentally selecting 0dB attenuation. Most analyzers have attenuators with the range 0dB to 70dB with ten or five dB step selection.

Input Filter:This limits the amount of RF energy to the band of the analyzer. In an RF analyzer this

band is the whole RF bandwidth of the instrument. In a microwave spectrum analyzer (for example > 3GHz) this filter is a tunable bandpass filter which tunes with the sweep, called a microwave preselector.


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The Mixer:There are many mixing stages in a spectrum analyzer, we can explain the function of the analyzer by treating it as a simple one stage receiver.

Mixers are three-port devices that convert a signal from one frequency to another, the mixer may be called, a frequency translation device.

The RF input signal is applied to the input port, and the Local Oscillator signal to the L port. The LO is a high level signal (> +10dBm) compared to the RF (< –10dBm).A mixer is a non-linear device, this means that besides the wanted frequency translated signal, there are a number (theoretically an infinite number) of signals we don’t want.The wanted signals from the mixer are (fR – fL) or (fL – fR), positive quantities.It is this difference frequency that is of interest in the spectrum analyzer, We call this signal the IF signal, or Intermediate Frequency signal.

Microwave Spectrum Analyzers.The IF is a relatively low frequency, so the LO must generate frequencies over the same nominal frequency range as the analyzer’s input frequency range. In microwave spectrum analyzers, therefore, to avoid the expense of providing a fundamental LO which would be a wide range microwave LO, multipliers may be used or more usually harmonic relationships at

the mixer output are used. (fR –n.fL) or (n.fL – fR)


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The Intermediate Frequency (IF) or Resolution Bandwidth (RBW) Filter:

The IF filter is also known as the Resolution bandwidth filter. When you change the RBW on the analyzer, you change the bandpass width of this filter.

The IF filter is a bandpass filter that is used as the "window" for detecting signals.Spectrum analyzers typically give several choices of RBW settings. By having a broad range of variable RBW settings, the analyzer can be optimized for a given measurement condition.

The example in this illustration shows that as the filter is narrowed, selectivity is improved and two closely spaced input signals may be resolved. This will, however, slow down the sweep speed.

The resolution bandwidth filter will be discussed in other parts of this class.


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The IF signal detector:

The detector converts the IF signal to a baseband or video signal so it can be viewed on the instrument's display. It is the envelope of the IF signal which is proportional to the input RF spectrum amplitude, so the detector is called an envelope detector.It is the output of this envelope detector which is needed to deflect the trace in the y-axis, or amplitude axis, of the display. (In older spectrum analyzers the waveform from the envelope detector would be amplified and applied to the y-axis of a CRT display.)

In spectrum analyzers using analog CRT displays, great efforts were made to maintain the trace on the screen especially during long sweeptimes, so called storage CRT’s were employed, which took much practice to adjust properly.


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The x-axis of the display can be thought of as being made up of "bins" (or trace elements) from which the analog detector output is digitally sampled. A natural question is: what point in the bin do we use for our data point?

In positive peak detection mode, the peak value of the signal over the duration of one trace element is used, whereas in negative peak detection mode, it’s the minimum value. Positive detection mode is typically used when analyzing sinusoids,(CW and stationary signals), but is not good when noise, or noise-like measurements are needed, because the value of the noise level is biased positively by the positive peak process.

In sample detection, since the sample is taken at evenly spaced time intervals, there is no bias in the sampled data, so the statistics of noise and noise like signals are preserved. This detection mode is used for measuring noise or noise-like signals.


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The video filter is a low-pass filter that is located after the detector and before the ADC.It is used to average or smooth the trace that is seen on the display, as shown on the slide. As the video filter BW is decreased the trace will become smoother . A change in video BW does not change the mean noise level. The small decrease in displayed noise level observed, when the VBW filter is reduced and the positive peak detector is in effect, is because the displayed noise is biased above the mean by the +ve peak detection process.The smoothing of the displayed noise allows signals close to the noise level to be observed. Remember that the displayed trace represents signal + noise, even when the noise is smoothed.


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Video or trace averaging is a feature of most modern analyzers, in this case averaging is performed over two or more sweeps on a point-by-point basis. At each display point the new value is calculated from the new measured value and the previously averaged data.

Aavg = new average valueAprior avg = average from prior sweepsAn = measured value on current sweepn = number of current sweep

navgprioravg A

nA

nnA .1.1


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The LO is a voltage controlled oscillator (VCO) which provides the LO signal for to the mixer. A sweep generator provides the ramp voltage to tune the LO in proportion to the ramp voltage. It also controls the horizontal position of the trace displayed on the spectrum analyzer display. This makes the display X axis the frequency axis. The ramp voltage may switched from the LO while still sweeping the display, in this mode the analyzer behaves like a fixed tuned reciever. This is called zero span or zero scan, the tuned frequency may be changed like changing stations on an AM radio.


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The IF gain is used to adjust the vertical position of signals on the display without affecting the signal level at the mixer. When we change this level, the reference level is changed accordingly.The RF attenuator value and IF amplifier gain, are coupled. When the RF input attenuator is changed, the IF gain will automatically change so that signals will remain stationary on the screen.


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The Displayed Average Noise Level, DANL, is also a function of RBW:

The internally generated noise in a spectrum analyzer is random and has no discrete spectral components. Also, its level is flat over a frequency range that is wide compared to the RBW ranges.This means that the total noise reaching the detector (and displayed) is related to the RBW selected. Since the noise is random, it is added on a power basis, so the relationship between displayed noise level and RBW is a ten log basis. In other words, if the RBW is increased (or decreased) by a factor of ten, ten times more (or less) noise energy hits the detector and the displayed average noise level (DANL) increases (or decreases) by 10 dB.


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This is a time domain view of the signal flow through the analyzer. Some aspects such as the way the display shows the IF filter shape will be more clear with this view. Notice how the envelope detector will have a repetative IF pulse when the signal is CW.


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What needs to be known about a spectrum analyzer in order to make sure that it will make particular measurements, and make them adequately? The most basic performance parameters are:

1) what is a safe power level?

2) what's the frequency range?

3) what's the amplitude range (maximum input and sensitivity)?

4) how different in amplitude can two signals be similtaneously present and be measured accurately?

5) how accurate are the measurements made with a spectrum ananlzer?

Although not in the same order, we will describe each of these areas in terms of what they mean, and why they are important.


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Before connecting the signal to a spectrum analyzer (or any instrument) be sure that there is no charge on the cable and be aware of input limitations. These are usually printed close the terminals.Static precautions are usually observed very strictly in production environments and should be taken seriously in less structured situations. Although the effect of static discharge may be obvious if it destroys the instrument input, often the effect is gradual, causing a progressive deterioration in performance.


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In summary, let's discuss all the "ranges" associated with a spectrum analyzer. Typically the term "dynamic range" only refers to the ability to measure two signals at the same time.

Display range refers to the calibrated amplitude range of the CRT display. For example, some analyzers with a display having 8 divisions might only have a 70 dB display range (with 10 dB/ div) because the bottom division is not calibrated, while others will have a full 100dB display and calibrated amplitude range. Measurement range is the ratio of the largest to the smallest signal that can be measured under any circumstances - not at the same time. The upper limit is determined by the maximum safe input level, +30 dBm (1 Watt) for most analyzers. Sensitivity sets the other end of the range.Dynamic Range:. . . it depends on what you are measuring.The other four ranges (signal/noise, signal/third order distortion, signal/second order distortion, and signal/noise sidebands) are when measuring two signals at the same time, and therefore are called dynamic range specifications. The ability of the analyzer to similtaneously large and small signals is limited by the dynamic range, When specifying dynamic range, it is important to qualify the term with the type of measurement to be made. The various specifications which can improve dynamic range will relate to the cost of

the spectrum analyzer.


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The features are categorized into application areas in order to better describe their function.

The first group, under Basic Operation, are some of the key features that enhance the use of the analyzer for any application. The others refer to a specific application, although the feature is not necessarily used only in that application.


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