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VC 2011 Wiley Periodicals, Inc.

SIMPLE METHOD TO DESIGN ATRI-BAND BANDPASS FILTER USINGASYMMETRIC SIRs FOR GSM, WiMAX,AND WLAN APPLICATIONS

Wei-Yu Chen,1 Yi-Hsin Su,2 Hon Kuan,2

and Shoou-Jinn Chang1

1 Institute of Microelectronics and Department of ElectricalEngineering, Advanced Optoelectronic Technology Center, Centerfor Micro/Nano Science and Technology, National Cheng KungUniversity, Taiwan; Corresponding author:mark00.chen@gmail.com2Department of Technology Electro-Optical Engineering, SouthernTaiwan University, Taiwan

Received 23 September 2010

ABSTRACT: This paper presents a simple method to design a tri-band

bandpass filter (BPF) using asymmetric stepped-impedance resonators(SIRs) with one-step discontinuity. Only two SIRs are needed in the filterstructure. The tri-band passband responses can be achieved by properly

selecting the length ratio (u) and the impedance ratio (R) of the asymmetricSIRs. The performances can be further well tuned by the lengths of twoenhanced coupling structure and gap of two asymmetric SIRs. A filter

example for the applications of GSM at 1.8 GHz, WiMAX at 3.5 GHz andWLAN at 5.2 GHz was designed and fabricated. The measured results are in

good agreement with the full-wave simulation results.VC 2011Wiley

Periodicals, Inc. Microwave Opt Technol Lett 53:1573–1576, 2011; View

this article online at wileyonlinelibrary.com. DOI 10.1002/mop.26037

Key words: tri-band; bandpass filter; stepped impedance resonator

1. INTRODUCTION

Recently, the development of multiservice wireless system and

mobile communication system are attractive in academic

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 53, No. 7, July 2011 1573

community and commercial electronics. Especially, the mobile

electronic products with the combination of global system for

mobile communications (GSM), worldwide interoperability for

microwave access (WiMAX), and wireless local area network

(WLAN) are popular and common nowadays. In such multiband

communication systems, bandpass filter (BPF) is used and plays

an important role in the RF front-end of the communication sys-

tem. To meet the multiband requirement, the multiband filters

were intensively proposed and investigated by using the open-

stub structures, the multimode resonators, such as stepped im-

pedance resonator (SIR), and the stub-loaded resonator [1–5]. In

the previous study [6], one of the authors adopted several reso-

nators and the multilayer technique to achieve tri-band

responses. However, the method suffers from the fabricated and

circuit complexities.

In this article, we propose a new and simple design method

for the tri-band BPF by using only two asymmetric SIRs, which

have only one-step discontinuity in the SIR structure, different

from the conventional SIRs. By properly selecting the length ra-

tio (u) and the impedance ratio (R) of the asymmetric SIRs, the

tri-band passband response can be achieved. The designed tri-

band BPF for GSM, WiMAX, and WLAN applications was fab-

ricated and measured. The measured results of the fabricated

sample are in good agreement with the results of full-wave elec-

tromagnetic (EM) simulation [7].

2. DESIGN OF TRI-BAND FILTER

In the first step, the resonant modes of the asymmetric SIR as

the main block of the proposed filter structure shall be dis-

cussed. Figure 1 shows the practical layout of the proposed

structure for the tri-band BPF by combining two asymmetric

SIRs and two enhanced coupling input/output ports. The con-

ventional SIR has two-step discontinuities, whereas the proposed

asymmetric SIR has only one-step discontinuity. The use of the

asymmetric SIR has the advantages of simple structure, easy

fabrication as well as the tunable resonance modes in the struc-

ture. Figure 2 shows the structure of the asymmetric SIRs,

which are constructed by connecting a high-impedance section

(Z1) with a low impedance (Z2). The impedance ratio R is

defined as R ¼ Z2/Z1. The input impedance Yin seen at one port

into the SIR is derived in the following equation [8]:

Yin ¼ jY2Rðcot h2 � tan h2Þ þ ðcot h1 � tan h1Þ

12ðcot h1 � tan h1Þðcot h2 � tan h2Þ � 2R

(1)

where y1 and y2 are the electrical lengths of the microstrip sec-

tion with characteristic impedance Z1 and Z2, respectively. It isknown that the resonance condition of the SIR is obtained as Yin¼ 0 in Eq. (1). However, to obtain more design freedom, the

length ratio (u) of asymmetric SIR can also be set as a variable

and adjusted to achieve the higher order resonant modes. The

length ratio (u) is defined as:

u ¼ h2h1 þ h2

¼ h2ht: (2)

Combine with (1) and (2), several resonant modes of the asym-

metric SIR dependent on R and u can be found. Figure 3 shows

the fundamental and higher order resonant modes versus length

ratio u as function of the impedance ratio R ¼ 0.25, 0.45, 0.5,

0.65, and 1.

In the second step, the obtained impedance ratio R and

length ratio u of the asymmetric SIR shall be chosen for achiev-

ing the required three resonant bands. For example, in this arti-

cle, a tri-band filter example will be designed for application of

GSM at 1.8 GHz, WiMAX at 3.5 GHz, and WLAN at 5.2 GHz,

basically based on the proposed structure. To meet this require-

ment, the impedance ratio R and length ratio u can be explicitly

determined as 0.65 and 0.85, respectively. Based on the obtained

impedance ratio R and length ratio u, the strip width and the

physical length of the SIRs can be obtained.

In the third step, the coupling gaps between the SIRs are

tuned by using the EM simulation to obtain the desired coupling

coefficients to achieve the passband performances of the

designed filter. Giving the designed targets: tri-band frequencies

at 1.8, 3.5, and 5.2 GHz, 3-dB fractional bandwidths (FBW) of

three passbands of 10, 6, and 6%, the lumped circuit element

values of the low-pass prototype filter are found to be g0 ¼ 1,

g1 ¼ 1.0177, g2 ¼ 1 [9]. Based on the coupling theory as

described in [10], the coupling coefficients are found to be Mf1

¼ 0.11, Mf2 ¼ 0.061, and Mf3 ¼ 0.06, where the f1, f2, and f3

Figure 1 Proposed circuit layout for the tri-band BPF

Figure 2 Geometrical diagram of the asymmetric stepped-impedance

resonator

Figure 3 Fundamental and higher order resonant modes versus length

ratio u as function of impedance ratio R ¼ 0.25, 0.45, 0.5, and 0.65

1574 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 53, No. 7, July 2011 DOI 10.1002/mop

are defined as the center frequencies of GSM, WiMAX, and

WLAN, respectively. To obtain the desired filter responses of

the tri-band, the coupling coefficients are used to determine the

suitable spacing of the asymmetric SIRs. The gap between two

asymmetric SIRs is finally determined as 0.2 mm to satisfy the

theoretical coupling coefficients.

Moreover, the responses of the tri-band BPF are also varied

with the gap of two asymmetric SIRs and the length of the

length of the enhanced coupled structure. As shown in Figure 4,

it is clearly observed that the appearance of transmission zeros

aside the second passband can be adjusted by the lengths of two

enhanced coupling structure. To obtain the optimum performan-

ces of the first and the third passband, the length of the

enhanced coupling structure is properly determined while L1 ¼15.5 mm and G1 is fixed at 0.5 mm. The performance of the

first passband is also adjustable as changing the gap, G1,

between each asymmetric SIRs. As shown in Figure 5, the first

passband is properly determined while G1 is set as 0.5 mm and

L1 is fixed at 15.5 mm.

3. MEASURE RESULTS

In the final step, the designed filter was fabricated on the sub-

strates of Duroid 5880 with dielectric constant er ¼ 2.2, loss tan-

gent d ¼ 0.0009 and thickness h ¼ 0.787 mm, after the BPF

was optimized by using a full-wave EM simulation [7]. Photo-

graph of the fabricated sample is shown in Figure 6(a). The

overall size of fabricated BPF is about 113.4 � 1.9 mm2, i.e.,

approximately 1.009kg � 0.016kg, where kg is the guided wave-

length on the substrate at the center frequency of the first pass-

band. The filter example was measured by an network analyzer

HP 8510C. The measured results show the return loss |S11| of 5dB, insertion loss |S21| of 1.2 dB, and FBW ¼ 11% at 1.8 GHz,

|S11| of 24 dB, |S21| of 1.8 dB and FBW ¼ 6.2% at 3.5 GHz and

|S11| of 14 dB, |S21| of 2 dB and FBW ¼ 6.1% at 5.2 GHz, as

shown in Figure 6(b). It is found that the center frequencies of

the three passbands are closely matched between the measured

results and the simulated results, thus verifying the proposed

design concept and theoretical prediction of the tri-passband.

Moreover, the excellent features indicate that the proposed

method for the tri-band BPF is useful to be utilized in the filter

design for the multiband applications.

4. CONCLUSIONS

In this article, a simple method to design a tri-band bandpass fil-

ter has been proposed. The proposed tri-band filter structure

used only two asymmetric SIRs. Design procedure in detail is

addressed in this article. First, these three passbands are

achieved by properly selecting the length ratio (u) and imped-

ance ratio (R) of the asymmetric SIRs, and then the performance

are further tuned by the lengths of two enhanced coupling struc-

ture and the gap of two asymmetric SIRs. The measured results

of the fabricated sample showed the return loss |S11| of 5 dB,

insertion loss |S21| of 1.2 dB; and FBW ¼ 11% at 1.8 GHz, |S11|of 24 dB, |S21| of 1.8 dB; and FBW ¼ 6.2% at 3.5 GHz and

|S11| of 14 dB, |S21| of 2 dB and FBW ¼ 6.1% at 5.2 GHz.

Figure 4 Simulated performances of the resonance frequencies by

changing the spacing L1 (W1 ¼ W2 ¼ W5 ¼ 0.2 mm, W3 ¼ 0.5 mm, W4

¼ 1.2 mm, L2 ¼ 54.2 mm, L3 ¼ 9.9 mm and G1 ¼ 5 mm)

Figure 5 Simulated performances of the resonance frequencies by

changing the spacing G1 (W1 ¼ W2 ¼ W5 ¼ 0.2 mm, W3 ¼ 0.5 mm, W4

¼ 1.2 mm, L1 ¼ 15.5 mm, L2 ¼ 54.2 mm and L3 ¼ 9.9 mm)

Figure 6 (a) Photograph (b) measured and simulated frequency

responses of the fabricated tri-band BPF, fabricated on the substrates of

Duroid 5880 with dielectric constant er ¼ 2.2, loss tangent ¼ 0.0009

and thickness h ¼ 0.787 mm. (W1 ¼ W2 ¼ W5 ¼ 0.2 mm, W3 ¼ 0.5

mm, W4 ¼ 1.2 mm, L1 ¼ 15.5 mm, L2 ¼ 54.2 mm, L3 ¼ 9.9 mm and

G1 ¼ 5 mm). [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 53, No. 7, July 2011 1575

Moreover, the measurement agrees well with the simulation,

verifying the proposed design concept.

REFERENCES

1. X. Guan, Z. Ma, P. Cai, Y. Kobayashi, T. Anada, and G. Hagi-

wara, Synthesis of dual-band bandpass filters using successive fre-

quency transformations and circuit conversions, IEEE Microwave

Wireless Compon Lett 16 (2006).

2. H.J. Lin, X.W. Shi, X.H. Wang, C.L. Li, and Q. Li, A novel CPW

dual passband filter using the split-modes of loaded stub square

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3. C.F. Chen, T.Y. Huang, and R.B. Wu, Design of dual- and tri pass-

band filters using alternately cascaded multiband resonators, IEEE

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4. M.H. Weng, H.W. Wu, and Y.K. Su, Compact and low loss dual-

band bandpass filter using pseudo-interdigital stepped impedance

resonators for WLANs, IEEE Microwave Wireless Compon Lett

17, 2007.

5. Y.C. Chang, C.H. Kao, M.H. Weng, and R.Y. Yang, A novel com-

pact dual-band bandpass filter using asymmetric SIRs for WLANs,

Microwave Opt Technol Lett 50, 2008.

6. M.H. Weng, H.W. Wu, K. Shu, R.Y. Yang, and Y.K. Su, A novel

tri-band bandpass filter using multilayer-based substrates for

WiMAX, European Microwave Conference (EuMC), Munich, Ger-

many, 2007.

7. IE3D Simulator, Zeland Software, Inc., Fremont, CA, 2002.

8. Y.C. Chang, C.H. Kao, M.H. Weng, and R.Y. Yang, Design of the

compact wideband bandpass filter with low loss, high electivity and

wide stopband, IEEE Microwave Wireless Compon Lett 18, 2008.

9. G.L. Matthaei, L. Young, and E.M.T. Jones, Microwave filters, im-

pedance-matching networks, and coupling structures. McGraw-Hill,

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10. J.S. Hong and M.J. Lancaster, Microstrip filter for RF/microwave

application. Wiley, New York, 2001.

VC 2011 Wiley Periodicals, Inc.

A SMALL-SIZE PENTABAND HAND-SHAPED COPLANAR WAVEGUIDE-FEDMONOPOLE ANTENNA

Alireza Mallahzadeh, Ali Foudazi,and Sajad Mohammad Ali NezhadShahed University, Persian Gulf Highway, Tehran, Islamic Republicof Iran; Corresponding author: alinezhad@shahed.ac.ir

Received 29 September 2010

ABSTRACT: A novel small-size simple structure pentaband coplanarwaveguide (CPW) fed monopole antenna is presented. The antenna

configuration is composed of several narrow strips that are connected toa 50-ohm CPW feed line. The length of each strip, which acting asresonance path, is kg/4 at operating frequencies. The antenna is

designed to operate over the worldwide interoperability for microwaveaccess and wireless local area networks frequency bands of 2.4, 3.3,

3.7, 5.2, and 5.8 GHz. It was found that the results of simulationand measurement agree well over each separated frequency bands.VC 2011 Wiley Periodicals, Inc. Microwave Opt Technol Lett 53:1576–

1579, 2011; View this article online at wileyonlinelibrary.com. DOI

10.1002/mop.26025

Key words: monopole antenna; CPW-fed; pentaband

1. INTRODUCTION

In recent decade, wireless local area networks (WLAN) and

worldwide interoperability for microwave access (WiMAX)

have evolved at very high rate for modern communication sys-

tems. Thus, design an antenna that can operate at WLAN and

WiMAX frequency bands have increasingly demanded.

Different studies were paid attention on planar antennas that

were designed to operate at WLAN and WiMAX frequency

bands. In comparison to nonprinted antennas, printed antennas

due to their small size, low profile, lightweight, and low cost are

more attractive [1]. Among different structure of printed anten-

nas, printed coplanar waveguide (CPW)-fed antenna is more

appropriate for WLAN and WiMAX applications, due to its

omnidirectional radiation coverage, little dependence of the

characteristic impedance on substrate height, single metallic

layer structure, easy integration to monolithic microwave inte-

grated circuits, and comparatively higher bandwidth [2, 3].

Recently, there has been an increase in the use of antenna

operating at several frequency bands for WLAN and WiMAX

systems. Several techniques to obtain multiband CPW antenna

were reported in literatures. One of the techniques is the design

of a wideband antenna and introducing notches to obtain the

multiband behavior [4]. In the second technique, by creating

multi-independent resonance paths, multiband CPW antenna is

obtained. A dual-band CPW-fed closed rectangular ring monop-

ole antenna with a vertical strip [5], a triple-band CPW-fed F-

shaped monopole antenna [6], and a dual-frequency double

inverted-L CPW-fed monopole antenna [7] were reported. In the

third technique, by inserting parasitic elements near or behind

the radiating element, multiband CPW antenna can be achieved

[8].

According to literature review of CPW antenna so far,

designed multiband CPW antennas are capable to operate as

quad-band antenna. Also, designed multiband CPW antennas by

Figure 1 The configuration of proposed multiband antenna: (a) single

resonance patch and (b) pentaband antenna. [Color figure can be viewed

in the online issue, which is available at wileyonlinelibrary.com]

1576 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 53, No. 7, July 2011 DOI 10.1002/mop