IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008 251

Torque Ripple and EMI Noise Minimization inPMSM Using Active Filter Topology and

Field-Oriented ControlKayhan Gulez, Member, IEEE, Ali Ahmed Adam, and Halit Pastaci

Abstract—This paper proposes an active filter (AF) topologyto reduce the torque ripple and harmonic noises in apermanent-magnet synchronous motor. The topology consistsof an insulated-gate bipolar transistor AF and two resistance–inductance–capacitance high-pass electromagnetic interference(EMI) noise filters, i.e., one in the primary and the other inthe secondary circuit of the coupling 1 : 1 transformer. The AFis characterized by detecting the harmonics in the motor phasevoltages by comparing the measured phase values with the ref-erence voltages generated as a function of the motor parametersand control setting values under field-oriented control. The AFuses the hysteresis voltage control method, while the motor maincircuit uses the hysteresis current control method; thus, the twocontrol methods independently work together to provide an almostsinusoidal voltage to the motor windings. The simulation resultsshow total harmonic distortion drops of greater than 13% withEMI noise damping down to ∼−10 dB as well as considerablereduction in torque ripple.

Index Terms—Active power filter, electromagnetic interference(EMI), permanent-magnet synchronous motor (PMSM), rotorreference frame, torque ripple, voltage harmonic.

I. INTRODUCTION

V ECTOR control with hysteresis controllers provides avery simple method to control the speed and torque of a

permanent-magnet synchronous motor (PMSM). It is desirablethat the voltages and currents provided to the motor terminalsdo not include harmonic components; in other words, they areperfect sine waves. However, various harmonic components arecontained in the motor windings due to many causes, such asstructural imperfectness of the motor, harmonics in the controlsystem associated with measurement noises, switching harmon-ics, and harmonic voltages supplied by the power inverter,which constitute the major source of unavoidable harmonicsin PMSM especially when the sampling period is greater than100 µs. These harmonics cause many unwanted phenomena,such as electromagnetic interference (EMI) noise, which affectsthe motor control system and the torque ripple, which thenprovides mechanical vibrations and acoustic noise.

Manuscript received March 3, 2005; revised September 21, 2007.K. Gulez and H. Pastaci are with the Department of Electrical Engineering,

Electrical–Electronic Faculty, Yildiz Technical University, Istanbul 34349,Turkey (e-mail: gulez@yildiz.edu.tr; hpastaci@yildiz.edu.tr).

A. A. Adam was with the Department of Electrical Engineering,Electrical–Electronic Faculty, Yildiz Technical University, Istanbul 34349,Turkey. He is now with the Faculty of Engineering Science, Omdurman IslamicUniversity, Omdurman, Sudan (e-mail: aliadam999@yahoo.com).

Digital Object Identifier 10.1109/TIE.2007.896295

Recently, many researchers have tried to reduce the torqueripple and harmonics in PMSM. Yilmaz et al. [1] presentedan inverter output filter topology for pulsewidth modulation(PWM) motor drives to reduce the harmonics of PMSM. Theproposed filter by Yilmaz et al. is composed of a conventionalRLC filter cascaded with an LC trap filter tuned to the in-verter line frequency. The scheme shows some effectiveness inreducing the switching harmonics; however, it requires tuningto adjust the trap filter for the switching frequency, and thatthe voltage harmonics are still high. Harrori et al. [2] andKim et al. [3] have proposed a suppression control method ofthe motor frame vibration and the rotational speed vibration ofPMSM by utilizing feedforward compensation control with ageneration of compensation signals to suppress the harmoniccontents in the d−q control signals by repetitive control andFourier transform. However, their work has nothing to do withthe switching harmonics and voltage harmonics provided bythe PWM inverter that supplies the motor. Many researchers[4]–[6] have addressed the active filter (AF) design to re-duce or compensate harmonics in the supply side by injectingharmonics into the line current, which has no effect on thecurrent supplying the load. Degober et al. [7] have proposedan approach to minimize the torque ripple of the surface-mounted PMSM caused by back electromotive force (EMF)harmonics. The approach used self-tuning multiple-frequencyresonant controllers in the Concordia reference frame with goodresults. However, the coefficients of the resonant controllershould be reevaluated according to the rotor speed while themotor operates, and the excitation current waveforms should bepredetermined according to the commanded torque and rotorposition. Stamenkovic et al. [8] have provided a model thatcan identify the torque ripple experienced with PMSM basedon measurement performed on a typical PMSM. They providedresults suitable for designing active and passive torque ripplecompensation. Gasc et al. [9] have proposed an approach toreduce the ripple torque without position sensor. The schemeutilizes a reduced-order torque observer and a Kalman filter.The scheme provided good results. However, accurate speed,currents, and line voltages are necessary to define the positionand load torque for the observer operations. Yun et al. [10]have proposed a variable step-size normalized iterative learn-ing control (VSS-NILC) scheme to reduce the periodictorque ripple. The VSS-NILC is combined with the existingproportional-integral current controller to minimize the meansquare torque error. The provided simulation results show someimprovements in minimizing the torque ripple. In [11]–[19], a

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252 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008

modification in the control algorithm and/or voltage modulationhas been carried to overcome the torque ripple problem. In[11] and [12], complicated and expensive multilevel invertershave been used to reduce the torque ripple and the fixingswitching frequency. In [13], a smooth torque has been obtainedby repetitive control techniques to the current control in afield-oriented PMSM drive. In [14], a method to reduce thecommutation torque ripple in a sensorless motor drive has beendeveloped. The developed method measures the commutationinterval from the terminal voltage of the motor and calculatesthe required PWM duty ratio to suppress the commutationtorque ripple. In [15], switching techniques, which insert zero-voltage vectors and/or more nonzero-voltage vectors to theconventional switching table, for ac drives with direct torquecontrol, have been achieved. In [16], an approach called con-duction angle control has been proposed to minimize the torqueripple in a doubly salient PMSM. In [17] and [18], space vectormodulation has been used to reduce the torque ripple with goodresults. In [19], a new direct torque control algorithm for the in-terior PMSM has been proposed to improve the performance ofhysteresis direct torque control (HDTC). The algorithm uses theoutput of the two hysteresis controllers used in the traditionalHDTC to determine two adjacent active vectors. It also uses themagnitude of the torque error and stator flux linkage positionto select the switching time from the suggested table structureto reduce the complexity of calculation. The simulation resultsshow adequate dynamic torque performance and considerabletorque ripple reduction. However, the hardware requirement isrelatively expensive.

In this paper, we propose a filter topology to reduce thetorque ripple and harmonic noises in a PMSM controlled byfield-oriented control (FOC) with current hysteresis controllers.The filter topology consists of an insulated-gate bipolar tran-sistor AF and two RLC filters, i.e., one in the primary circuitand the other in the secondary circuit of a coupling transformer.The AF is characterized by detecting the harmonics in themotor phase voltages and uses the hysteresis voltage controlmethod to provide an almost sinusoidal voltage to the motorwindings.

II. PROPOSED FILTER TOPOLOGY

When the PMSM is controlled by FOC with hysteresiscurrent controllers, the motor line currents are controlled tooscillate within a predefined hysteresis band. Fig. 1 shows thecurrent waveform and the associated inverter output voltageswitching.

In the figure, the inverter changes state at the end of asampling period only when the actual line current increases ordecreases beyond the hysteresis band, which results in a highripple current full of harmonic components.

To reduce the severity of this ripple, two methods can bementioned. The first method is to reduce the sampling period,which implies very fast switching elements. The second methodis to affect the voltage provided to the motor terminals in sucha way as to almost follow a sinusoidal reference guide. The lastmethod is adopted here. On that account, the AF topology, as

Fig. 1. Current waveform and associated inverter voltage switching of theFOC equipped with hysteresis current controllers.

Fig. 2. Basic structure of the proposed filter topology.

shown in Fig. 2, is used to affect the inverter voltage waveformto follow the required signal voltage as in Fig. 1.

Fig. 2 shows a schematic of the basic structure of the pro-posed filter topology, including the AF, coupling transformer,RLC filters, and block diagram of the AF control circuit.

In Fig. 2, Vsig is the desired voltage to be injected in order toobtain a sinusoidal voltage at the motor terminals, and VAF isthe measured output voltage of the AF. VAF is subtracted fromVsig and passed to the hysteresis controller in order to generatethe required switching signal to the AF. The AF storagecapacitor CF , which operates as the voltage source, shouldcarefully be selected to hold up to the motor line voltage. Thesmoothing inductance LF should be large enough to obtain analmost sinusoidal voltage at the motor terminals. The referencesinusoidal voltage V ∗, which should be in phase with the maininverter output voltage Vinv, is calculated using the informationof the motor variables.

The proposed filter topology consists of three parts, i.e.,one is the voltage reference circuit based on the space vector

GULEZ et al.: TORQUE RIPPLE AND EMI NOISE MINIMIZATION IN PMSM USING AF TOPOLOGY AND FOC 253

calculation, another is the AF part, and the other is the couplingpart, which consists of a 1 : 1 transformer and two RLC filters.In the coming sections, first, the operating principle of thevoltage reference control circuit will be explained, then, thetwo other parts will follow.

A. Voltage Reference Signal Generator

The effectiveness of the AF is mainly defined by the algo-rithm that is used to generate the reference signals required bythe control system. These reference signals must allow currentand voltage compensation with minimum time delay. In thispaper, the method used to generate the voltage reference signalsis related to the control algorithm of the motor, which uses themotor model in the rotor d–q reference frame and the rotor FOCprinciples with monitored rotor position/speed and monitoredphase currents. The motor model in this synchronously rotatingreference frame is given by[

vsd

vsq

]=

[R+ pLsd −PωrLsq

PωrLsd R+ pLsq

] [isd

isq

]+

[0eB

](1)

Te =32P (ψF isq + (Lsd − Lsq)isdisq)) (2)

wherevsd, vsq d-axis and q-axis stator voltages;isd, isq d-axis and q-axis stator currents;R stator winding resistance;Lsd, Lsq d-axis and q-axis stator inductances;p = d/dt, differential operator;P number of pole pairs of the motor;ωr rotor speed;ψF rotor permanent magnetic flux;eB = PωrψF , generated back EMF due to ψF ;Te generated electromagnetic torque.

Under base speed operation, the speed or torque controlcan be achieved by forcing the stator current component isd

to be zero while controlling the isq component to be directlyproportional to the motor torque Te as

Te =32PψF isq. (3)

The instantaneous q-axis current can be extracted from (3).Hence, by setting isd to zero, the instantaneous d- and q-axisvoltages can be calculated from (1) as

Vsd = − PωrLsqisq (4)Vsq =Risq + pLsqisq + PωrψF . (5)

Once the values of the d- and q-axis voltage components areobtained, the Park and Clarke transformation can be used toobtain the reference sinusoidal voltages as

v

∗a

v∗bv∗c

= K

1 0−1/2

√3/2

−1/2 −√

3/2

[cos θ − sin θsin θ cos θ

] [Vsd

Vsq

]

(6)

where K is the transformation constant, and θ is the rotorposition.

Fig. 3. Simplified power circuit of the proposed filter topology.

Fig. 4. Coupling circuit between the AF and the main inverter on one side andthe PMSM on the other side.

B. AF Compensation Circuit

Fig. 3 shows a simplified power circuit of the proposedtopology (the passiveRCL filters are not shown). In this circuit,Vdc is the voltage of the main inverter circuit, and V ±

CF is theequivalent compensated voltage source of the AF. In order togenerate the required compensation voltages that follow thevoltage signal vsig, bearing in mind that the main inverterchanges switching state only when the line current violates thecondition of the hysteresis band and that the capacitor voltagepolarity cannot abruptly change, the switches sw1 and sw2 arecontrolled within each consecutive voltage switching of themain inverter to keep the motor winding voltages within theacceptable hysteresis band.

The motor line current im is controlled within the motormain control circuit with hysteresis current controller to providethe required load torque; therefore, two hysteresis controllersystems (i.e., one for voltage and the other for current) areindependently working to supply the motor with an almostsinusoidal voltage.

In Fig. 3, when the switching signal (e.g., 100) is sent tothe main inverter, i.e., phase a is active high while phases band c are active low, then, following the path of the current imin Fig. 3, the voltage provided to the motor terminal can beexpressed as

Vs =23

(Vdc − V ±

CF − 32LF

dimdt

). (7)

The limit values of the inductor LF and capacitor CF can bedetermined as follows.

254 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008

TABLE IMOTOR PARAMETERS

TABLE IIL-TYPE FILTER PARAMETERS

Fig. 5. Motor phase voltage before and after applying the AF.

During a sampling period Ts, the change in capacitor voltagecan be calculated as

∆VCF =1CF

Ts∫0

imdt. (8)

So if the maximum capacitor voltage change is determined asVdc, the minimum capacitor value can be calculated as

CF ≥∣∣∣∣∣∫ (n+1)Ts

nTsimdt

V dc

∣∣∣∣∣ =∣∣∣∣Ts • imav

Vdc

∣∣∣∣ (9)

where imav is the maximum of the average current change thatcan occur per sample period.

The limit values of the smoothing inductance LF can beexpressed as

1(2πfsw)2CF

< LF ≤∣∣∣∣∣

VLFmax

32 max

(dim

dt

)∣∣∣∣∣ (10)

where the lower limit is determined by selecting the resonancefrequency of the combination CFLF to be less than the inverterswitching frequency fsw to guarantee reduced switching fre-quency harmonics. The upper limit is calculated by determiningthe maximum voltage drop across the inductors VLFmax and themaximum current change per sampling period dim/dt.

Fig. 6. Injected voltage from AF.

Fig. 7. Motor lines current before and after applying the AF.

Fig. 8. Motor torque before and after applying the AF.

C. Coupling

The coupling between the main inverter circuit and the AFcircuit is achieved through a 1 : 1 transformer, and to attenuatethe higher-frequency EMI noises, the LCR filters are used atthe transformer primary and secondary windings, as suggestedin Fig. 4.

The important point here is that the resonance that may arisebetween the capacitor C1 and transformer primary winding andbetween the capacitor C2 and motor inductance winding shouldbe avoided when selecting capacitor values.

At a selected cutoff frequency, the currents iCR1 and iCR2

derived by the RLC filters are given by

iCR1 =zT

zT +√R1 + 1/sC1

im1

iCR2 =zPMSM

zPMSM +√R2 + 1/sC2

im2 (11)

where zT and zPMSM are as defined in Fig. 4.

GULEZ et al.: TORQUE RIPPLE AND EMI NOISE MINIMIZATION IN PMSM USING AF TOPOLOGY AND FOC 255

Fig. 9. Rotor speed before and after applying AF.

Fig. 10. Phase a current (upper) and its spectrum (lower) before connectingthe AF.

At the selected cutoff frequency, these currents should belarge compared to im1 (which is drawn by the transformer)and/or im (which is drawn by the motor). On the other hand,at the operating frequency, these currents should be very smallcompared to im1 and im. Another point in the selection ofthe RLC parameters is that the filter inductors are essentiallyshorted at the line frequency while the capacitors are opencircuit, and for the EMI noise frequencies, the inductors areessentially open circuit while the capacitors are essentiallyshorted; thus, a considerable amount of EMI noises will passthrough the filter resistors to the earth and cause a frequency-dependent voltage drop across the inductors that in turn willhelp in smoothing the voltage waveform supplying the motor.

Fig. 11. Phase a current (upper) and its spectrum (lower) after connectingthe AF.

III. SIMULATIONS AND RESULTS

To simulate the performance of the proposed filter topology,Matlab/Simulink was used. The effectiveness of the filter topol-ogy was shown by providing the filter into operation while themotor is running.

The PMSM is star connected with earth return. The motorparameters are shown in Table I, while the passive filter param-eters are shown in Table II. The AF capacitor that was used is200 µF, and its inductors are 200 mH.

A. Motor Performance

The simulation results with 100-µs sampling time are shownin Figs. 5–13. Fig. 5 in particular shows the phase voltageprovided to the motor terminals. Observing the change of thewaveform after switching on the AF (at time = 0.15 s) intothe circuit, it is clear that the phase voltage approaches asinusoidal waveform. Fig. 6 shows the injected voltage fromthe AF. A better waveform can be obtained by increasing theAF inductance LF . However, the cost and size of the AF willincrease, so an acceptable inductance value can be selected toachieve less than 2% of the total harmonic distortion (THD).

The motor performances before and after applying the AFare shown in Figs. 7–9. In Fig. 7, the motor line currentsshow considerable reduction in noise and harmonic componentsafter applying the AF, which is reflected in a smoother currentwaveform.

256 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 55, NO. 1, JANUARY 2008

Fig. 12. EMI noise level before connecting the AF.

Fig. 13. EMI noise level after connecting the AF.

The torque response in Fig. 8 shows a dramatic drop intorque ripple from 3.2 to 0.2 N · m after applying the AF,which will result in reduced motor mechanical vibration andacoustic noise. This reduction is also reflected in a smootherspeed response, as shown in Fig. 9.

B. Harmonics and EMI Noise Reduction

The status of the line current harmonics and the EMI noisebefore and after connecting the AF are shown in Figs. 10–13.In Fig. 10, the spectrum of the line current before connectingthe AF shows that disastrous harmonics currents with THD of∼15% have been widely distributed with a dominant harmonicsamplitude of ∼16% in the range of thirtieth to fiftieth harmonicorder. After connecting the AF, the THD is effectively reducedto less than 1.5% with dominant harmonics amplitude of ∼1%in the range greater than the eight harmonic order, as shown inFig. 11.

The EMI noise level before connecting the AF in Fig. 12shows a noise level of ∼13 dB near the operating frequency,

∼−15 dB at the switching frequency (5 kHz), and less than∼−42 dB for the highest frequencies (> 0.3 MHz). Whenthe AF is connected, the EMI noise level is tuned down to∼−10 dB near the operating frequency, ∼−30 dB at theswitching frequency, and less than ∼−57 dB for the highestfrequencies (> 0.3 MHz), as shown in Fig. 13.

IV. CONCLUSION

In this paper, a new AF topology has been presented andanalyzed. The filter topology combines the compensation char-acteristics of the series AF and the L-type passive EMI filters.The proposed topology has been shown to be capable of re-ducing the torque ripple and current harmonics, and providingan almost sinusoidal voltage to the motor terminals, which wasreflected in a smoother line current waveform. The harmonicsdetection method is based on the same logic of the motorcontrol algorithm without time delay so that accurate referencesignals are expected. The topology has also shown effectivenessin reducing the EMI noise level that harms the motor controlsystem.

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[14] D.-K. Kim, K.-W. Lee, and B.-I. Kwon, “Commutation torque ripplereduction in a position sensorless brushless DC motor drive,” IEEE Trans.Power Electron., vol. 21, no. 6, pp. 1762–1768, Nov. 2006.

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Kayhan Gulez (S’92–M’00) was born in Istanbul,Turkey, in 1970. He received the B.S., M.S., andPh.D. degrees from Yildiz Technical University(YTU), Istanbul, in 1992, 1995, and 1999, respec-tively, all in electrical engineering.

Between 1994 and 1997, he was a ResearchAssistant with the Department of Electrical andElectronics Engineering, Engineering Faculty, CelalBayar University, Manisa, Turkey. In July 1997, hewas with the Department of Electrical Engineering,Electrical and Electronics Faculty, YTU. Between

1997 and October 1999, he continued to carry out his studies at the samedepartment. Between October 1999 and November 2002, he was a Re-search Associate in a JSPS project and other short-term projects with KeioUniversity, Yokohama, Japan, and with the Tokyo Metropolitan Institute ofTechnology, Tokyo, Japan. He is currently with the Department of ElectricalEngineering, YTU, where he has been an Assistant Professor since March2003. His major research interests are artificial neural networks and controlapplications; the control of electric machines, control systems, EMC, andEMI control methods; active, passive, and EMI filter design methods; andapplications for EMI noise and harmonic problems, on which he has over150 scientific papers and technical reports in various journals and conferenceproceedings.

Prof. Gulez was the recipient of three Science Grand Awards from YTU (in1998, 1999, and 2000), and two Best Paper Awards from the 5th World Multi-Conference on Systemics, Cybernetics and Informatics, and the Modeling andSimulation Conference in 2001.

Ali Ahmed Adam received the B.Sc. degree inelectrical engineering from Khartoum University,Khartoum State, Sudan, in 1991, the M.Sc. de-gree from Baghdad University, Baghdad, Iraq, in1997, and the Ph.D. degree in electrical engineeringfrom Yildiz Technical University, Istanbul, Turkey,in 2007.

Since 1991, he has been with the Faculty of En-gineering Science, Omdurman Islamic University,Omdurman, Sudan, where he is a Lecturer. His mainresearch interest includes power electronics, control

of electrical machines, digital control, microcontroller- and microprocessor-impeded systems, and active power filters.

Halit Pastaci was born in Macedonia in 1948. Hereceived the M.S. degree from Istanbul TechnicalUniversity, Istanbul, Turkey, in 1972 and the Ph.D.degree from Yildiz Technical University (YTU),Istanbul, in 1980.

From 1981 to 1982, he was with Colombia Uni-versity, New York, NY, where he carried out hisresearch activities. Since 1992, he has been withYTU, where is currently a Full Professor. Between1994 and 2001, he was a Consultant with the IstanbulMunicipality for the “Subway control systems of

Istanbul.” He has over 50 papers in various journals and conference proceed-ings, and is the author of ten books on his study areas. His research areas aresystem and biomedical control, neural network and fuzzy logic control andapplications, industrial electronics, and transportation systems.

Prof. Pastaci was an Associate Editor of the IEEE TRANSACTIONS ON

INDUSTRIAL ELECTRONICS between 2000 and 2005.