band 3-BLC functions well at the assigned center frequencies of

2.45 GHz. This methodology enables BLC to be reduced in area

by 61.64% (from 1262.42 mm2 to 484.19 mm2) with good toler-

ance to compromising the performance at the center frequency

while at the same time achieving 50% bandwidth within the

operating frequency band.

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VC 2013 Wiley Periodicals, Inc.

TWO NEW ELECTRIC COUPLINGSTRUCTURES FOR DOUBLE FOLDEDSUBSTRATE INTEGRATED WAVEGUIDECAVITY FILTERS WITH TRANSMISSIONZEROS

Guang Yang, Wei Liu, and Falin LiuDepartment of Electronic Engineering and Information Science,University of Science and Technology of China, Hefei 230027, China;Corresponding author: yangg@mail.ustc.edu.cn

Received 30 December 2012

ABSTRACT: Two electric coupling structures for double foldedsubstrate integrated waveguide (DFSIW) cavity filters are presented inthis article. The coupling strength can be easily tuned, and the coupling

mechanism is investigated. With electric coupling realized by theproposed structures in the main path and magnetic coupling by aconventional coupling structure in the cross coupling path, an X-band

cross-coupled filter is designed, which possesses a transmission zero oneither side of the passband. A prototype of the filter is fabricated to

validate the predicted frequency response, and the measured resultsagree with the simulated ones. VC 2013 Wiley Periodicals, Inc.

Microwave Opt Technol Lett 55:1815–1818, 2013; View this article

online at wileyonlinelibrary.com. DOI 10.1002/mop.27708

Key words: bandpass filter; substrate integrated waveguide; transmis-sion zero

1. INTRODUCTION

The substrate integrated waveguide (SIW) technique has proved

to be a low-cost solution for high performance millimeter wave

filters. Due to its advantages of high Q, light weight, and easy

integration with planar circuits, this technique is also becoming

popular with filters operating in lower microwave frequency

bands. However, a relatively large size of SIW filters to micro-

strip ones constrains their application. Half mode substrate inte-

grated waveguide (HMSIW) [1], substrate integrated folded

waveguide (SIFW) [2], and some more complicated structures

have been proposed to tackle this problem, and with these modi-

fied structures miniaturized SIW filters are designed [3–8].

A double folded SIW (DFSIW) cavity is an SIFW resonator

of a quarter guided wavelength, as shown in Figure 1, or it may

be regarded as an SIW cavity folded along the two symmetrical

planes of AA0 and BB0. It is shown that the quality factor of a

DFSIW cavity is comparable to that of an SIW cavity, even

though the area reduction is as much as 75% [3]. Typical

DFSIW filters are direct coupled ones [4,5], where adjacent cav-

ities are magnetically coupled by inductive windows. Besides,

Figure 1 (a) Configuration of a DFSIW cavity, (b,c) Magnitude of the

fundamental electric field distribution in an SIW cavity and a DFSIW

cavity. [Color figure can be viewed in the online issue, which is avail-

able at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 8, August 2013 1815

cross-coupled filters with DFSIW cavities have also been pro-

posed [6–8]. In [6], both the main coupling and the cross cou-

pling are achieved by identical coupling structures, and thus the

filter response bears no transmission zero (TZ). The filter in [7]

has a pair of TZs, but the cross coupling is implemented

between half guided wavelength SIFW resonators, which

increases the occupied area. A capacitive vane coupling struc-

ture is proposed to achieve electric cross coupling in [8], and

with inductive window coupling structures in the main path the

designed filter has two TZs.

This article proposes two new electric coupling structures for

DFSIW cavities, which are simpler than that in [8] with via walls

between adjacent cavities removed and facilitate the design of

cross-coupled DFSIW filters with TZs. Based on these new struc-

tures, a four pole cross-coupled filter is implemented, which is

specified to operate at 10 GHz with a fractional bandwidth of

4.5% and two TZs at 9.5 GHz and 10.5 GHz, respectively.

2. FILTER DESIGN

The configuration of the proposed filter is demonstrated in Fig-

ure 2(a), which has three metal layers and two dielectric layers.

The first metal layer excluding input/output lines constitutes

ground planes together with the third metal layer by metallic

vias through both dielectric layers. The second metal layer is

connected with input/output lines by metallic vias through the

first dielectric layer. Four DFSIW cavities make up resonant ele-

ments of the filter, with the layout shown in Figure 2(b).

The filter is designed using the coupling matrix synthesis

method. A four order lowpass filter with a single pair of TZs

and a passband return loss of 20 dB is chosen as the prototype.

The coupling matrix and the external quality factor are found to

be [9]:

M 5

0 20:0395 0 0:0065

20:0395 0 20:0341 0

0 20:0341 0 20:0395

0:0065 0 20:0395 0

2666666664

3777777775

Qe1 5 Qe4 5 21:144

(1)

where Mi,i11 denotes the coupling coefficient between resonator

i and resonator i11, while Qe1 and Qe4 represent the input and

output external quality factors.

2.1.Proposed Electric Coupling StructuresThe orthogonal slot of a DFSIW cavity behaves as a magnetic

wall boundary, where the electric filed is much stronger than

the magnetic field and electric coupling is likely to occur. How-

ever, the metallic via wall between adjacent DFSIW cavities

shields the electromagnetic field in either cavity and prevents

inter-resonator coupling when slots of two DFSIW cavities are

located by the common side via wall. As the orthogonal slots by

the common side via wall both behave as magnetic walls, the

common side via wall can be removed without interfering the

field distribution in a DFSIW cavity. Moreover, coupling

between adjacent cavities is now possible and the consequent

coupling structure is much simpler.

The first coupling structure is shown in Figure 3, which is

motivated by interdigital microstrip filters. Adjacent cavities are

interdigitally positioned so that the strong electric field region is

in the vicinity of the weak electric field region. An eigen mode

solution analysis is conducted by HFSS to find out the first two

modes. As shown in Figures 3(c) and 3(d), the odd mode has a

lower resonant frequency than the even mode, which indicates

that the structure realizes electric coupling. Figure 4 is the

equivalent circuit, where both LC tanks share the same resonant

frequency, that is, f0 5 1/(2pL1C1) 5 1/(2pL2C2). The odd and

even mode resonant frequencies of the coupled tanks are derived

in Eq. (2), and fodd is less than feven due to electric coupling.

fodd 5f0ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi

11Cm L11L2ð Þp

feven 5 f0

(2)

where fodd and feven correspond to the resonant frequencies of

the odd and even mode, respectively.

Unlike the first coupling structure, in the second coupling

structure cavities are positioned so that the strong electric field

Figure 2 (a) 3D configuration of the proposed filter (ML: Metal

Layer, DL: Dielectric Layer), (b) 2D layout of the middle metal layer.

[Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 3 (a) Configuration of the first coupling structure when l1 5 0,

(b–d) configuration, lower mode and higher mode when l1> 0. [Color

figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 4 Equivalent circuit of the electric coupling structure

1816 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 8, August 2013 DOI 10.1002/mop

region is in the vicinity of the strong electric field region, as

shown Figure 5(a). However, in this case the coupling coeffi-

cient is very small and does not satisfy that of the coupling ma-

trix. Actually, this structure may be regarded as a half guided

wavelength SIFW resonator with a slot in the center. As the slot

hardly disturbs the surface current distribution, the achieved

coupling is very weak. For stronger coupling, the structure in

Figure 5(b) is proposed, and by eigen mode analysis it realizes

electric coupling, as shown in Figures 5(c) and 5(d). With a T

shaped slot, the surfaced current distribution is disturbed more,

resulting in much more current component perpendicular to the

slot, and thus the coupling gets much stronger. The coupling is

strengthened with the increase of the dimension l1, which facili-

tates tuning of the coupling coefficient, as shown in Figure 6.

2.2. Filter Coupling TopologyFigure 7 shows the coupling topology of the proposed filter.

Unlike [8], the main path coupling is realized by the proposed

electric coupling structures, while the cross coupling between

resonators 1 and 4 is realized by the conventional inductive win-

dow coupling structure, which is magnetic in nature.

Figure 5 (a) Configuration of the second coupling structure when

l1 5 0, (b–d) configuration, lower mode, and higher mode when l1>0.

[Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 6 Coupling coefficient between R2 and R3 versus l1. [Color fig-

ure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 7 Coupling topology of the proposed filter

Figure 8 Structure of the GCPW to strip line transition. [Color figure

can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 9 Coupling coefficients (a–c) and external quality factor (d)

versus filter dimensions

Figure 10 (a,b) Top and bottom metal layer of the first substrate,

(c,d) Top and bottom metal layer of the second substrate, (e) Assembled

filter prototype. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 8, August 2013 1817

The input and output lines of the filter are grounded coplanar

waveguides (GCPW) [8], which are connected with the tapped

strip lines by two metallic vias through the first dielectric layer.

The configuration of the transition structure is shown in Figure 8.

2.3. Realization of the FilterRelations of the coupling coefficient and the external quality factor

to filter dimensions are derived and shown in Figure 9 [9]. Initial

values of these dimensions are determined according to Eq. (1), and

after tuning and optimization, they are finally decided as follows:

l1x 5 l1y 5 l2x 5 l3x 5 l4x 5 l4y 5 7.15 mm, l2y 5 l3y 5 5.65 mm, s12

5 s34 5 0.36 mm, s23 5 0.4mm, w14 5 3.3 mm, o 5 1.85 mm, s1 5

s3 5 s4 5 s5 5 s6 5 0.3 mm, s2 5 0.4 mm, l1 5 4 mm, l2 5 4.6

mm, w1 5 0.55 mm, w2 5 8 mm, w3 5 2.55 mm, w4 5 2.85 mm,

d1 5 0.5 mm, d2 5 1 mm, p1 5 0.97 mm, p2 5 p4 5 p6 5 1 mm,

p3 5 0.9 mm, p5 5 0.85 mm.

3. RESULTS AND DISCUSSION

Both layers of the filter are fabricated by single layer printed

circuit board (PCB) process on a 20 mm thick Rogers5880 sub-

strate with permittivity of 2.2 and loss tangent of 0.0009, which

are assembled together afterwards. The middle metal layer is

printed on both layers of substrate to ensure good contact after

assembly. A photograph of the fabricated filter is shown in Fig-

ure 10. Figure 11 shows the measured results compared with the

simulated results by HFSS from 9 to 11 GHz. The measured

insertion loss (IL) is around 2.7 dB and the return loss is better

than 10 dB from 9.79 to 10.34 GHz. There are two TZs at 9.53

GHz and 10.71 GHz of the stopband, which greatly improves

the selectivity of the filter. The first spurious passband (FSP)

appears at 22.2 GHz, as illustrated by Figure 12. A comparison

of the filter performance to other DFSIW ones is listed in Table

1. The performance of the filter is expected to improve if fabri-

cated by low-temperature co-fired ceramic (LTCC) process.

4. CONCLUSION

Two novel electric coupling structures are presented in this arti-

cle for DFSIW cavities. With grounded via walls between

DFSIW cavities removed, these structures are simple and realize

electric coupling easily. A cross-coupled filter with two TZs is

designed, which has an electric main coupling and magnetic

cross coupling topology, and a prototype is fabricated to verify

the predicated response. The proposed coupling structures enrich

the DFSIW coupling structure family and facilitate the design of

cross-coupled filters with TZs.

REFERENCES

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VC 2013 Wiley Periodicals, Inc.

Figure 11 Simulated and measured S parameters of the filter. [Color

figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 12 Wideband transmission response of the filter. [Color figure

can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

TABLE 1 Performance Comparision

Work Frequency/ FBW IL (dB) TZs (GHz) FSP (GHz)

[4] 4.675G/7.5% �2 None 9.65G

[5] 5.55G/10% 1.15 None 10G

[6] 7.8G/6% �3 None 16.8G

[8] 8G/7.5% 2.6 7.5/8.45G 16G

This work 10G/4.5% 2.7 9.53/10.71G 22.2G

1818 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 8, August 2013 DOI 10.1002/mop