wireless world circard

Wireless World Circard Series 7:I

Power amplifiers-l 1

Basic power amplifiers Emitter current (and hence 1~) is defined because the p.d.across R, equals the p.d. defined across R, minus the T/be ofTrl. For silicon transistors this is 0.6V andis stable to within

RI lll#o-l

K

+v, Supply: +15v 10 or 20% for most transistors under most operating condi-

Tr: BFYSO tions. The resulting p.d. across R8 is again a compromise

% R,: 1.2kS2 between high values for better stability and low values for

R,: 12052 minimum wasted power - not less than OSV and not greaterthan say 20% of supply voltage as a guide for power stages.

ct R,: lOS-2T: 3.25:l turns ratio Capacitor Ca decouples Rs to prevent negative feedback

aa Quiescent current : 70mA within the required frequency range. As R, may be a lowR3 c2 Output power into 2552 resistance, Cz must then have high capacitance.

load : -400mW for 10%distortion Class C

The basic principle behind class C amplifiers is simple, theClass A efficient realization difficult. The transistor conducts only onThe classic transformer-coupled class A amplifier has been positive peaks of the input signal with the RC time constantsuperseded for most purposes, but may still be applied where determining the angle in the cycle for which conductiongood isolation is required between source and load, or where continues, the base-emitter of the transistor acting as a diodethe optimum impedance for maximum undistorted output is and allowing C to charge during the peak. The current in thevery different from the load impedance. Resistors R1 and Rz output circuit is then in the form of pulses of current of whichfix the base potential of Trl provided the current through them the fundamental term flows in the load if the LC circuitis much greater than the base current. This base current is the resonates at the fundamental frequency. A high-Q circuitrequired collector quiescent current divided by transistor ensures that the harmonics are sharply attenuated giving goodhFE. These parameters fix the value of RI + R, by the approxi- output waveform simultaneously with high efficiency. A widemate relationship R, + Ra = hFE Vs/m ZC. The VdUe of m range of load and source impedances can be accommodatedthe ratio of divider current to base current, is a compromise by introducing suitable LC networks at input and outputbetween stability and wasted power. Typically m = 5 to 20. (see card 6).

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Typical data

iI%l bV’ +V Supplies V = f6V,

TP, V’ = &15VTr,: BFR41

m%

L

+ RI Trz: BFR81I IC,: 741 RL

RL R,: 47052Tr2 RL: 1552 Class D____________-- -J

Output power 0.92W at In the class D amplifier, one or more transistors act as-v* - V

64 % efficiency switches, connecting the drive point of an LR series circuit toOutput voltage swing: the supply lines. This delivers a square wave to the LR circuit1OSV pk-pk for f6V and provided the reactance of the inductor is high at the

Class B supply switching frequency there is little output. If the duty-cycle of

The complementary pair of transistors acting as emitterthe input waveform is altered the output will have a mean

followers comprise the basic class B push-pull stage. Tran-level which is a function of the duty cycle. A frequency lowerthan that of the basic switching frequency is used to modulate

sistor Trl conducts during the positive half-cycle and Tr,during the negative half cycle. For input voltages close to zero

the pulse-width/position of the square wave generator and thelow voltage is then a function of that signal voltage. For ideal

neither transistor conducts as each requires a finite base- transistors there is no power lost at the switching frequencyemitter voltage for conduction to commence (-0.5V for and the overall efficiency can approach 100%. Diodes clampsilicon devices). Non-linearities at low-levels make direct the output voltage to the supply lines. The drive voltage mustvoltage drive at the bases unattractive, with the resulting be large enough to saturate the transistors.cross-over distortion being very apparent in badly designedamplifiers of this time. If the output stage is included within Further readingthe feedback loop of a high-gain amplifier the negative feed- Oxborne, M. R., Design of tuned transistor power amplifiers,back reduces the distortion very considerably. At high Electronic Engineering, 1968, pp.436-43.frequencies the falling gain of the op-amp prevents the feed- Stewart, H. E., Engineering Electronics, Allyn & Bacon 1969,back from being fully effective and the crossover reappears. pp.589~642.Voltage-gain as shown is unity, but standard feedback Birt, D. R., Modulated Pulse Amplifiers, Wireless World,networks may be used to obtain any desired voltage gain. 1963, pp.76-83. (Also subsequent articles and letters.)Output may be increased to 1.75W into 852 but heat-sinkingis then advisable. If the objectionable audio effects of cross- Cross referencesover are to be minimized biasing networks are inserted Series 7, cards 4, 5, 9, 10, 11 (class A), 2, 3, 7, 8 (class B),between the transistor bases. 6 (class C), 12 (class D).

Q 1973 IPC Business Press Ltd.

Wireless World Circard Series 7: Power amplifiers-2

Servo amplifier

FREQUENCY ( H z )

Typical performanceSupplies : & 15V, 235mAQuiescent current :&1.8mAA,: 741Trl: BFRSITr,, Tr,: TIP3055Trs: BFR41R1, R,: 15MR,: 4752; Ra, R,: 4.7kSZR,, R,: 18052C, : 2nF; C,: 1OOpFC,: 4.7nF; Cd, Cg, C,:470nFDr, D,: SP2; D,: lN914RL: 18aRisetime z 30~s (4.8Vpk-pk at 1kHz)Vi*: 2.07V r.m.s. with-out clipping

In servo systems a servoamplifier is needed when a high-powerload must be driven from a low-power source. Amplifier A1acts as a see-saw amplifier having its gain determined by R2/Rowhich can be adjusted to accommodate a wide range of inputsignal levels from a transducer. With no input signal, theoutput power transistors are virtually cut off, the only drainfrom the supply being the quiescent current of the operationalamplifier (around 2mA). Hence the base-emitter junction ofTr, is forward-biased by only about 350mV due to the p.d.across R,. The base-emitter junctions of Tr, and Tr, would beforward-biased to a smaller extent unless R, was greater thanR,. However, including Ds and making RB = R, produces thedesired bias with D, providing some temperature compensa-tion for the base-emitter voltage of Tr,. The amplifier has aclass B push-pull output stage so that a bipolar input signalproduces class B currents in its supply leads. These currentsare used to provide the base drive to the compound powertransistors which supply the load currents to R1 in push-pull.Transistors Tr, and Tr, form a Darlington pair while Tr, andTr, are its complementary equivalent. The Darlingtonconfiguration is used to provide high current gain to ensurethat the load current is much larger than the amplifier’squiescent current. To guard against instability, R1 and C1provide feedback around the operational amplifier and Rs andCs provide feedback around the power stage. Bandwidth ofthe amplifier is controlled by C,R, time constant which canbe held fixed when the gain is varied by Rz, if CZ is alsoadjusted. Diodes D1 and Da protect the output transistorsagainst breakdown when the load is highly inductive.

-+v

Component changesUseful range of supplies: f6 to &- 18V.Output power and efficiency fall as supply voltage is reduced:typically Pout is 0.8W and efficiency is 65% with &6V at1kHz. With maximum drive, Pout falls as RL increases: forsupplies of f15V, typically, Pout is 12.6W for RL = 6.852 andPout = 3.8W for RL = 2552. Total harmonic distortion falls

b %t

D4 R2Tr1

~

R, Cl- -- + aa

RL,

R6 Tr2

A -

+a5 a3

RE2 ,D3

- V

as drive increases: typically 0.45 % for vi, = 2.8V and 5.3 %for fin = 150mV (supplies 115V, RL: 1852 and f = 1kHzsinewave).

.l In principle, any other feedback configuration may beused; for example taking the input signal to the non-inverting

Circuit modification input of the operational amplifier and grounding the inputl The Trr-Tr, and Tr,-Tr, Darlington pairs in the output end of Rd converts the feedback to a series-applied form withstage may be made single n-p-n and p-n-p transistors. Ideally, the accompanying increase in input impedance. (See circuitthese transistors should have high current gains to provide a right.) The operational amplifier may be supplied withpeak load current that is significantly in excess of the quiescent differential input signals if desired.current in the amplifier. They also need to have a higher powerrating and the combination of high power, high current gain Further readingand wide bandwidth is not an easy specification to meet at Campbell, D. L. & Westlake, R. T., Build a high-currentlow cost. The use of single BRF81 and BRF41 transistors servoamplifier with i.cs, Control Engineering, December 1969,provides a reasonable compromise. pp.91-4.l A modification which can improve stability while allowing Garza, P. P., Getting power and gain out of the 741-typesome quiescent current in the output stage, i.e. biasing in class op-amps, Electronics, 1 Feb., 1973, p.99.AB, is obtained by including resistors in the equivalent emittersof the drive transistors, increasing the p.d. across R, and R, Cross referencesand/or placing a diode in series with R, and R,. The resistors Series 7 cards 1 & 12.in the emitters can be selected to provide the required quiescent Series 2 card 4.current. (See circuit left.) Series 4 card 8.

Q 1973 IPC Business Press Ltd.


Tr, : TIS4j; Tr, : TISSOICI: l/6 SN7406R,: 470;R,: 1OOsZR,: 1052D,: PSlOl ; C,:680pFInput pulse height : 4VDuration : 600nsP.R.F.: SOkHzRise time: 20ns

Circuit descriptionThe complementary symmetry output stage commonlyused in class B amplifiers is equally applicable to pulse out-puts. The problem here is that using only a single transistor inthe output will only allow any capacitive load to have eithera fast rise time or a fast fall time, but not both. Or if the outputstage is operated in class A, it needs a quiescent current greatlyin excess of the charging current required by the capacitorto achieve a high rate of rise and/or fall. The class B push-pullstage shown has Trr driving the capacitor in the positivedirection when a positive-going edge is applied at the baseconnection, while Trz drives the capacitor in the negativedirection. Rise and fall times are now determined by thecurrent flow in the capacitor, which on the positive-goingedge is limited by the base current that can be supplied by R1

Output pulse: rise time49 ns; fall time 32ns;pulse height : x V1(Rise and fall timesmeasured between 10%and 90% levels). Varia-tion of rise and fall timewith several capacitiveloads shown opposite.Some small distortioneffects on input drivepulse were not apparenton the output pulse.

as D1 is allowed to conduct. On the negative-going edge, currentthrough Rz is significantly greater and could cause excessivecurrent flow in Tr, but the diode is reverse biased and R,takes the place of limiting action previously provided by R1.It is not possible in a simple circuit of this kind to choose asimple bias network for R, and Rz which would give the samebias drive current in both directions.IC1 is a open-collector high-voltage output device whichpulls the potential at the bases of Trr and Tr, to a low valuewhen in conduction, and when out of conduction allows thebias to rise towards V1 via Rr.

Rl Tr4-3 __R4

Tr2

Component changesl Transistors Trl and Tr, can be replaced by BFR41 & zero and hence pulse rise times will be slightly larger than inBFR81 or BC125 & BC126 with poorer rise and fall times. the original circuit.Typical comparison l An alternative arrangement is shown right. If the drive

rise time (ns) fall time (ns) voltage goes positive, the Zener diode transfers current to theTIS45/50 12 12 base of Tr, which brings Tr, into conduction, clamping theBC125/126 28 14 output to the negative supply rail, with very small saturationBFR41/81 38 15 effects. Conversely, if the output swings negative Tr, conducts

l For each capacitor value, overshoot on leading and and clamps the output to the positive rail, i.e. the peak-to-peaktrailing edges of output pulse is approximately 25% of pulse output swing into the load is almost equal to the supply raillevel. values.l Resistive load: lOOQ, V,: 14V, V,: +5V; output pulseexcursion is from 1.6 to 12V. Further readingPulse width: 6~s. Useful frequency range 3 to 1OOkHz. Texas Instruments Technical Seminar 1972, m.o.s. memoryCorresponding mean current from supply 1.5 to 30mA d.c. drivers.l ICI: SN75451A or SN7407 for greater output voltage levels SGS-Fairchild, Industrial Circuit Handbook, 1967, p.38.and faster rise times. Williams, P., Voltage following, Wireless World, vol. 74, 1968,

pp.296.Circuit modificationl Rise and fall times for the circuit left are given centre. Cross referencesThe lower level of drive pulse from the i.c. is approximately Series 6, cards 1, 2 & 8.

0 1973 IPC Business Press Ltd.


Push-pull class A power amplifier

Trz, Tr,: TIP3055Tr,: BFR41C,: 2,OOO~FC,: 470,eFC,: 1OOpF

TCl R1, R,: 22052

R,: 25OQR,: 47052R,: 1208R: 352Supply: 12 to 14v

Circuit descriptionClass A push-pull amplifiers have at least two active devicesin the output stage, and each device should operate under thesame quiescent conditions. A drive circuit using one or moredevices provides antiphase signals to the output pair whichshould have matched parameters. Thus a minimum of threetransistors is called for and more are commonly required. Byusing current phase-splitting, a simple circuit results whichstill gives adequate efficiency and distortion figures. The keyfeature of the circuit is that the current in Rz remains constantthroughout the a.c. wave form while its d.c. value can beadjusted to set the desired quiescent current. Bootstrapping

Typical performanceFor supply of 13V, quies-cent current of 950mA,max. output for 3%distortion is 12V pk-pkinto 552 (3.6W). Meancurrent falls to 820mAat max. output. Fullpower bandwidth: 20Hzto 1OOkHz. Hum and

noise: 80dB below fulloutput. Quiescent cur-ren t : 1.25A @ 1 3 V .Output power: 5W into352 @ 5% i.m.d. Distor-tion: < l.l%, 1W into3 5 2 , 1OOHz to 10kHz.Voltage gain N -2. In-put impedance N 25052.

via C1 ensures that any increase in the collector potential ofTr, is transferred via the emitter follower action of Tr, toreappear at the junction of R1 and Rz. Hence the charge inp.d. across Rz approaches zero except at very low frequencieswhere the reactance of Cr becomes significant. As there is nochange in Rz current, any increase in Tr, current increases thebase current of Tr3 while reducing the base current of Tr, bysubstantially the same amount. Accurate current phase-splitting together with matched current gains of Tr,, Tr, keepthe distortion low. Overall negative feedback via R, definesthe output quiescent voltage as a multiple of the base voltageof Tr, (N 1.3V) and the ratio RJR, scales this base voltage upto half the supply voltage, i.e. the output transistors operatewith equal Vce as well as equal 1~.

Component changes gain, allows 100% d.c. series-applied feedback and has inputTr,, Tr,: Power transistors with closely matched hrn at feedback and load all referred to same supply line. Thisoperating current. Quiescent power (at least twice max. eliminates bootstrap capacitor provided speaker can tolerateoutput) determines types and heat sinks. direct quiescent current of driver stage. For output at mid-2N3055 for PO > 5W. MJE521 for PO > IW. point of RB z R7. Voltage gain % (R,/R,) $1. Reactance ofBFYSO, BFR41, etc., for PO < 1W. C, < R6 at lowest frequency of interest. Typically Rq, R, : 1 toTr,: BFYSO, BFR41, 2N3053 for most applications. lOks2, Rg, R,: 20 to 2OOksZ. Other values as before.C,: Reactance < RI, at lowest freq. Typically 200 to 5000,~F. l For higher input impedance, input potential divider mayC,: Reactance <RI at lowest freq. Typically 100 to 500,~F. be bootstrapped. Interchanging locations of R,, Cz allows R,Ri, R,: Set output current Vs/2(R, + R,) w 2zs/hFE. One to be bootstrapped, almost doubling input impedance.resistor made variable to adjust mean current. Typical range l Quiescent current depends on current gains of Tr,, Tr,.1OOQ to lkS2 (higher values for low-power circuits). By monitoring circuit mean current and using result to controlR,: Sets voltage gain N - RJR, and input resistance N Rg. drive current Tr,, mean current can be made constant, e.g. forR,, R,: Set output voltage (quiescent) to N 2 Vbe [(R3/R4) + 11. Tr, a germanium transistor, Dr a silicon diode, mean p.d.Current in RB, Rd to 5 to 20 times base current of Tr,. Typical across R, is controlled at 0.4V.values R,: 100 to 5OOQ. R,: 30052 to 3kS2.

Farther readingCircuit modification Linsley Hood, J. L., Simple class A amplifier, Wireless World,l Open-loop gain of the original circuit is low and feedback vol. 75, 1969, pp.148-53.that can be used may not reduce distortion sufficiently. Simple Markus, J. (ed.), Improving signal transfer in Electronicsbias circuit leaves the output at a fixed multiple of Vbe rather Circuits Manual, 1971, p.19.than at the supply centre point, i.e. resistors require readjust- Allison, W., Self-biasing class A power amplifier, Wirelessing for different supply volts. Adding Tr4 increases open-loop World, vol. 78, 1972, p.577.



High-voltage amplifier+v

RI

V0ThITT2

R2

Typical performanceSupply : + 1OOvTrl: MJE340Trz: 2N3810R,: lOk.C2; R,: 45652Quiescent voltage: 52VVB: 1ov

Input signal: 7V pk-pk

Gain constant up to2OkHzVariation of output withRa not decoupled shownopposite.Effective output impe-dance of transistor con-

Output voltage: 84V pk- figuration 5MQ.pk

The characteristics required by an amplifier may include highvoltage gain and in some applications the ability to withstandhigh output voltages simultaneously. Such a combination isnot available within a single device, but the circuit shownarranges that the necessary input impedance gain charac-teristics are obtained by TrZ and the high voltage charac-teristics by Tri. The input characteristics aimed at were thatthe device should behave with a defined gain, so that the wholesystem could be considered equivalent to a value. TransistorTr, is thus a field-effect transistor whose gain is controlled bythe quiescent current, which may be set by R,. The drain ofTr, feeds into the emitter of Tr, whose base is maintained at aconstant potential, just high enough to ensure that Tr, has aquiescent voltage that is above its pinch-off value. The biasvoltage should be obtained from a low impedance circuit.

Hence TrZ is operating into a low impedance, while Trr isvirtually a common-base stage and has thus the highestvoltage rating that it could possibly have. The current at thecollector of Trl is essentially the same as the emitter currentas the current gain from emitter to collector is nearly unity.There is no significant Miller/Blumlein effect between thecollector of Tr, and the gate of Tr, as the voltage swing at thecollector is isolated from the gate of TrZ. The capacitancebetween Trr collector and base is now effectively a capacitanceto ground rather than to the input of the amplifier. Howeverthis capacitance still affects the output characteristics, as it isin parallel with R, for a.c. and determines the bandwidth ofthe amplifier. The problem is more severe than in many low-voltage amplifiers because R, will have a much higher valuefor a given quiescent current because the p.d. across it may bein excess of 1OOV. This is the usual penalty to be paid for ahigh-voltage gain, i.e. the associated high load impedancewill have a longer time constant for a given capacitance. Thevoltage rating is close to the VCE breakdown of Tr,.

Circuit description

Component changesl Decouple Rz with 150pF capacitance to retain gm of thecombined transistors. Output 82V pk-pk for an input signalof 2.4V pk-pk. Low frequency cut-off then 5Hz.l Range of Vn 8 to 11V - value not critical, with no signi-ficant effect on performance.

l Supply may be increased up to 300V with appropriatechanges in R1 and R, to control quiescent voltage. Typically(i) SUPPIY: +2OOV, VQ: llOV, R,: 122R, Vin: 7V pk-pk; Vout:180V pk-pk; R,: 1OkQ. (ii) supply: +3OOV, VQ: 15OV, R,:lSkR, R,: 68kQ Vout: 2754 pk-pk.0 Increase of +V from 100 to 2OOV, maintaining circuitresistors constant reduces h.f. cut-off by approximately 20%indicating that this is dependent more on external componentsrather than operating conditions.

Circuit modificationsl F.E.T. gm is controllable by varying negative feedback.A wide range of control is possible with the circuit shown left.Because the gate is at positive, RE can be large for chosenquiescent value of drain current, the feedback being varied viaC without then altering the d.c. state of the circuit.

l F.E.T. g, can be boosted by adding a p-n-p bipolartransistor to achieve a complementary pair (centre), or ann-p-n transistor for a Darlington pair (right), as the outputimpedance may be considered to be approximately l/gm. Theeffective output impedance is less than that of the f.e.t alone.

Further readingDesigners casebook. Electronics, 1 Feb., 1973, p.99.Greiter, O., Transistor amplifier output stages, WirelessWorld, vol. 69 1963, pp.310-3.High voltages switched with a single transistor, 400 Ideas forDesign, Vol. 2, 1971, Hayden.

0 1973 IPC Business Press Ltd


Class C power amplifierTypical dataSupply: 12vTr,, Trz: BFR41R,: 10052; R,: 5052(carbon)C,: 180pF; C,: 36OpFC,: 47pF; C,: 1OnFC,: 5OOpF; C,: 190pFC,: 805pF; L,: 2.7fiHL,: 2.16,~H; L,: 2.38mHL,: 230pH; L,: 1.51fiH

Circuit descriptionMany class C amplifiers find application in the v.h.f. and u.h.f.bands, special transistor fabrication techniques being used tooptimize their performance. For correct design it is necessaryto establish a suitable model for the transistor behaviour underclass C conditions, some manufacturers providing the appro-priate data. In general, this data is not available for class Cdesigns operating at frequencies lower than about lOMHz, sothat a successful circuit normally results from a breadboardversion using variable capacitors. The circuit shown abovewas produced on this basis where Cz, Cs, C,, Cg, C, wereoriginally fixed capacitors ‘padded’ out with variables.Source and load resistance were 5OQ and the output powerobtained at 7.2MHz was 1.41 W with a drive signal producing

250mA supply current. Overall efficiency was only 47% (seegraphs) but taking account of the d.c. drop (3.52V) across ther.f. choke L, efficiency rises to 66.5 %. Hence L4 should havelow resistance, but its effect is less noticeable at lower currents.Transistors Tr, and Tr, were general-purpose transistorsconnected in parallel to reduce dissipation problems. Thetuned networks in the input and output circuits should matchthe source to the transistors and the transistors to the load formaximum power transfer. Careful layout is essential and thecircuit can easily oscillate as L,, L1 and the collector-basecapacitance of the transistors form the basic arrangement of aHartley-type oscillator.

Component changesThe circuit can operate over a limited frequency range and awide range of supply voltages and power levels provided theinput and output networks are re-adjusted to cater for thechanging values of transistor input and output resistance andcapacitance.

J-0 +“cc

Alternative general-purpose transistors can be used, such asBFYSO.Single transistor can be used when reduced power is accept-able.Input transformer can be dispensed with if alternative inputand output networks used (see over).

Circuit modificationsCorrect design procedures for class C r.f. power amplifierstend to be highly analytical due to the need to consider thecorrect choice of input and output coupling networks, theirworking Q-factors, degree of harmonic rejection, possiblecauses of spurious oscillation and the d.c. operating conditions.For a successful design the impedances at the transistor inputand output terminals must be known under the desired operat-ing conditions. Use of small-signal parameters leads to con-siderable errors in a class C design as the voltage and currentswings are so large in such a power amplifier. When class Ctransistor data is available it is normally provided in the formof equivalent parallel input resistance and reactance andparallel output capacitance as a function of frequency andpower output. The equivalent parallel output resistance is

given approximately by R = V2cc/2.Pout. Even with this dataavailable a choice must be made from the large number ofpossible input and output coupling networks. Often a T-configuration is suitable for both networks as shown left. Thesenetworks complex-conjugate match the source to the transistorand the transistor to the loads. Both networks introduce lossesdue to component imperfections. Choice of the workingQ-factors is a compromise between losses in the couplingnetworks, their selectivity and realizable component values.If the loaded-Q is high the capacitors will be small, the

.selectivity will be high but the losses will be large. A lowworking Q-factor implies the opposite. When the availabledata is correctly interpreted it will normally still be necessaryto tune the amplifier for optimum performance, for exampleby adjustment of C1 to C.,. Complete design procedures aregiven in the first three references.

Further readingMotorola, application note AN-282: Systemizing r.f. poweramplifier design, 1967.Hilbers, A. H., On the input and load impedance and gain ofr.f. power transistors, Electronics Applications, vol. 27, 1967,pp.53-60.Mulder, J., On the design of transistor r.f. power ampli-fiers, Electronic Applications, vol. 27, No. 4, pp. 1967, 155-171.Markus, J. (ed.), ~-MHZ, 3-W amplifier, in Electronic CircuitsManual, 1971, p.15.

Cross referencesCircard series 7. card 1.



Bridge output amplifiersTypical dataICI, IC,: 741R,: 10kQ potR,: 10kQSupplies: f 15VRI,: 2kS2Output voltage: 15Vr.m.s. into 2kSZ (173’r.m.s. o/c) fork = 0.1 to1.0 at 1kHz.

Circuit descriptionMost power amplifiers have a single-ended output, deliveringto the load a voltage whose peak-to-peak value is at mostequal to the total supply voltage. If transformers/inductors areallowed such single-ended stages may produce peak-to-peakoutput voltage swings of up to double the supply voltage, butonly if the transistor breakdown voltages are equally high. Theeconomic and performance limitations imposed by trans-formers point to the need for an alternative output configura-tion for increased output voltage swing. If the load is takenbetween the outputs of two amplifiers delivering invertedoutputs of equal magnitude, then the load voltage being thedifference between the two has twice the magnitude of eachseparately. The method is illustrated using standard opera-tional amplifiers, but is applicable to amplifiers at all power

(1) Max output including increased drive at h.f.(2) Max output for distortion < 5%.

0 +3 26 f6 +12 *ViSUPPLY VOLTAGE (V)

levels, where the constraint of a grounded load needs to be met.This particular configuration offers the advantage that a singlepotentiometer controls the gain of both channels. The exactbalance is adjusted if required by setting Rz = RI. Equalmagnitudes of output are ensured for this condition assumingideal amplifiers because the two resistors carry equal currentwhile their junction is a virtual earth point. A further advan-tage of this circuit is the high input impedance. As only oneamplifier has a common-mode signal, the amplitude responsediffers somewhat, but the difference is only significant at thosefrequencies where the characteristic of each amplifier hasdeparted significantly from the ideal. Slew-rate limiting, anoutput circuit phenomenon, determines the highest frequencyat which large output voltages are obtainable with lowdistortion.

Circuit modificationsl Using two separate inverting amplifiers, with second setfor a gain of - 1, control over both outputs is obtained byvarying the gain of the first. As both are used as virtual-earthstages feed-forward compensation may be used to obtainstable performance with considerable increase in slew-rateand cut-off frequency.l Current capability of the output stages can be increased byany of the ways suggested on the cards describing class B/class A amplifiers. The simplest addition is a pair of comple-mentary emitter-follower combinations. Output currentcapability may be increased by one or two orders of magnitude,but the output voltage swing is slightly reduced because of thebase-emitter p.d. of the transistors. Crossover distortion maybe minimized by the addition of diode/transistor biasingnetworks to the transistor base circuits.l An alternative to the bridge circuit for increased voltageswing is the principle of supply bootstrapping of which this isone version.l Replace amplifiers by any compensated type (307, etc.);alternatively use uncompensated types (748, 301, etc.) withappropriate compensation capacitor (reduced compensationpossible with increased gain leading to higher slew rate).

l Resistor values non-critical but RI = Rz gives push-pulloutput (circuit usable as phase-splitter for succeeding stages).Resistor Rz may be made adjustable to take up tolerances ifoutputs are required to be given ratio, leaving tapping pointon potentiometer to vary total gain. Typical values forR1, Rz; 1kQ to 250kR. Higher values lead to offset, drift andadditional h.f. limitations; lower values absorb too much ofthe available output current.l If unity gain is sufficient, IC1 may be replaced by voltagefollower, R1 replaced by fixed resistor.

Further readingGreiter, O., Transistor amplifier output stages, part 1, bridgecircuits, Wireless World, vol. 69, 1963, pp.17-20.Del Corso, D. & Giordana, M., Simple circuit to double theoutput-voltage swing of an operational amplifier withincreased slew rate, Electronics Letters, vol. 8, pp.151/2.Ayer, J., Proportional d.c. motor control requires low-levelinputs, 400 Ideas for Design, Hayden 1971, p.225.Marshall, A. D., Differential input and output with op-amps,Wireless World, Jan. 1973, p.31.



Class B quasi-complementary outputTypical performanceSupply : +2ovTrl: BFR81; TrZ, Tr3:TIP3055Tr,, Tr,, Tr,: BFR41R1, R,: lSkJ2; R,: 1kSLR,: 47052; R,: 33052R,: 1.8kSZ; R,: 8.2kGR,: lkS2; R,: 852C,: 1OOpF; C,: 22,~F;C,: 10pFMain d.c. output: 1OVInput signal: 2.6V pk-pk

Circuit description .This is a circuit of a class B push-pull amplifier in whichtransistors Tr, and Tr, complement the pair Tr, and Tr,. Touse n-p-n transistors in the output stage for economy, theconfigurations of the two sections are different, i.e. Tr, and Tr,are connected as a Darlington pair and Tr, and Tr, as acomplementary pair. They receive essentially the same a.c.drive. but with the bases separated by Tr,. Tr, and Tr, conduct

Output signal: 6.7Vpk-pkOutput power: 5.4 wattsHarmonic distortion:5.8%Quiescent current :0.41AGraphs of harmonic dis-tortion and efficiencyversus output power for :!loads of 1552 and 8Q g “,“,shown opposite. ; 0 1

.“TP”: POW,‘, (W)4 5

for positive-going output signals and Tre supplies base currentdrive to Tr, and Tr, for negative-going output signals.Transistor Tr, is used in the so-called amplified diodeconfigura-tion in which the potential difference between the bases of Trland Tr, is set as a multiple of the Vbe of Tr, by the potentialdivider R,, R,, i.e. R, can be adjusted to give the desiredquiescent current in transistors Tr2 and Tr,. A forward biasis available which may allow the transistors to conduct to asmall extent, just sufficient to minimize the crossover distor-tion that can never be entirely absent. Transistor Tr, is aninverting amplifier with overall negative feedback through R,,the values of RB and R, determining the dc. output potential

in conjunction with R,. Because R5 is decoupled, tk a.c.properties of the arrangement are determined by the ratio ofR7 to the source resistance. Resistors RI and Rz are centre-tapped and this point is taken to the output via Cr, whichbootstraps R, so that the current through it remains constantthroughout the cycle of output voltage swing.

(i) (ii)

between input and output circuits being twice the Vbe of asingle transistor, (ii) uses complementary Darlington pairswith only one base - emitter path between input and output.Each pair comprises two inverting stages with 100% series -applied negative feedback giving unit gain.

Component changesAdjustment of R, to avoid just visible crossover distortiongives a quiescent current of 7mA.

Circuit modifications Further readingl To avoid dangerous overcurrent in either of the outputstage transistors, the current may be limited by adding series

New uses for the LMlOO regulator, National Semiconductor

resistors Re between the emitters and the output terminal (left).application note AN%7.

l Middle circuit shows an alternative arrangement, addingGrebene, B., Analog integrated circuit design, Van Nostrand

transistors Tr, and Tr,. These are normally non-conducting1972, pp.163-7.

except under overload conditions, i.e. as the output currentAmplifier efficiency (Letters), Wireless World, vol. 75, 1969,

increases the voltage drop across Re, or Re, causes Tr, or Tr,p.381.

to turn on and divert the base current available to Tr, or Tr,,Hartz, R. S. & Kamp, F. S., Power output and dissipation in

limiting the output current to I/be/&.class B transistor amplifiers, RCA publication AN-3576. (Also

l Alternative configurations for the output stages are shownin publication SSD-204A, p.594.)

right (i) requires low and high power n-p-n and p-n-p tran- Cross referencessistors to make up the Darlington pairs, the minimum p.d. Series 7, cards 1, 2 & 3.



Broadband amplifier

Typical performanceSupply : +2OV, 118mATr,: BFR41; Tr, : BFYSOR,: 3352; R,: 15052R,: 22052; R,: 22kQR,: 12052; RL: 5052(carbon)L,: 1.7pH; L,: 220pHPower gain x 14dB3dB bandwidth 64kHz to16mHz

Circuit descriptionIn many applications the transfer of power to a load at maxi-mum efficiency is not the primary consideration. Often, powergain is required for small input signals over a wide frequencyrange without introducing significant intermodulation andharmonic distortion. The common-base stage offers the bestlinearity of voltage gain against collector current, the latterchanging in sympathy with this input signal. The emitterfollower, while not providing voltage gain, gives a currentgain of the same order as a common-emitter stage and istherefore very useful for transferring power to a load, Toobtain this transfer with little distortion, it is necessary tooperate the emitter follower at a relatively high quiescent

current even for quite small input signals. The circuit uses acommon-base stage feeding the load via an emitter follower.To maximise the gain-bandwidth product, Tr, and Tr,operate in regions where their current gain is much smallerthan the normal values and they therefore have relativelyhigh quiescent currents. The input resistance of the commonbase stage is inverse to its quiescent current, so that a highcurrent allows the amplifiers input resistance to be matchedto that of the source by a suitable choice of R1. Resistor R,is determined from the required voltage gain (Av) for equalsource and load resistances R, cz Avhpe2RL/(Av + hp&.Inductor L1 is included to offset the capacitive loading due toTrZ and strays to maintain the gain at high frequencies. Todeliver as much output current to RL as possible at highfrequencies choke L, is included in series with R,.

Component changesWith Vcc (min) = +5V, vi, (max) x 14OmV r.m.s., supplycurrent is 30mA, and Pout w 1lmW.Tr, and Tr, can both be BFYSO or BFR41.Tr, can carry a much smaller quiescent current, using forexample an ME4103, with increased values of R2, Rs and R,.RI can be increased or decreased to allow matching to sourceresistances greater or less than 508 respectively.

Circuit modificationsIf the input signals are very small, output powers of aroundhalf a watt can still be obtained over a wide bandwidth bycascading a pair of amplifiers on the type described. When thegain-bandwidth product of the amplifier is not the mostcritical requirement and a higher efficiency is needed, thequiescent current in Tr, may be drastically reduced. ResistorsR, and R, would then need to be increased, with a correspond-ing increase in R1, if this is to be the means of controlling thequiescent operating conditions. The lower Tr, current may bechosen to make the natural input resistance of the stage, in theabsence of RI, the value required to match the source.

0 1973 IPC Bur;iness Press Ltd.

l Input resistance may be defined using shunt-applied feed-back, as shown left, where the emitter of TrI is d.c. or a.c.grounded, the feedback is not decoupled and the voltage gainis determined by the ratio RAIRR. The input resistance islargely that of RA except at high frequencies where the feed-back falls and the impedance at TrI base must be considered.l Inclusion of R,, as shown right, may be applied to both theprevious circuits to allow an output to be taken from thecollector of Tr,. To maximize the signal swing in the collectorcircuit of TrZ the bias network must be readjusted to leave asmall voltage at Tr, emitter, say by reducing R1 and R, in theoriginal circuit. The output resistance is approximately Rg;this stage is therefore convenient for feeding directly into anyother low impedance stage, such as that left, with RA removed.This mismatch can often be of advantage in extending thebandwidth of the amplifier.

Further readingHirst, R., Wideband linear amplifier, Wireless World, vol. 75,1969, pp.1 68-70.Meindl, J. D. & Hudson, P. H., Low-power linear circuits,IEEE Journal of Solid-State Circuits, vol. 1, 1966, pp.lOO-11.Lo, A. W. (and others), Transistor for Electronics, Chapter 9,Prentice-Hall, 1955.Griffiths, H. N., Simple wideband amplifier, Wireless World,vol. 75, 1969, p.478.

Cross referencesSeries 7, cards 1, 4, 5 & 10.


Class A op-amp power booster

Typical performanceTrl: BFR81Tr,, Tr,: TIP3055R,: 352; R,: 2508 pot.R,: 22052; Rll, R,: 1OldlC1 : 2,OOO~F; C, : 470pFC,: 10pFSupply voltage: 12VQuiescent current :1.25A (set by R,)

Circuit descriptionAvailable operational amplifiers have limited output currents,but may have a voltage swing approaching supply values. Thecircuit shown is class A buffer amplifier of unity voltage gainwhich may be added to such amplifiers to increase theiroutput current to 1A or more. In addition the circuit is a verysimple version of the voltage follower, having a low d.c.offset between input and output, a voltage gain very close tounity and a high input impedance. With the bootstrap tech-nique applied the amplifier is capable of driving low loadresistances to within < 1V of each supply line.

If a constant current flows in Rz, then as the base potential ofTrl increases the emitter current of Tr, decreases and with itthe collector current.

Output power for 1%t.h.d. into 352 load: 4.2W(supply current falls to1.05A at full output).Output voltage swing towithin about 0.7V ofsupply lines for 30 load Eand about 0.15V for 1552 ; dBload. : o

E$ 1:F5 EE!

z:sB}?&*lo? 1.5w ,nt0 3rl

This fall is fed to the base of Tr, causing it to conduct less,while the fall in emitter current releases more of the constantcurrent in R, to flow in the base of Tr,. Provided the currentgain of Tr, is reasonably high, the magnitudes of the basecurrent charges in Tr,, Tr, are equal but the signs are opposite.This represents an approach to ideal current phase-splitting.The constant current in R, is provided by the bootstrapcapacitor Cz, such that any change in the potential at the baseof Tr, is coupled via the follower action to the positive end ofRz, i.e. with no resulting change of p.d. across Rz in the idealcase. Resistors R, or R, require to be variable to set the

-

1

output current and stability of that current then depends onhFE variation in Tr,, Tr,. The base-emitter p.ds of Tr,, Tr,substantially cancel, as they can readily be chosen for junctionarea ratios matching the quiescent current ratios. As a class Aamplifier, maximum theoretical efficiency is 50%. At fulloutput the load power may approach 40 % of supply power inpractice, but the quiescent power is somewhat higher than thesupply power at full load.

Circuit modificationsl The good d.c. offset characteristics allow the amplifier tobe used as a voltage follower with d-c. coupling to the load.Bootstrapping should be retained unless the amplituderesponse is required to extend to d.c., as it swings the junctionof Rz, R, above the supply on positive signal swings. Hence itcan drive Tr, base far enough positive to saturate Tr, hardmaking maximum use of available supply voltage. If the loadis to be a.c. coupled but may carry a small quiescent current,the load resistance R1 may replace R,.l Any other constant-current circuit may replace the boot-strap arrangement, e.g. a f.e.t. either with gate strapped to

source as shown or with a resistor in the source lead to definesome lower value of current.l Although the distortion of the buffer stage above is low,the addition of a high voltage gain amplifier such as an op-ampcan increase the voltage gain to (RB/RA) + 1 while providingsufficient overall feedback to make distortion very low. Thewide bandwidth of the buffer stage together with its unity gainminimizes the risk of instability at high frequencies. Shouldthis be troublesome an op-amp with external compensationmay be used with increased compensation capacitor.

Further readingBelcher, D. K., Inexpensive circuit boosts op-amp outputcurrent, 400 Ideas for Design, vol. 2, Hayden, pp.l-2.Bloodworth, G. G., D.C. amplifier with unity voltage gain,Electronic Engineering, 1965, pp.1 12-4.Electronic Circuit Design Handbook, Current boosters fori.c. op-amps, Tab, 1971, p.161.

Cross referencesSeries 7, cards 2, 4 8c 8.



D.C. power amplifierTypical performanceICI,: LM305 or LMlOOTrI: MJE271Supplies: f15VR1, R,: 15052; R,: 15kQR,: 1kQ; R,: 1.8kQR,: 10kQC,: 1pF (tantalum);C,: 10,~F

Circuit descriptionThis circuit uses a voltage regulator, i.e. package to supply anoutput stage Tr, where the amplifier is to be used primarilywith a unipolar signal, though it can also be interpreted as aclass A output stage which can be a.c. coupled to a load. Thei.c. regulator contains its own reference voltage and a separatefeedback point (terminal 6) which allows the potential at thecollector of Tr, to be set to some stable value which is amultiple of the internal reference voltage, that multiple beingset by R,, R4 and R, the quiescent current in Tr, is then set bythe bias resistor R2 in conjunction with this predeterminedvoltage.

Input signal: 2.8V r.m.s.at 1OOHz

2 8

Maximum output volt- : 6age before symmetrical 2clipping 6.2V r.m.s.

w 4

Output power: 250mW$9 2

Harmonic distortion: :0.35 % at lkHz, and s yoO Ik 10k0.32% at 20kHz. FREOUEN‘Y, t (Hz)

An a.c. signal can be superimposed at pin 6 via C, and R,the circuit then behaving as a see-saw amplifier, as the referencevoltage leaves the feedback terminal 6 as an a.c. virtual earth.For d.c. purposes, the circuit may be treated as series appliedfeedback. The peak current in the load is limited to a fractionof the quiescent current for negative excursions; as the voltagegoes negative the p.d. across Rz falls and with it the currentthrough Rz. The current in Rr for this voltage exclusion cannever exceed Rz even when the transistor current falls to zeroin the positive direction; however, much greater currents canbe provided through Tr,. The amplifier is thus an inefficientclass A amplifier whose effectiveness can be improved byreplacing Rz by a constant-current stage which can sustain agiven peak current in R1 almost equal to the quiescent value,even for large voltage excursions in the negative direction.Capacitor C1 is used to suppress h.f. oscillation and a lowinductance type must be used.

1--1%b

Onset of slew-rate limitation occurs at 70kHz for an outputsignal level of 16V pk-pk when the signal level is reduced to3 to SV pk-pk by reducing the input signal. Voltage gain isflat up to lOOkHz, with 3dB fall-off occurring about 250kHz.

Circuit modificationsl Resistor R1 is replaced by the Baxandall constant-currentcircuit shown left Trr: BFR81, Trz: TIP3055 R,: 1852, R,:3.9kQ. This permits a much greater input signal level beforepeak clipping occurs. Resistor R, is chosen for approximately alOOmA constant quiescent current in the path a-b (about 1.Wis available at terminal 6). If the transistor Tr, output currentis 200mA pk-pk, then the output current swing in load Ri istwice that for the case when Rz is 15052 with the same quiescentcurrent. A comparison of instantaneous currents for the twopossible circuits between a and b is tabled above.

l The regulator may be replaced by the operational ampli-fier emitter-follower circuit, shown right. To maintain the d.c.stability of the output, the non-inverting terminal must beconnected to a suitable stable reference voltage. If the d.c.power supply is stabilized, then this may be a tapping on apotential divider connected across the supply. For minimumdrift, the effective resistance seen at both input terminals of theop-amp should be comparable.

Further readingNew uses for the LMlOO regulator, National Semiconductorapplication note AN-8, 1968.Amplifier efficiency (Letter), Wireless World, vol. 75, 1969,p.535.

Cross referencesSeries 3, card 8.Series 7, card 12.



Class D switching amplifier+” Typical performance

Tr;: BFR81Tr,: BFR41D,, D,: SP2Supply: flOV(4 to 15V)R1, R,: 18OQ, R,: 1OkQR, : 10052, R6: 4.7kQR,: 47052; R,, R,: 1kQC,, Cz, C,: 1OOnFZL: 1mH (r = 0.952)+1552

C3

Circuit descriptionBasically, the circuit is an astable oscillator, generating asquarewave that is used to drive a complementary pair ofoutput transistors into conduction on alternate half-cycles ofthe squarewave. The output transistors thus switch the voltageto the load at a frequency that is much higher than that of thesignals to be amplified. The squarewave generator is designedaround the operational amplifier At which uses positive feed-back via R, and R,. The periodic time of the squarewave fedto RB depends on the time constant R5Cl if R, is much greaterthan Rg. To obtain a realistic switching frequency with

Vbias : -640mV to setmean load voltage tozero with fin = 0.Switching frequency:27.8kHz, max 40 kHz.With fin = 0, supplycurrent is f20mA; withVi,, = 3.4V pk-pk1OOHz; current isx 130mA; power in 15-Q

load x 1.66W; residual“carrier” x 300mVacross 15Q; overallefficiency 64%; outputstage efficiency % 76 %;3-dB bandwidth~60OHz. Withrectangular input atlOOHz, output rise andfall times % 600,~s.

reasonable components and also to obviate the need for largeinput signals a compromise must be made in the value of R,.Current in R, flows alternately in R, and R, producing p.dsacross these resistors that are sufficient to switch on Tr, andTr, respectively. The signal applied to R, causes the mark-to-space ratio of the output waveform from the astable to varyin sympathy with the instantaneous value of Vi,,, so that themean value of the voltage applied to the load also variesdirectly with the input signal. If the load impedance has anexternal filter, or is by its nature self-filtering such as with amotor, then the power drawn from the amplifier at theswitching frequency is low and the useful signal power in theload will be high.

If the mark-to-space ratio of the squarewave generated bythe astable is not unity with I/in = 0, it can be made so by asuitable choice of the bias supply and Rs. Diodes D1 and Dzprotect Tr, and Tr, against breakdown when the load im-pedance is highly inductive.

Circuit modificationsl The bias source to set the mark-to-space ratio of thesquarewave to zero can be obtained by a potentiometerconnected between ground and the appropriate supply line.l While an inductor is normally used in series with a resistiveload to filter out the h.f. sauarewave. anv suitable low-nassfilter can in principle be connected between the junction ofTr, and Tr, collectors and R1. Another possible method is toconnect a capacitor in parallel with the inductive smoothingchoke so that it is resonant at the switching frequency. Forexamnle. with a choke of 1mH and r = 152. a parallel canaci-

V?.fcomplementary pulse-width-modulated switching waveforms.The bridge of power transistors is connected across a single-ended supply. Component details are given in the firstreference.

tance-of 16nF would be resonant at switching frequency of4OkHz. At signal frequencies less than about 5OOHz, the

Further reading

impedance of this tuned network is inductive, having a maxi-National Semiconductor, data sheet and application notes

mum impedance of about 352.on the LM3 11 voltage comparator, 1970.

The complementary pair of transistors forming the outputCamenzind, H. R., Modulated pulse audio and servo power

l

stage can be replaced by a bridge-type network as shown left.amplifiers, International Solid-State Circuits Conference,

The four transistors are fed with complementary pulse-width-University of Pennsylvania, Philadelphia, 1966, pp.90/1.

modulated squarewaves which cause the transistors to beMeidl, J. D., Micropower circuits, Wiley, 1969, pp.61,64 & 65.

switched on and oflin pairs. With Tr, and Tr, on current flowsGarza, P. P., Getting power and gain out of the 741 type

in the load in one direction and is reversed when Tr, and Tr,op-amp, Electronics, 1 Feb., 1973, p.99.

are switched on. Cross referencesl Another practical form of bridge output stage is shown Series 7, cards 1 & 2. Series 2, card 4. Series 3, card 1.right using a pair of voltage comparators to generate the Series 4, card 8.


wireless world circard

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