3106 ieee transactions on microwave theory and …

3106 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 10, OCTOBER 2005

Compact and Broad-Band Millimeter-WaveMonolithic Transformer Balanced Mixers

Pei-Si Wu, Chi-Hsueh Wang, Tian-Wei Huang, Senior Member, IEEE, and Huei Wang, Senior Member, IEEE

Abstract—Three broad-band miniature monolithic transformersingly balanced diode mixers for operation in the microwave andmillimeter-wave bands are reported in this paper. The coupled-line equivalent models are used to synthesize the initial design ofthese transformers up to 50 GHz. The first one is a broad-bandspiral transformer mixer, and the second one is a 21-GHzMarchand-type transformer mixer. These two mixers with chipsizes around 0.29 mm2 exhibit bandwidths of 105% and 54.5%,respectively. We also propose a 30-GHz single-coiled transformermixer, which has comparable performance with the first twomixers and reduced chip size. The single-coiled transformer mixerachieves a bandwidth of 100% with the chip size smaller than0.25 mm2. In order to save chip area, all these transformers pro-vide broad-band matching to the diodes directly. To the authors’knowledge, these mixers achieve the widest bandwidths withthe smallest chip sizes among all passive balanced mixers usingmonolithic-microwave integrated-circuit processes in dc–40-GHzfrequency range.

Index Terms—Diode, mixer, monolithic microwave integratedcircuit (MMIC), transformer.

I. INTRODUCTION

ALTHOUGH microwave mixer design is well developed, itstill remains a challenge to develop high-performance and

low-cost monolithic mixers [1], [2]. For low-frequency appli-cations, Gilbert-cell mixers have good performance and smallchip area. Above 10 GHz, Gilbert-cell mixers also have attrac-tive function, but they need an extra matching circuit to enhancetheir bandwidth [3]–[5]. Gilbert-cell mixers are fully differentialcircuits, but may require single-ended to balanced transforma-tion for connection with other circuit components. Passive bal-anced mixers, which use Lange couplers [6], Marchand baluns[7], or rat-race baluns [8], are widely used in microwave andmillimeter-wave frequencies. The balun sizes of these passivebalanced mixers are proportional to the wavelength and oftenoccupy most of the chip area. Alternatively, the transformermixer with meandered coupled lines demonstrates a compactdesign in the - and -band [9], [10]. In [9] and [10], thesizes of the transformer mixers are reduced significantly (4–13times) compared to the mixers using the above-mentioned pas-sive baluns.

Manuscript received January 13, 2005; revised April 7, 2005. This workwas supported in part by the National Science Council under Grant NSC93-2752-E-002-002-PAE, Grant NSC 93-2213-E-002-033, Grant NSC93-2219-E-002-024, Grant NSC 93-2752-E-002-003-PAE, and Grant NSC93-2219-E-002-025.

The authors are with the Department of Electrical Engineering and GraduateInstitute of Communication Engineering, National Taiwan University, Taipei106, Taiwan, R.O.C. (e-mail: [email protected]).

Digital Object Identifier 10.1109/TMTT.2005.855122

Fig. 1. Mixer structure of: (a) conventional singly balanced mixer and(b) transformer mixer.

In this paper, we discuss three singly balanced transformermixers using GaAs-based monolithic-microwave integrated-circuit (MMIC) technology. The first one is the broad-bandspiral transformer mixer. The second is the 21-GHz Marc-hand-type one [11]–[13], which demonstrates the first -bandMarchand-type transformer. We further proposed a 30-GHzsingle-coiled transformer mixer. The single-coiled transformeris modified from an -band CMOS design in [14] and [15], andextended to -band in this study. To minimize the chip sizes,all these three transformers provide broad-band matching tothe diodes directly. Among these three mixers, the broad-bandspiral transformer mixer has the widest bandwidth, the 21-GHzMarchand-type mixer obtains the best isolation, and the 30-GHzsingle-coiled mixer achieves comparable performance and thesmallest chip size. To the authors’ knowledge, these mixersachieve the widest bandwidths with the smallest chip sizesamong all passive balanced mixers using MMIC processes.

II. DESIGN METHODOLOGY

The conventional singly balanced mixer uses a four-port hy-brid, such as a 180 rat-race balun or a 90 Lange coupler [16],as shown in Fig. 1(a). In this mixer, RF has the same phaseand the local oscillator (LO) has 180 phase difference to thediodes. A low-pass filter is often utilized to filter out the IFsignal. Using this mixer type, LO-to-RF isolation is determinedby the port-to-port isolation of the four-port hybrid. An alterna-tive configuration, which is shown in Fig. 1(b), is utilized in thispaper. Since the transformer is a three-port element, only the LOfeeds through it and the signal has 180 phase difference to thediodes. RF feeds through a high-pass filter and IF is taken outthrough a low-pass filter. The LO-to-RF and LO-to-IF isolationare directly related to the magnitude and phase balance of thetransformer.

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The circuits are designed using a 0.15- m GaAs pseudo-morphic high electron-mobility transistor (pHEMT) MMICprocess provided by WIN Semiconductors, Taoyuan, Taiwan,R.O.C.1 In order to design a miniature transformer mixer, adiode size should be selected for the best conversion loss atmatched impedance. To save the chip area of matching circuits,the transformer balun can be designed to provide broad-bandmatching to the diodes directly instead of matched for 50 . Forminimum size consideration, only a single and small capacitoris used for the RF high-pass filter, and a series inductor and ashunt capacitor are used for the IF low-pass filter.

Three different types of transformers are utilized to imple-ment the mixers. For these three types of transformers, the mul-ticoupled-line model in EDA software such as ADS and Mi-crowave Office are utilized to synthesize the transformers inthe initial designs. A full-wave electromagnetic (EM) simulator(Sonnet Software, Liverpool, NY) [17] is used after the initialdesign to predict the performance more precisely.

A. Conventional Transformer

The first type of transformer is the conventional transformerusing two oppositely wrapped twin coils connected in series. Asimplified circuit diagram of it is shown in Fig. 2(a). Port 1 isconnected through two coils to ground; ports 2 and 3 are con-nected from ground to coil. At the center frequency, an exci-tation signal at port 1 will couple to ports 2 and 3 with equalmagnitude and phase difference of 180 .

This transformer is divided into two coils; each coil can bemodeled utilizing multicoupled-line elements. Fig. 2(b) showsone coil layout structure of the transformer. The line length fromport a to port c (gray line) is the same and symmetric as the linefrom port b to port d (black line). The gray and black lines aremetals on the same layer. The total line length of one coil(including the gray and black lines) is

(1)

where is the inner square width in Fig. 2(b), is the numberof coupled lines on each side, and and are, respectively,the linewidth and line gap. Each linewidth is equal and, thus, sois each line spacing. The area of one coil is

(2)

By neglecting the corner effect, one coil can be modeled usingfour -coupled lines [18], as shown in Fig. 2(c). Instead ofmodeling coupled lines i and iii individually, one eight-cou-pled microstrip line can be used for the coupling considera-tion. Similarly, coupled lines ii and iv are modeled by anotherset of eight-coupled microstrip lines. In order to use a simplereight-coupled-line model, as shown in Fig. 2(d), coupled line ivis interchanged with coupled line iii. It turns out that the sim-ulation results for Fig. 2(c) and (d) are nearly identical, when

is the same or larger than the line gap and the total linelength is less than twice the center frequency wavelength, which

1WIN 0.15 �m Power (10 V) pHEMT Design Kit (rev. 0.3.1), 2003.

Fig. 2. (a) Simplified circuit diagram. (b) One coil layout structure of theconventional transformer. (c) Equivalent multicoupled line. (d) Simplified2n-coupled-line model (n = 4) of one coil.

is true for most cases. Therefore, one coil can be modeled uti-lizing one -coupled microstrip lines, as shown in Fig. 2(d).The line length of each microstrip line in Fig. 2(d) is the av-erage of the total line length of one coil, and it can be calculatedby

(3)

In Fig. 2(c), the current flow of coupled line i (ii) is in thereverse direction of coupled line iii (iv) so the insertion loss



Fig. 3. Insertion losses of the conventional transformer as a function of theratio of the inner square width to the substrate height ID=H .

Fig. 4. Chip photograph of the conventional transformer. The chip size is0.5� 0.48 mm .

and magnitude imbalance will both deteriorate as decreases.Fig. 3 shows insertion losses of the conventional transformer asa function of the ratio of the inner square width to the substrateheight . In order to further decrease the coupling effect,we place a ground at the center of the coils. If via-holes areplaced in the coil center directly, the imbalance can be improved.For size consideration, small squares connected to via-holes areutilized in this case. The chip photograph of this transformer isshown in Fig. 4. The chip size is 0.5 0.48 mm .

The design procedure of this transformer is summarized asfollows.

1) Calculate of the designed center frequency.2) To minimize the chip area, the linewidths and line gaps

are selected to the design rule limit of 5 m.3) Determine the inner square width . For both size and

coupling consideration, 50 m is selected.4) Let , then the number of turns of one coil

can be calculated by (3).5) Use a multicoupled microstrip line model to synthesize

the initial design of this transformer.6) Finally, the full-wave EM simulator is utilized to deter-

mine the final layout.

The transformer was tested via on-wafer probing. We usedan HP8510C network analyzer to measure the small-signal dataup to 50 GHz. The three-port -parameters are extracted fromthe two-port measurements using a port-reduction method [19],[20]. However, since the transformer is designed to match to thediodes instead of 50 , the simulated and measured -param-eters under the 50- system do not exactly reflect the perfor-mance of the transformer in the mixer circuit. Fig. 5(a) showsthe measured total power lossesof this transformer, it is below 9 dB from 5 to 30 GHz. Thesimulated and measured insertion losses and phase differencesbetween ports 2 and 3 of the transformer are shown in Fig. 5.The multicoupled-line model can be used to predict the initialvalues of line gaps, widths, and lengths. After that, we can startfrom these initial values for the optimization of the transformerperformance. Neglecting the corner effect of the coils makes themulticoupled-line model results slightly different from EM sim-ulations. The magnitude difference between EM simulation andmeasured results are less than 2 dB up to 30 GHz.

B. Marchand-Type Transformer

The second type of transformer is the Marchand-type trans-former [11], [12]. Unlike the design in [11] and [12], thelinewidths and line gaps of this transformer are adapted to thedesign rule limit of 5 m to minimize the chip area, therefore,this transformer operates in the higher frequency and occupiesa much smaller chip area. A simplified circuit diagram of theMarchand-type transformer is shown in Fig. 6. This configu-ration is very similar to the conventional transformer, but withport 1 connected through two coils and then to an open circuitinstead of to ground. This transformer can be considered as aMarchand balun with each coil to be a quadrature coupler.

For the mixer to match the diodes, the total line length ofone coil is approximately of the center frequency. The innersquare width of the coils is still a concern for magnitude balanceand insertion loss. In order to obtain good insertion loss and littleimbalance of and , via holes are placed in the centerof the coils to improve them, as mentioned before in the con-ventional transformer design. The chip photograph of this trans-former is shown in Fig. 7. The chip size is 0.72 0.42 mm .

The simulated and measured results are shown in Fig. 8. Theimbalance of this transformer is very small (from 10 to 25 GHz).Fig. 8(a) also shows the measured total power losses of thistransformer (it is below 7 dB from 10 to 30 GHz). The cou-pled-line model can be also used as the initial design, and theagreement between EM simulations and measured results is lessthan 2 dB up to 30 GHz.

C. Single-Coiled Transformer

The third type of transformer is the single-coiled transformer.This transformer is modified from an -band CMOS design[14], [15]. Due to the restriction of the air-bridge design rule,we modified the structure to the GaAs process and extended it tothe -band. A simplified circuit diagram of the single-coiledtransformer is shown in Fig. 9(a). Its simplified circuit diagramis similar to the conventional one; the difference is that all thelines are intertwined to one coil instead of two coils. By thissingle-coiled structure, the size of this transformer is more com-pact than the other two transformers. The structure of this trans-



Fig. 5. Simulated and measured results of: (a) jS j, (b) jS j, and(c) amplitude and phase difference of S and S of the conventionaltransformer. Measured result of total power loss (1�jS j �jS j �jS j )is also shown in (a).

former is shown in Fig. 9(b). The gray metal, the metal withpolka dots, and the metal with oblique stripes are all metals onthe same layer. The difference in pattern between metals is only

Fig. 6. Simplified circuit diagram of Marchand-type transformer.

Fig. 7. Chip photograph of the Marchand-type transformer. The chip size is0.72� 0.42 mm .

for the clarity. The total line length in the -direction ( )and in the -direction ( ) are

(4)

(5)

where and are the inner square widths in the - and-directions in Fig. 9(b), respectively; is the number of turns

parallel to the -axis, which is an even number. and are,respectively, the linewidth and line gap. The line lengths fromports 2 and 3 to ground are the same, but they are not a halflength of port 1 to ground as the other two transformers. Thearea of this transformer is

(6)

The total line number of upper and lower sides, which is dif-ferent from the number of left and right sides , is . In-stead of using one section coupled line to model the transformer,one -coupled line and one -coupled microstrip lineare used to synthesize this transformer. The line length of the

- and -coupled microstrip line areand , respectively.

Similar to the previous two transformers, the inner squarewidth is also an important factor of this transformer. One via-hole is put in the center to decrease the coupling effect betweenthe opposing sides of the coil.

The design procedure of this transformer is similar to the con-ventional transformer; the only difference is that the number ofturns is decided by

(7)



Fig. 8. Simulated and measured results of: (a) jS j, (b) jS j, and(c) amplitude and phase difference of S and S of the Marchand-typetransformer. Measured result of total power loss (1�jS j �jS j �jS j )is also shown in (a).

which means that the line length from ports 2 or 3 to ground isequal to . The chip photograph of this transformer is shownin Fig. 10. The chip size is 0.58 0.45 mm .

Fig. 9. (a) Simplified circuit diagram. (b) Structure of the single-coiledtransformer.

Fig. 10. Chip photograph of the single-coiled transformer. The chip size is0.58� 0.45 mm .

The simulated and measured results are shown in Fig. 11. Theimbalance between ports 2 and 3 is similar to the conventionaltransformer. Fig. 11(a) also shows the measured total powerlosses of this transformer; it is below 8.7 dB from 5 to 35 GHz.The coupled-line model is still valid for the initial design, whilethe EM simulations are within 1.5-dB differences to the mea-sured results up to 25 GHz.

III. MIXER IMPLEMENTATIONS AND MEASUREMENT RESULTS

The three transformers are separately applied to three singlybalanced mixer designs. The diode sizes of these mixers areall two-finger 20- m devices. The diode is realized by con-necting the drain and source of a pHEMT as the cathode of theSchottky diode, while the gate metallization is realized as theanode. The cutoff frequency of the two-finger 20- m Schottkydiode is 301 GHz. The whole circuits are simulated by the cir-cuit simulator (HP/EEsof Libra). All these mixers are measuredvia on-wafer probing. We used an Agilent E8247C PSG as theLO source, and an HP83650B signal generator as the RF or IFsource.

A. Broad-Band Spiral Transformer Mixer

The chip photograph of the broad-band spiral transformermixer with the conventional transformer is shown in Fig. 12.The chip size is 0.63 0.46 mm . The simulated and measured



Fig. 11. Simulated and measured results of: (a) jS j, (b) jS j, and(c) amplitude and phase difference of S and S of the single-coiledtransformer. Measured result of total power loss (1�jS j �jS j �jS j )is also shown in (a).

results of this mixer are shown in Fig. 13. The conversion loss isbetter than 10 dB from 10 to 32 GHz. Between 19–23 GHz, theconversion loss of this mixer is better than 6 dB. The RF-to-IF

Fig. 12. Chip photograph of the broad-band spiral transformer mixer. The chipsize is 0.63� 0.46 mm .

Fig. 13. Simulated and measured: (a) conversion losses, RF-to-IF isolations,(b) LO-to-IF isolations, and LO-to-RF isolations of the broad-band spiraltransformer mixer for down conversion, of which LO power is 13 dBm and IFis fixed at 1 GHz.

isolation is not good at low frequency because only a series in-ductor and shunt capacitor are used for the IF low-pass filter.



Fig. 14. Chip photograph of the Marchand-type transformer mixer. The chipsize is 0.65� 0.45 mm .

The LO-to-RF isolation is good at low frequency, but degradesat high frequency. The degredation is due to the imbalance fromthe transformer, specifically the magnitude imbalance. Since theimbalance of this transformer at low frequency is very small,and the low-pass filter has good rejection at high frequency, theLO-to-IF isolation is better than 33 dB from 9 to 34 GHz.

B. Marchand-Type Transformer Mixer

The chip photograph of the Marchand-type transformer mixeris shown in Fig. 14. The chip size is 0.65 0.45 mm . Fig. 15shows the simulated and measured results of this mixer. It canbe observed that the conversion loss is better than 10 dB from12 to 21 GHz. Between 15–18 GHz, the conversion loss is betterthan 6 dB. Due to the same reason as the broad-band spiraltransformer mixer, the RF-to-IF isolation is not good at low fre-quency. The LO-to-RF and LO-to-IF isolations are better than30 dB from 12 to 21 GHz due to the good magnitude and phasebalance of the Marchand-type transformer.

C. Single-Coiled Transformer Mixer

The chip photograph of the single-coiled transformer mixeris shown in Fig. 16. The chip size is 0.58 0.42 mm . The sim-ulated and measured results of this mixer are shown in Fig. 17.It can be seen that the conversion loss is better than 10 dB from10 to 30 GHz. From 15 to 23 GHz, the conversion loss is betterthan 5 dB. The poor RF-to-IF isolation at low frequency is sim-ilar to the other two transformer mixers. The imbalance of thetransformer degrades the LO-to-RF isolation. Due to the charac-teristic of the single-coiled transformer, the LO-to-RF isolationis good at low frequency, but becomes worse at high frequency.The LO-to-IF isolation is better than 30 dB from 9 to 33 GHzbecause the imbalance of this transformer at low frequency isvery small, as shown in Fig. 11(c), and the low-pass filter hasgood rejection at high frequency.

In these three mixers, the LO-to-RF isolations are directlyaffected by the magnitude and phase imbalance of the trans-formers. The Marchand-type transformer has the lowest imbal-ance among these three transformers, therefore, its LO-to-RFisolation is also the best. The LO-to-IF isolation is not onlycaused by the imbalance of the transformer, but also by the re-jection of the low-pass filter at the IF port. All these mixers havebetter than 30-dB LO-to-IF isolations. The RF-to-IF isolationsare determined by the rejection level of the high-pass filter in

Fig. 15. Simulated and measured: (a) conversion losses, RF-to-IF isolations,(b) LO-to-IF isolations, and LO-to-RF isolations of the Marchand-typetransformer mixer for down conversion, of which LO power is 13 dBm and IFis fixed at 2 GHz.

Fig. 16. Chip photograph of the single-coiled transformer mixer. The chip sizeis 0.63� 0.46 mm .

the RF port and the low-pass filter in the IF port, therefore, theisolations of the mixers are about the same. The cause for thevariation of conversion losses is more complicated, and cannotbe determined directly from the transformers.



Fig. 17. Simulated and measured: (a) conversion losses, RF-to-IF isolations,(b) LO-to-IF isolations, and LO-to-RF isolations of the single-coiled trans-former mixer for down conversion, of which LO power is 13 dBm and IF isfixed at 1 GHz.

Among these three mixers, the broad-band spiral transformermixer has the widest bandwidth with an aspect ratio of greaterthan 1 : 3. The Marchand-type one has the best isolations, butthe bandwidth is much smaller than the others. The bandwidth islimited due to the impedance matching between the diodes andtransformer. The single-coiled transformer mixer has compa-rable performance compared to the other two with the smallestchip area.

IV. CONCLUSION

Three broad-band compact singly balanced transformermixers have been designed, fabricated, and tested. Themulticoupled-line equivalent models are proposed to synthesizethese transformer mixers in the initial designs. Conventionaland Marchand-type transformers have been used to realize themixers with bandwidths of 105% and 54.5%, and conversionlosses better than 10 dB from 10 to 32 and 12 to 21 GHz,respectively. The individual chip size of each chip is only

approximately 0.29 mm . The proposed single-coiled mixerhave a bandwidth of 100%, the conversion loss is better than 10dB between 10–30 GHz and below 5 dB from 15 to 23 GHz.Its chip size is very compact, only 0.25 mm .

ACKNOWLEDGMENT

The chip was fabricated by WIN Semiconductors, Taoyuan,Taiwan, R.O.C., through the Chip Implementation Centerof Taiwan, Taiwan, R.O.C. The authors would like to thankH.-Y. Chang, M.-F. Lei, and M.-C. Yeh, all with NationalTaiwan University, Taiwan, R.O.C., Prof. Y.-S. Lin, NationalCentral University, Taiwan, R.O.C., and J.-S. Fu, The Univer-sity of Michigan at Ann Arbor, for their helpful suggestions.

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Pei-Si Wu was born in Changhua, Taiwan, R.O.C.,in 1980. He received the B.S. degree in electric en-gineering from National Taiwan University, Taipei,Taiwan, R.O.C., in 2002, and is currently working to-ward the Ph.D. degree at the National Taiwan Univer-sity.

He is currently with the Graduate Institute ofCommunication Engineering, National TaiwanUniversity. His research interests include microwaveand millimeter-wave circuit designs.

Chi-Hsueh Wang was born in Kaohsiung, Taiwan,R.O.C. in 1976. He received the B.S. degree inelectrical engineering from National Cheng KungUniversity, Tainan, Taiwan, R.O.C., in 1997, andthe Ph.D. degree from National Taiwan University,Taipei, Taiwan, R.O.C. in 2003.

He is currently a Post-Doctoral Research Fellowwith the Graduate Institute of Communication Engi-neering, National Taiwan University. His research in-terests include the design and analysis of microwaveand millimeter-wave circuits and computational elec-

tromagnetics.

Tian-Wei Huang (S’91–M’98–SM’02) received theB.S. degree in electrical engineering from NationalCheng Kung University, Tainan, Taiwan, R.O.C., in1987, and the M.S. and Ph.D. degree in electrical en-gineering from the University of California at LosAngeles (UCLA), in 1990 and 1993, respectively.

In 1993, he joined the TRW RF Product Center,Redondo Beach, CA. From 1998 ti 1999, he waswith Lucent Technologies, where he was involvedwith local multipoint distribution system (LMDS)fixed wireless systems. From 1999 to 2002, he was

involved with RF/wireless system testing with Cisco Systems. In August 2002,he joined the faculty of the Department of Electrical Engineering, NationalTaiwan University. His research has focused on the design and testing ofmonolithic microwave integrated circuits (MMICs) and RF integrated circuits(RFICs). His current research interests are MMIC/RFIC design, packaging,and RF system-on-chip (SOC) integration.

Huei Wang (S’83–M’87–SM’95) was born inTainan, Taiwan, R.O.C., on March 9, 1958. He re-ceived the B.S. degree in electrical engineering fromNational Taiwan University, Taipei, Taiwan, R.O.C.,in 1980, and the M.S. and Ph.D. degrees in electricalengineering from Michigan State University, EastLansing, in 1984 and 1987, respectively.

During his graduate study, he was engaged inresearch on theoretical and numerical analysis ofEM radiation and scattering problems. He was alsoinvolved in the development of microwave remote

detecting/sensing systems. In 1987, he joined the Electronic Systems andTechnology Division, TRW Inc. He was a Member of the Technical Staffand Staff Engineer responsible for monolithic-microwave integrated-circuit(MMIC) modeling of computer-aided design (CAD) tools, MMIC testingevaluation, and design. He then became the Senior Section Manager of theMillimeter Wave Sensor Product Section, RF Product Center, TRW Inc. In1993, he visited the Institute of Electronics, National Chiao-Tung University,Hsin-Chu, Taiwan, R.O.C., and taught MMIC-related topics. In 1994, hereturned to TRW Inc. In February 1998, he joined the faculty of the Departmentof Electrical Engineering, National Taiwan University, as a Professor.

Dr. Wang is a member of Phi Kappa Phi and Tau Beta Pi.


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