a microwave fet power amplifier 6 ghz, 6 wdigitool.library.mcgill.ca/thesisfile62629.pdf · a...
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A MICROWAVE FET POWER AMPLIFIER as, a T.W.T. substitute, operating at 6 GHz, 6 W -
by:
. Luiz H.A. Duque, B. Eng (Fundaçao Valeparaibana de Ensino, /
S.J. Campos, S.P., Brazil)
A thesis submitted to the Faculty of Graduate Studies and Research in partial fulfillment of the
degree of Master of Engi neeri ng
Date January 1983
Department of Electri cal Engineering
McGill University Montreal, Que.
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~Theory,which 19 concerned with the immutable essence of things beyond the mutable region of human affairs, can obtain practical validity only by molding the manner of life of men engaged in theory"
, JÜRGEN HABERMAS
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ABSTRACT
Thi s thes i s desc ri bes the deve l opment of a 6 Watt FET power ampl ifier operated in the C band (5.95 -6'.45 GHz). Thi s five stage MIe
power amplifier provides a small 'signal gain of 46 dB, 18% power added
efficiency and over 500 MHz ydB bandwidth. .. . . .. -
The FET power amplifier features good efficiency, low noise figure, low AM to PM distortion and excellent linearity. During the
development of thi s ampl i fier, a comparati ve study between push-pull class AB and hybrid coupled cla'ss A amplifier was made. The hybrid -coupled class A configuration ',s used in the power stage of tne' amplifier due to the tight phase distortion requirements that could
not be met with the push-pull class AB configuration.
The characteristics ,of the amplifier indicate that it can be a suitable replacement for the travelling wave tube in future analog and
di gi tal communi cation systems.
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RESUME
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Cètte the-se décrit la conception et la réalisation dlun
amplificateur de puissance à Transistor à Effet d~ Champs'fonctionnant dans la bande C (5.95 -6'.45 GHz). Cet ,amplificateur, composé' de cinq étages IIMIen, fourni t, un gai n tçta1 pui ssance de s'ortie de 0 Watts avec larg~ur de bande 3_1 dB de 500 MHz.
petit si gnal de 46 dB, une "
une effi caci té de 18'1, et une
Les particularités de cet amplificatèur sont une bonne efficacité, un faible facteur' de bruit, peu de distorsion amplitude à phase et 'une
, e~c~llente linéarité. Pour la conception de cet amplificateur un~ étude comparative entre les ampl iff cateùrs 3 montage symêtri que de classe AB et les amplificateurs à couplage hybride de classe A 'a été effectuée, cette dernière configuration a été adoptée pour ,1 étage de
sortie parce que la spécification de la distorsion amplitude 3 phase
admissible ne pouvait être rencontrée avec un étage à montage symétri que de cl asse AB.
les résultats obtenus montrent qu'un ampliJicateur de ce type remplacerait adéquatemen~ un '-:ube à Ondes Progressives utilisé dans les radios analogiques et numiriques en hyperfréquences. C
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ACKNOWL EDGMENTS
1 would like ta thank a11 those to whom 1 am ,indebted during IllY thesis 'work at McGil1 University ~d Northern Telecom (Ana1og and Radio Transmission Division, St. L:u'rent, Quebec). In particular l want ta thank my research"directors, Dr. T. Pavlasek and Dr. P. Bura
for. 'their const~~t support 'durfng the execution of this work. 1 am
also very gratef~l to Mr. W. '~rzew1bcki. manager of the Radio design group (Northern Tel eçom) for hi s constant encoùragement and support. Speci al thanks are also due to Miss M. Horan who~ kindly typed the
manuscripts of this thesis. '
Finally, 1 would like ta thank these special friends, Lucie Girard»" Alix Glorieux. and Igor Rossine for, their patience and
encouragemen:t duri ng the wri, t1n9 of th1 s thesfs.· :.
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TABLE OF CONTENTS Page ' '
ABSTRACT i
RESUME ii
ACKNOWLEDGEMENTS' "
. TABLE OF CONTENTS
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LIST OF ILLUSTRATIONS vi;
/ , 1. INTRODUCTION 'J
2. FIELD EFFECT TRANSISTOR DEVELOPMENT 4 ~ 2.1 Microwave FET Structures 5
2.1.1 M~imization of gate width 12 \l>
2.1.2 Minimization of parasitics 13 )
2.1.3 ReductiOn of, thenmal impedance .. 14 .. f,'
2.1.4 FET structures and source-drain burnout 16
2.1.5 Fabricàtion technology 19 ~}
.1
2.2 FET re li abi li ty ) (
EFFltIENCY TWT's " 3. HIGH
20
24
3.1 NEC LD4353' TWT . 0\ 26 t!2~
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Page
3.2 Siemens RW890 26
1 3.3 Thomson - CSF TH 3600 27
/ \ . 3.4 J COl11llents 27
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4. 6 GHz, 6 WATT FET POWER AMPLIFIER f9
4.1 Specification 30
4.2 P.A. Line-up 32
4.3 Design Approach 36
4.3.1 Stabi,lity consideration 36
4.3.2 FITA-2 45
4.3.3 Matching networks 55
/ 4.3.3.1 L-Type matching netWork 55
4.3.3.2 [1-Type matchi n9 network 59
4.3.3.3 Transmission line transformers 60
4.3.3.4 Mfcrostrip steps 62
64 4.3.4 Comparative study of a push-pull class AB amplifier \.
and of a hybrid coupled class A power amplifier
4.3.4.1 Biasing cons-iderations 65 "
4.3.4.2 Circuit analysis 70
-,~, 4.3.4.3 .Amp 1 i f_1 er pe rformance 75
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( 4.4 Fi na 1 P .A. line-up and results 80 ' .
5. CONCLUSIONS 88
Appendix - A S Parameter measure~ent ,
) A-l
Appendi x -, B "RFOPT" ànd stability analysis B-1 (
Appendix - 'C F;ita - 2 ,C-l
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&' ... ~;' ,
Appendix - 0 Optimized modules D-1
" Append1x - E' Measurement ~et-ups E .. 1
Bibliography Ir , x
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Fi g. ,1
Fi g. 2 "
Fig. 3
Fig. 4
Fi g. 5
Fig~ .6
jFi 9.7
Fig. 8 Fig. 9 ,Fi g. 10 Fi g. 11
Fig. 12 Fig. 13
Fig. 14 Fi g. 15
Fig. 16
Fig. 17 Fig. 18
Fig. 19 Fig. 20
Fig. ,21
Fig. 22
Fig. 23 Fig. 24
Fig. 25 Fig. 26 Fig. 27 Fig. 28
LIST ,OF ILl~STRATIONS i
FET Structures
MESFET Cross section GaAs FE~ ___ ~mall signal equivalent circuit
Temperature rise under the,gate v~ substrate
thickness and gate to gate spacing.
MESFET structu res
Gas FET power output ve rsus frequency
Schematic diagram of a TWT It. State of 'the art for U.S. high-power TWT's
Initial P.A. Une up
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1 nput power versus output power characteri sti cs
Computer aided approach
MIe c~nfiguration Mi ter ; n a ri ght angl e bend
L-Type matching network
Graphical solution of L-Type matchfng
network
L ahd, T type matching network
II -Type matchi ng network
Graphical solution of lI-type matching network
Real-to-complex impedance match
Microstrip ~teps
FET quiescint operating points
FET B f as modes
"Ha 1 f-moon" bi as structure
Bi as structure optimization
Hybrfd coupled class A power amplifier
90° Hybri d
Optimized. Hybrid coupled class A amplifier
Push-pull class AB power ampli fier
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6
8 8
15
17 23 25
.25
33
34
37 45
54
55
56
58 59
59 60
62
65
66 67
69
70
71 72
73
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Fig. 29 Fig. 30 Fi g. 31 Fig. 3? Fig. 33 Fi g'. 34 Fig. ,35 Fi g. 36 Fig. 37 Fig. 38 Fig. 39
Table 1 Table II Table III
'w" Tabre IV
Fig. Al 1 Fig. A2 1
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Rat-race coupler -Optimized push7pull cl ass AB power amp l ifi er Amplifier bandwiath' \ Input power versus output power Drain current versus input power' Efficiency versus input power AM to PM conversion Test Jig P .A. final Line-Up Pin diode attenuator Direct10nal coupler
MESFET Contact Technologies Teflon-f1berglass vs al umi na Deviee Evaluation -------------
Results
Sma 11 si gn a 1 Large-signal
S-parameter arrangement S-parameter arrangement
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viii PAGE
73 'l, Il
74 77
'" 77 80 80 79 80 --
(83 84 84
18 35 82 -87 -
Al A2
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1. INTRODUCTION
Modem communication systems require high poweor microwave amp li fi ers with good OC to RF convers ion effi ci ency, low AM - to - PM
'conversion, high lineaTity, low noise figure, small size, low cast and
h i gh re li ab il i ty •
This thesis reports the study, analysis, design and practical
development of a 6 GHz, 6 watt Field Effect Transistor (FET) amplifier
to be used interchangeably for anal09 and digital (16Q.A.M.) microwave
'radiolink transmission systems.
In the analog application the amplifier is ta be used in a
compression mode, i.e., the drive level ta thé power amplifier will be
adJusted sa that the AM envelope modulation is compressed about 10 dB while maintaining AM/PM conversion at an acceptable level.
, This mode of operation avoids the need for an intennediate
frequency limiter pl us subsequent bandpass fil ter and group delay
equalfzer in' the transmitter input.
In the digital application the power amplifier is ta be used in
a lfnear mode. Hence the gain should be as lfnear as possible ove~the /'
dynamic range of the digital signal. It is assumed that the optimum
RMS power level for digital application will be appraximately 5 dB
below the ana109 operating level.
In ,the second chapter thfs thes1s reviews the extraordinary • 1 "
progress achfeved with Field Effect Transistors. Varfous microwave FET ,
structures are de,scribed and what is being done in the field by
various laboratories te achfeve better devices wfth respect to power
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capability, gain, reliability etc., is reviewed.
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The thirds; chapter descri bes
travelling wave tubes which are
medi um power hi gh effi ci ency the '~mai n contenders wi th FET
amplifiers for radio link systelW'applications.
The fourth chapter presents the design of a modular 6 GHz, 6 watt
FET amplifier. The "line-up" to meet the requirements for the power amp l ifi er i s shown; soft teflon fi bergl ass i s used for the MIe
substrate, and a camparisan with alumina substrate is made. The
computer ai ded desi gn approach used duri n9 the desi gn i s shawn and
stability considerations reviewed. Fita-2 J which is a software tool developed by the author, generates a table of microstrip line
J?arameters to accurately reproduce the optimized microwave integrated
circuit (MIe) topology, is presented. The matching networks used in the
'modules are shown and sorne other match; n9 structures revi ewed.
A comparative study of a push-pull cl ass AB arnpl ifi er and of a
hybrid coupled class A power amplifier, with the!'purpose to show the' suitabilîtyof push-pull class AB power amplifier in radio 1ink
application, is presented.
F1nally the chapter presents the eva14ation of the devices used in the power amplifier, the final "line-up" ~nd the rnea~ured results.
The fifth and canel udi 09 chapter presents the claims for _ originalfty
recolllllendati ons arising from this work for
sUllII1ari zes the resu l ts. of the work and makes
further investigations.
2
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A major contribution of this thesis is the realization of Cl 6 GHz, 6 Watt, 500' MHz bandwidth power amplifier that meets the main requirements for analog and digital transmission. The other major
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contribution is the investigation'\ of push-pull class AB microwave power amplification and also the investigation of hybrîd-coupled class A power ampl ifi er perfonnance wi th respèct to two di·fferent bi as modes
q,
of operation (constant gain vol tage and constant drain current) as discussed in detail in the body of this thesis.
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2. FIELD EFFECT TRANSISTOR DEVELOPMENT
After Shockley and his co-workers i~nted the bipolar transistor in 1948, he proposed in 1952 a new type of transistor which he called the .unipolar field-effect transistor (FET) in which the conductivity of a layer field [1].
l-band with
of semi-conductor is modulated by a transverse electric
In ~965 germanium transistors became available for
a noise figure ·under 6 dB. Also in 1965 R. Engelbrecht and K. Kurokawa [2] developed a wide band low noise L-band balanced
transistor amplifier and in 1968 the AT & T System adopte~ the balanced transistor amplifier in its Sand C-band microwave communication links. Since then, significant progress has been made in
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obtaining low-noise performance and high po~er capability~from bipolar transistors for frequencies reaching up to the X-band. At 8 GHz power transistors with 1 watt CW output power and 6'dB power gain have been reported [3]. At 3GHz silicon bipolar transistors delivering 15 watts C.W. output power with 4.8 dB gain and 38 percent collector efficiency have been developed [4]. Most recently (1980) a microwave dual transistor delivering 100 watts C.W. output power with a collector efficiency above 60t operating near 1 GHz has been reported [5J.
By 1971, however, ,major developments were achi eved in fi el d effect
transistors. Today GaAs metal semi-conductor field effect transistors (MESF,ET'S) have higher gains, higher power amplification efficiency
and lower noise figure than bipolar transistors. More important is the fact that FET's promise a great deal of potential for further advances
1n the near future because of the following:
4
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1. A large variety of FET structures (MESFET, JFET, IGFET) are suitable for microwave applications, and some' promising structures are now in an early stage of development.
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2. A variety of semi-conductor materi~ls are competing for application in FETls.
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3. Further miniaturization to sub-micron dimensions can be realized in most FET structures.
4. The possibility of integration of circuits on semi-insulating substrates enables device isolat(on with low parasitic capacitances, 10w 10ss interconnections, and high packing
,
density.
2.1 MICROWAVE FET STRUCTURES
A cross section of all FETls structures is given in Ffg. 1 which summarizes all the technologies available at ahi time. The field effect transistors with ]nsulated gates (IGFET I S are of interest for power ampl ifi cati on because they offer the fol ow; n9 advantages over MESFETls and JFET's:
1- Ve~ sma'l dependence of input capacitance and transconductance on the gate voltage.
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2. Independence of output capacitance on drain voltage. ,w
, 3. Larger gate vol tage range (-Vgs to + V 9S).
Due ta these factors, very 1 inear power amplification can be achieved with IGFET's. Unfortunately very little success has been
achieved with GaAs, and practical IGFET ' s have only been made on Si. However up to now very few useful results have been reported above 1
GHz. Recently a microwave silicon power MOSFET that can deliver 22 W
of output power at 1.1 Ghz With 8.5 dB gain has been reported (11).
The Junction Field Effect Transistor (JFiT) has shown sorne promising results for power amplification. For instance, Vergnolle (6)
has built a JFET with 1.5 pm gate length and 611111 total gate width
which delivers 1 watt output power with 6 dB power gain and 26 percent
power added efficiency at 6 GHz. Since then however not too many papers/have been published on microwave JFET'sjmost of the efforts seem to be concentrated on Power MESFET 15 despi te the fact that sorne new
promising structures like DIFETs (distributed interaction FET)' have been proposed (31).
$0 far, the Metal Semiconductor Field Effect Transistor (MESFET)
has be~n the most successful structure above 2 GHz for power amplification as well as low noise amplification. This is due ta the
fact that two critical dimensions, the gate length and channel . thickness can be very we11 controne~ and al so due to the easy
1
realizati on on GaAs compared to the other three structures shown in ffgure 1. Since Mead (7) and Hooper (8) who, in 1966 and 1967
respectively, presented the first devices using this -,'structure, the
devfces have shawn extrordinary progresse In March 1970, Middlehoek
(9) realized a silicon MESFET with I-pm gate length which presented a maximum frequency of oscillation compara~le to the best bipolar transistors at that Ume (fmax=12 GHz). The fi rst l-}Jm gaté GaAs
MESFET became avaflable in April 1970 and was described by Drangeid, et al(lO).
7
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Semi-lllllUletlnQ GeAa SuMtrate
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Fig- 3 GaAs FET small signal equivalent circuit
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The extrordinary emphasfs given to GaAs FET fnstead of silicon 1s due basfcally to the fact that 1n GaAs the electron bulk mobilfty is s1x timès' larger than 1n silicon and the peak drift velocity 1s 2 times b1gger. Therefore the paras1tfc res1stances are smaller. the
, . transconductance 1s larger. and the transit time of"\electrons in the high, field region is st}orter: These properti,es .provide a lower noise
figure, hi gher gain, and a h1gher eut-off frequency t which are all important characteristics for microwave transistors. The cross section of a MESFET structure and its sma'l signal equivalent circuit are given in Fig. 2 and 3 respectively.
The activity of the FET is determined by its maximum frequency of oscillationWmax at which the FET becomes passive. From the intrinsic model of the FET given in fig. 3 the admittance matrix Y in the following two port:
·(I):[V](V) 18)
15 given by
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'C Y ::1
where:
2 2 1 + CI) Ti
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Ti = CiR i and Go = l/Ro. The FET will be passive when
y + yT* ~ 0
gm
or
How in practice ~!» ;'('1' (lb) can then be approxirnated by
~x~1C1Go - 9m2 ~ (&)2maxgm2~12 ~ 0
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rèsulting in
') 10
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. Usually for a 1 }lm gate length FET. the maximum frequency of
oscillation 1s a60ut 30 - 40 GHz but for the extri nsic FET the ~ .
parasitic elements reduce Wmax cons; derably. The maximum avail able
gain, Ga, max of the extrinsic model can be approximated by:
Ga ,max ::: Cl~ Wr - II.) r 3. )
\ "
Wr = 9m!(Ci , + Cf) ~
Q = [4G (R. + R + !w.L ) + 2w.Cf
(R. + R + w.L )J-1 01 S 15 Ils 15
, From (3) it 1s seen that the FET presents a gai n roll-off of
about 6dB/octave.
At the present t1me the objectives which workers in the field are
striv1 n9 for are ta: .
a) Obtain the maximum gate width for the frequency of interest ta
handle h1gh Ids,'
b) ])n1m1Ze e1eétrical paras1t1cs.
\ \ j
c) Reduce ~hermal impedance.
d) Ach1eve a dev1ce capable of susta1nfng a high drain source
potent1al.
,
e) Improve dev~ce efffcfency and linearfty.
11
~l,
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f) Optimize the fabrication technology aiming for consistency and reproducibility.
2.1.1 MAXIMlZATION OF, THE GATE WIDTH
One approach to maximize the gate wi dth is that of the crossovér
structure which has been utilized very succes'sfully by NEC and
,Fujitsu(12). The m~in advantage of this approach, is !~ it makes the
most efficient use.of ~real estate" and minimizes t~~mber of bonds
" required to assemble a large device: The drawback of this process is
its complexity and thè additional parasitics involved in the .. ' capacitanèe of the crossover, but both manufacturers have been able to
produce high power internally matched devices.
The second approach simply involves paralleling a large number of
uni't cells by wir~ bonding. Many laboratories such as Texas Instuments
and Bell Laboratories have adopted this approach because' of the simple
te,chnology involved despite the' device assembly difficulty and , increased parasitics. Nevertheless gate widths of 9.6mm for operation at 9 GHz and 24 mm for operation at 4 GHz have bee'n successfully
assembled (13,14)
Il ,"
A novel via hole plated heat sink structure has been introduced
recently by FUJitsu. In this structure, the source grounding pads
which are fabri cated outs; de the acti ve area, are connecteod di rectly
to the plated heat sin~ through holes made Just underneath these pads.
Fujitsu cl aims that the thermal resi stance of vi a hol e FETs can be
improved by about 40% over that of sheet grounded FETs" the .source
inductance reduced by one-fourth and 9% better effi ci ency can be
achfeved.
'12
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L
, , \.
o
The 'third approach, ffrst introduced by RCA, and now used by MSC / and Mitsubtshi, consists of "fltp chipping" plated-up source pads to a package heat sink. to achieve' paralleling. A 1200 pm K-band power GaAs
FET 'has been developed by MSC and 27 dBm utput power with 5 dB gain at 21 GHz' have been reported (15). A 10 GHz, 10 watts, internally matchèd Flip-chip GaAs Power FET have also been reported by Mitsubishi (16) •
A unique chip-level cell combiner technique has been proposed by Hughes Atrcraft Co. (17), 2 Watts at 15 GHz and 1.25 Watts at 18 GHz
have been achieved using this technique.
A useful way of compari ng dt fferent dev; ces i s through the fi gu re of merit proposed by a research team from ReA Laboratori es which 111 ustrates the effect of total gate wi dth. The proposed fi gure of \ mèrit MF is given as
MF =
Where Pa 15 the RF added power (Pout-Pin) in "", f 1s the frequency 1 n GHz. and W 15 the total gate wt dth 1n microns. For the same power ovtput and the same frequency of operation, a higher value of MF indicates a smaller gate width. -As a consequence a higher MF imp11es easier-matching.
2.1.2 MINIMlZATION OF PARASITICS
From Equation 3 which gives the maximum available gain for a small-t '
'signal FET we see t~e importance of minimiz1ng the parasftics to achieve reasonable gain and high power output for microwave FETs.
J
13
.. '
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14 , ,
Ta maximize gâ";, a critical dimension in the device geometry is o
the gate length since the shorter the gate length, the greater the gm :'~
and a-sma~ler Ci is obtained resulting in a larger t.
Another important- parasitic effect that has ta be minimized in "power FErs is the source lead inductance Ls. Sinc~ the power FETs are obta,ined by combini ng several cell s with each cell having several , parallel gate fingers, it is evidént' that the 'higher the power and the higher the frequency more crucial the spurce lead inductance becomes.
The cell combining efffciency can be defined as: 1
11 = Pout (n cell s) nPout (l cell) cOhSt..gain
- ,
Different techniques hav~ been developed and are in development to minimize the source lead inductance and at this 'point it fs not
;
obvious which of the approaches for reducing source lead inductance is the best.
o
2.1.3 REDUCTION OF THERMAL IMPEDANCE
The FET, as any other solid state device, shows deteriorated '>-,
performance as, its operatfng temperature increases. The operating temperature is determined by bath ambient and dr1ving conditions. The
~ tel1Jj)erature rise in a FET is conveniently treated in tenns of the thermal impedance which 1s a function of many vari ables such as substrate thickness and gate-to-gate' separation. At the moment, besides the flip-chip mounted FET developed by RCA and used by ~C and Mitsubish1 all the others are mount~d upright in the heat sink sa that the heat must be remove~ through the GaAs substrate •
• J ... ,
!
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t f
1
t
(
()
Fi g. 4 shows the maximum temperature ri se under the gate as a function of substrate thfckness and gate-to-gate spaci ng.' From the figure it fs obvious that the substrate thickness plays a maJor role 1 n detennini ng the temperature rhe, but for substrate thickness of 50
pm or more the gate-to-gate spaci n9 becomes of great importance. • .
'oJ
• ~ .. 1 -l .. e .. • ~
- .. JJ
v - 3e~-;.te to $lIte a • 181' • \J8te ta 9".
25 50 75
Subltme Thlclcneu (.L'm)
o'
100
. 15
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() i
1 1
1 n the case of the flip chip mounted type the heat 1s removed
through the Au source pads but the heat also spreads into t~e GaAs
and this spreading resistance is the domi nant factor in the thennal res1stance of these dev1ces.
A study done by Wemple and Simon (18) showed that an upright FET,
with a substrate thickness of 50 }Jm gives the same thennal resistance as an equivalent flip chip mounted device.
As we can see the advantage of one configuration Qver another is
'dependent on the detail s of the dey; ce geometry and mounti ng confi guration. More generally one can say that for thick GaAs
substrates, flfp chfp mounting will have then1lal advantage while for thi n substrate upri ght FET can have thenna l advantage.
2.1.4 FET STRUCTURES AND SOURCE-DRAIN BURN OUT
ln 1976, Wemple and Niehans (19) demonstrated that avalanche at "imperfect" drain! ohmic contacts trlggers thennal runaway in the
substrate and suggested the' N+ epi' under the contacts woul d improve
device voltage capability. This technique is used at the moment by many manufacturers.
Fi gure 5 presents the schematic of va'ri ous MESFET structures used {?
at thfs time a,nd Table 1 summarizes the different contact technologiès in use to increase the source-drain burnout voltage.
16
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;.. ,
1
1
l 1
i 1
J
t
"
~i
,
(
~Al'(
o Basic
,~\
\ . , . .J
o Bell
@NEC
Fig- 5 .
@ Fujitsu
H.P,
Il' .H-
@ MSC
Mitsubishi
0, FUjitsu
17
..
O'
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L
• , ... Table l
1 LABORATOR~ TECHNOLOGY 1 OHMIC CONT.
1 1 1
1 1 1
1 1 N+ inla1d source and drain 1
1 FUJITSU 1 islands fonned by selective 1 Au Ge/Au
, 1 1 épi taxi al regrowth epi channel!
1 1 1 N+ inlaid source and -drain 1
1 islands fonned by ion !
1 H.P impl antation, channel formed 1 NiC r/Ge/Au
1 by island implantation. 1 \
1 1
1 N+ epi pl anar source and drai ~
1 BTL contacts, epi channel. 1 AuGe/Ag/Au
1 1
! Recessed channel wi th N+ 1 Au-Ge-Nl
1 BURNOUT @
" lOOma/mm
1
1 Bi gger
1 than 26 V
Bigger
than 30 V
52 V
1 MSC 1 contacts and self ali gned 1 (drain&source}1 B;gger
1
1
1
1 NEC
1
1 gates
1 1 Recessed channels,
1 epi channel.
1 Ti-W-Au(gate) 1 than 25 V
1 1 no N+ epi, 1 1 Up to
1 AuGe/Pt 1 22-25 V
1 1
So far. Nippon 1s one compal'\Y that has not adopted the N+ epi
under the contacts, contend1ng that N+ contacts are not required to
achieve high source-drain burnout. Nippon's recessed gate structure " reduces the f1eld at-the dra1n contact and burnout vol tage of up to
25V 1s achieved with this structure w1thout the N+ epi under the
contact.
Fuj1tsu have also recently adopted a recessed gate structure with
N+ contacts for some of the1r recent power FET' s. (FLC "1" famlly).
18
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2.1.5 FABRICATION TECHNOLOGY
Since material requirements for FETls are very stringent, the
technology for their fabrication must be capable of grow;ng
mul til ayer, thin uniform epitaxi al structures with dopi ng vari ati ons in the range of 1013_1018 cm-3 and producing abrupt transitions between the different layers. Surface smoothness is also necessary for device processing.
At the moment the materi al technology most widely used for
producing FET structures is Vapor Phase Epitaxy (V'.P.E.)(20,21,22).
Liquid-Phase epitaxy (LPE}{23) and ion implantati~n (24) have
demonstrated promise as a technique for producing material, but most
of the results have been on low nqise devices.
Another very promising technique is the Molecular Bearn Epitaxy (MBE) ow; ng to ; ts preci se control of impurity i ncdrporati on and
thickness. Ultra-short channel (O.25)Jm) low-noise GaAs FET I s prepared by MBE have been reported to exhi bit a noi se fi gure of 1.5 dB at 8 GHz
(23). 1.9 dB with an associated gain of 8.5 dB at 12 GHz have also
been reported (25). The MBE technique holds promise for high power FET application due to the higher gate breakdown voltage compared ta the FET I s prepared by LPE and VPE. Some high power FET with favorable
results have been already reported (26). Metal organic chemical vapor deposition (MO-COV) seerns to be an'Other very promising technique, Toshiba have reported a 0.5 pm gate FET with noise figure of 1.8 dB at 8 GHz and 11 dB gain, using this technology.
Other new technologies are being intensively investigate~ such as: electron beam lithography,self alignment patterning, ion-bearn milling, surface passivation, and laser alloying of high-temperature metallic
L
contacts ta alium rsen1de.
19
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The outlook for material technologies seems to be bright and many
improvements are expected ·in the near future •. Further better device
performance and quality associated with higher production yield
leading to lower cast are expected.
2.2 FET RELIABILITY
After about 10 years of evolutionary development, power GAaS FET's
have finally gained the confidence of high reliabilfty system l
designers. This confidence is substantiated by life tests performed by
the military services,systems houses, transistor manufacturers and above all by field results already obtained. MSC, NEC and Fujitsu have
carried out exte'nsive reliability studies of power GaAs FET' s as
described below.
MSC carried out studies on commercially available MSC 88004 devices (27). Accelerated 1 ife tests performed wi th these dey; ces
have shawn that under dc bias a MTTF of 5 x 106 hours at a ch~nnel temperature of 125°C is expected. The RF failure critericnused in all
testing was a + 1 dB change in power output at 6 GHz in a fixed tuned
circuit at a fixed RF drive level.
A t the Naval Research Laboratori es (NRL), li fe tes ts were made on
MSC 88002'5 (28). Spec1fying again failure as 1 dB degradation of
amplHier gain while operating '#at 125°C channel temperature the
results i~dicate an MTTF of 2.5 X 106 hours. Further tests at NRL of devices fram different manufacturers indicate that gold refractory
_gate metallization is highly reliable.
Recently, a new technology ta connect the source islands in the
flip-chip. type power FET called the Plated Source Bridge (PSB)
structure has been reported by Mitsubushi (29). This new technology
has improved the work efficiency, the production yfeld and the thermal
resfstance.
20
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'J
The thennal resistance improvement resulted in a two-fold increase r
in the MTTF (107 hours) compared to conventional flip-chip devices.
J
Some advantages of the MSC fl ip-chip device are that by using a
gold based 5chottky-barrier gate it is not as sensitive to static
discharges and al10ws much greater gate current than aluminum shottky-barri er gates (NEC, FUJ ITSU).
Nippon has carried out a reliability study (30) based on their C
Band 2 watt GaAs Mesfet NE8684. 1 n order ta detennine failure modes
and reasonable operating condition limits for stable operation, a
biased step stress test was made us'ing ten units. The conditions were
an ambi ent temperature Ta=75° C, a drai n current Id = 500 ma and a
time step of 1.68 hours. The dra i Il vol tage Vds was i ncreased step-
wise from 9V in IV steps. Carrel ation between OC and RF
characteristics was also investigated and it was found that a 1 dB RF
gain degradation at a 1 dB gain compression point (fo=6 GHz) corresponded ta an IOSS decrease of 20-25% and for an apprximately
60% decrease of IDS.
Based on the rel i abi l i ty study a qual ity assurance program has
been designed in a way which praetically eliminates the initial and
catastrophic failures. The gradual degradation failures are claimed ta
be s!JPpressed to an extremely low poi nt by conducting a hi gh
temperature storage sereeni n9 process and an MTFF of more':' than 108
hours at Teh=125°C is elaimed for Nippon's devices. So far N'ippon has
reported outstanding field results, by the end of 1980 more than 2,250
Nippon power GaAs FET's have operated for more than 4600 hours in the
field with total deviee-hours exceeding 25 X 106 hours. During
th1s time only four fallures have occurred. Therefore, the MTBF of
NEC GaAs FETs in the fiel d ; s 6.25 x 106 hours. However it must be
noted that when the power GaAs FET amplifier 15 used as a TWT
substf tute, seven or ef ght GaAs FET 1 sare generally used.
if 'l'ch = channel temperature
21
, r
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Therefore, the total MTTF of the power GaAs' FET amp l ifi er using
Nippon devices 1s expected to be 900000 ta 700000 hours based on field
data which is at least 5 times better than the state-of-art-TWTs
amp 11ft ers.
Fujitsu has recently carried out a life test on their FLC 30
dev1ces, that eonsisted basically of a burn-in test of these devices
ehoseo 'randomly and tested at Vds = 12V, Vgs = -2V, f = 6.2 GHz,
Pi!! = '0.3 -0.4 W. Tc __ = 90°C, Teh:; 200°C for 14000 hrs. For a power -
output degradation of 1 dB or 1 20% variation of Idss, a MTTF of 107
hours at a channel temperature of 1000 C i s predi cted.
RCA has al sa tested a totalO-f -60 FUJitsu power F ET' s of the
FLC-30 type fam1ly. The transistors were installed in 4 GHz amplifier
modules a~~Q ven to + 27.5 dBm output. Temperature stress was 190
to 215°C and, .. the faflure criterion was a 0.3 dB drop in output power.
Reported ffit 'results are a med; an 1 He of 2500 hours at 250°C and
6500 hrs at 190°C. In normal satellite operation, the device temperature 15 expected to be about 100°C and for this temperature the
medium li fe 1 s proJected to more than 106 hours •
. ' The present performance of power output versus frequency shawn in
fi gure 6 represents a marked 1 mprovement for GaAs FET' S over the past
severa l years. Further improvement in power output J effi ci ency , gai n and reliabl1ity is expected in the, future.
At the present time MTTF's in the order of 106 to 108 have
been reported for power fET' s from di fferent manufacturers and fi eld <
experience has ver1fied that the GaAs FET ampl if1er can provide
outstand1ng rel1able service as the"power amplifier in radio-relay
systems and satell ite transponders.
22
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23
.(
POWER OUTPUT
// ' (W)
40
0
20 0 0 ....
10 • ~
B
B ( <4
2 • TI
0 NEC
1 .6 M.TSUBISH. • 0 BELL LABS
... FU.lnSU
~ RCA .4 ~)
0 lSI • • MSC
.2 • HUGHES • ~
.1
1 2 5 10 20 .. 0 100
FREQUENCY (GHz )
Fig. 6 GaAs FET Power Output. versus Frequency
L
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"- 24
t> i ! t 1·
I 1 ! !
L
3. HIGH EFFICIENCY Twr's
Since the GaAs FET's are the principle contenders for replacing
TWT's in radio link systems it is important ta examine TWT properties
and characteristics in arder ta make comparisons between the two
competing devices. The following discussion therefore considers medium .. power high efficiency TWT ' s.
Since Kompfner invented the helix travelling wave tube (TWT) in
1944 (42), its basic circuit has changed little. Figure 7 presents a
simplified circuit of a TWT. ("
Even sa, if we compare the older model glass-metal tube with its bulky beam focusing assembly and the mo'dern high efficiency TWT
metal-cerami c type which employs PPM (periodi c pennanent mag'net) ft i~ possible ta measure the significant progress achieved with these '0 tubes. The state of the art power output for high power TWTs is shown
in Figure 8.
Medium power high efficfency TWTs in the 6 GHz band (5.9 - 6.4 o ,
GHz) amang others are manufactured by Ni ppon El ectrf c ( LD4353) ,
Thomson-CSF (TH3600) and Siemens (RW89D). . ' Due ta the lower power consumption, increased packing density and
fmproved reliability (MTTF= 200000hrs), these tubes are very
attractive for use in radio relay systems.
A brief description of these TWT's and their electrical
characteristics is presented below,' based on manufacturer's data
sheets and measured results (43).
~ 1
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L
E lee tron VII"
Siow-wo". Uruc turc ~---_ .. ~-----Collectar
RF output
1 Helix
RF input
Anod.
1 1
,
Figure 7 Schematic Diagram of a TWT p
(W ) io . - ... 100 -- 1· .
•
10 -t- I-~ - • .. -
1-1-~
1- !-1-1 .
1 1 10
IFE
Cathode
/
t(GHz)
Figure 8 State of the art for U.S. high-power TWTs
Heoler
25
\
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\ 3.1 NEC L04353 TWT
This tube is of metal-ceramic construction with partial glass, a
coaxiallycoupled(SMA connector)and conductive cooling. It\,employs a
periodic permanent magnet focusing stack using samarium cobalt magnets with a high energy product. The TWT has a grounded helix and a doubled depressed collector for improved reliability and efficiency, which is quoted as 35.4~ at saturated output power (22.2 watts). AdJustment of
the helix and a[Jode voltages is available and necessary to cover the
specified freque~y band. The TWT allows a maxfmum RF drive of + 10 ...
dBm. At 6.4 GHz, EW= 3.05 Kvac, Po = 10W, the noise figure and the AM o
to PM conversion factor 1s specified respectively as 23.0 dB and 3.3
degrees IdB. ,The overall weight of the TWT plus power supply is 3.3
Kg.
3.2 SIEMENS RW89D .
These tubes are of metal-ce~ic type, coaxially coupled (N-type
connector). conduction cooled and employ periodic pennanent magnet beam .f0cusing. A doubled depressed collector is used and an' efficiency
of 30\ at saturated output power (15 watts) i s a~h1 eved. The only adJustment to thfs grounded helix TWT is ~he grid 2 voltage which
alters the cathode current anfl hence the output power by means ,of a 13
position switch on the boQy of the power supply. Measurements carried
out with this TWT showed a worst case AMIPM distortion of 2~3 °/dB, a -, nohe figure better than 21.4 dB, a maximum output power variation in
any 30 MHz band of 0.2 dB and a group. delay illich larger than .:!:. 90.2 nsec variation over a 30 MHz band. The overall weight of the TWT plus
power supply is 7.5 Kg. .< -
.. EW = helix voltage <:>
26
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o \ •
( ,
3.3 THOMSON-CSF TH3600
These tubes are also of metal-ceramic type, coaxiallycoupled (SMA
connector) and conduction cool ed. They al so employ peri odi c permanent magnet beam focusing. An impregnated cathode and double depressed collector is used in this TWT to improve re1iability and efficiency , which 1s quoted as 38% at saturated output power (15 watts). The anode voltage is the on1y adjustment available through a 12 position 'potentiometer located on the power supply. The maximum RF' dri,ve al10wed is + 5 d8m. The maximum AM/PM conversion and noise figure is specified respectively as 4°/dB and 24 dB. The output power variation
1 n any 30 MHz band i s specf fi ed as .3 dB and presents a ve ry sma 11
group del ay of !. .2 nsec (worst case) vari ation over any' 30 MHz band.
,3.3 Kg i s the overall weight of the TWT pl us power supply.
3.4 COMMENTS
These state of the art TWT ' s p'resent outstanding perfonnance compared to the al d type TWT and for many reasons they are extremely
attractive for use in microwave radiolink. products- not the least of which are the reduced size, weight and power consumptfon. Being
cooled by conductiçm,) the fact that coaxial connectors' are used
instead of waveguide and the fact that they were initially designed
for space applications suggests that the MTTF will nct be degraded. {MTTF = 200.0QOhrs.}
However. sorne new radio system designs are of a fonn for which TWT's are' becoming inappropriate. > Thus for example, none of these
. ~
TWT 1 s woul d meet the specification tor the present RA3-T6C FM Radio
system manufactured by Northern Tel ecom because of the i nabl1 i ty of these amplif1ers ta be operated in the lfmiting mode (Le., at + 41
dBm output power 2°/dB max AM/PM and 0 dB min. AM/AM compression) and also because the Siemens TWT has excessive group delay and the
,Thompson-CSF has excessive AM/PM.
\
27
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\
Another drawback of the TWTs is the fact that they are inherently 1"
non-lfnear dey; ces and therefore in order to be used in di gital radio /'
systems i t is necessary to back-off the power at l east 3dB more than • a solid state power amplifier to provide good linearity.
)
28
,J
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J,-
, , ,
, ~'
i'
l \ r
/
,
, ,
4. 6 GHZ, 6 WATTS FET POWER AMPLIFIER
This chapter discusses the develapment of a modular 6 GHz, 6 Watt microwave integrated circuit (MIe) FET power ~plifier for analog and digital radio application using MSC and Nippon devices. "'
The modular design approach was chosen because it offers many
advanta{es when campared w,ith a single in.tegrated unit. For instance, individual'-' modules can be tuned in test maunts prior ta "drop-in"
integration into ~he amplifier assembly, enabling optimum perfonnance to be achieve~ on each, module, and red~cing the time required for
tuning. repairs can be quickly carried out by s1mply replacing modules, and problems which arise upon integration can be readilYcidentified.
MSC devices were chosentor the amplifier "line- up" because MSC is a North American supplier w ose FET's have high reliability figures; Nippon series 868898 FETls ere used. in the power stage instead of the
MSC 88012 because the MSC devi ce di d not meet the requi rement as explained further in section 4.4.
In the analog application, the amplifier Jis intended to be used in
a compression mode, i.e.,the drive level to the P.A. will be adjusted so that A.M. envelope modulation ~s compresed about 10 'dB whl1e
maintaining AM/PM conversion at an acceptable level.
In the digital application the p.A. will be used in a linear mode over the approx1mately 12.5 dB dynarni c' range of the di gital si gnal •
29
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"
\
1
1
4.1 SPECIFICATION
As a consequence of the application needs outlined above, the power amplifier was designed to meet the followfng specifications: . , .,
1-
Analog Digital
1 nput Power (dBm)- \, - 5 - 0 - 13 - 4
- Output Power (dBm) Min. 37- Max.40 Min. 32- Max 34 - ~
Frequency Range (MHz) 5925-6425 5925-6425
Minimum - 10 dB AM/AM Pout 37 dBm
Maximum -10 dB AM/AM Pout 40 dBm
Smal1 Signal Gain 42 + 3 dB "
Small signal gain stability + 0.5 dB
with time and temperature
'j
"Pout @ -10 dBl AM/AM" stability + 0.5 dB with time and temperature
AM/PM @ -10 dB AM/AM Pout 4 deg/dB.
30
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f Output return 1055 1 nput..;. ~etturn 1055
1
Il 'Slope- (small signal and saturated)
Ripple (saturated)
Group Delay response Noi se Fi gure Harmonic Output
Power Dissipation
1 nput Connector
Output Connector
Output Moni tor
Output Detector
Cost (1982$)
Isodaptor 10ad rat1ng
RF leakage
oc Connector
( )
31
26 dB 9 dB
-.,,,&-- 0 .25/30MHz 0.1 dBi30MHz 0.03 dB/30 MHz
0.2 nsec/30MHz 7 dB
-20 dBc
40 Watt
SMA(female)
CRM137
-30 dBc + 2 dB (Calibrated)
0.5 Volt at + JO dBm
$1700
-~ -----~- -----
, 1 Watt
-70 dBc wfth waveguide adaptor at 4" from PA
to be speci fi ed
•
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\
, 1 i i
1 r
t 4.2 P .A. LINE-UP
To meet the specification shown above the amplifier 1 ine-up given in Figure 9 was initially adopted based on the devices, data sheet.
The level diagram shown in Figure la i1lustrates the expected power
amplifier performance as far as Pin vs Pout is concerned •.
This figure indfcates the essential linearity of the first and intennediate stages and indicates the non-linear and saturation properties of the final output which the amplifier development and design are intended ta achieve.
It was decided ta fabricate the MIC's on soft teflon-fiberglass substrate, Duro1d 5880, Sr = 2.2, instead of alumina.
A comparison between the teflon fiberglass material Duroid 5880 and the alumina substrate is given in Table II.
The table shows the advantages of teflon-fiberglass over alumina
which make the choice of Duroid 5880 for this particular application evident (Ref. 45, 46, 47). These advantages make it possible to satisfy the need for mechanical reliability, ease of mechanical and electric circuit fabrication as well as low cost.
..
32
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-
'" ~
1 1 , 1 1
IIp 1
INPUT MODULE
.. -?-" ~-.~ "~1-- ~_ "'-.<-
TEMPERA.TURE COMPENSATION MODULE
DRIVER MODULE
..
INITIAL P,A. LINE UP Flg.9
l ' 1 1 1 , 1
-' •
OUTPUT MODULE
~ , ,
o/p
w w
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, ~
i· t
1
i
f
( )
L-
P out
(dBm)
40
30
20
10
o
o
PAR. MSC88012
J' --_. MsC88012 . 4
MSC88004
MSC88002
MSC88001
__ ---ilI------MSC88000
1
~ 1 --..-- .c---...
t T.c..lou
~--------~--------r-------~--------~----------------~ -10 o 10 20 30 P.
ln
(dBm)
FIV. 10
PIN vs POUT
"
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35
-1 .....
1 DUROID 1 ALUMINA 1 5880
1 1 Dielectric Constant 1 2.2 1 9.9
1 1 Loss Factor ,
1 0.0008 1 0.00004
1
Thermal Conductivity 1
Low 1
High
U seful Tem erature -60 to 200°C 1 U to 500°C
Machinabil i ty Excell ent Poor , r
C racki ng None Frequent
" Etching Definition Good Excellent
Flexibility Good Very Poor
P rocessi n9 Standard PCB Special' Number of $ubstrate Interconnections Law
1
High
Vibration Damping Excellent l,
Poor
Cl 1 Water Absorption Low
1 Law
Relfabi1ity Excellent Good
Trimm1ng Easy Difficult Component Mounting Sol der by hand Sol der on
Reflow Machine FET Mounting Sol der by hand 1 Preheat and
.1 sol der by hand 1
Production Cost Low 1 High
( \
TABLE II
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-4.3 DESIGN APPROACH
The basic design approach utilized for the design of each module is shown in Figure Il. The input and driver modules were designed and optimized for maximum gain while the output module was optimized for
maximum output power,
Since the devices were not available on time to have their S-parameters characteri zed, the data sheet s-parameters were used for
the desi gn of the matchi n9 networks and because of that, sorne bench fine tuning was expected.
Later on, the devices were characterized -fOr large signal
(511,522) and srnall signal (512, S21) parameters and their values plus the measurement arrangement are shown in Appendix
A.
4.3.1 STABILITY CONSIDERATION
A major concern dur; n9 the desi gn of the amp 1 ifi er was to ensure that every amplifier module would be operating in a safe stable
region, before integration.
Consider1ng the FET as a 2 port network one can write:
4.1 )
4.2 )
36
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I-I
! ,
1 )
COMPUTER AIDED DESIGN APPROACH
CIRCUIT
SPECIFICATION
INITIAL CIRCUIT'
DESIGN
,--( sm_lth ....L-.c~_art )-----,1 1 COMPUTER
OPTIMI ZATION (RFOPT)
DEVIeE EVALUATION +
SELECTION
l-fODULE
OPTIMI ZATION
-------
MODEL$]
Figure 11
.,.- P.. ': input retutn 108s
~ ': output return 10BB
------- ---DESIGN DATA
+ SYNTHESIS
,
NO
YES
INTEGRA TION
37
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38
Manipulating the above equations yie1ds:
SI = S + S12S21 fL ,
1> 4.3 11 11 1 f S
- L 22
S12S2,fs s22 = S22 + l - r
ss
11 4.4
Wher~ 511 is the input reflection coefficient with arbitrary '-
ZL and 522 is the output ref1ection coefficient with arbitrary
ZS·
To simultaneously match the input and output of a this network
impl1 es °that 511 :. r: and s~· rt- Sol vi 09 4.3 and 4.4 \ s imu1 taneous 1y l eads to the source and load refl ecti on coeffici ents
(rSM ~ rLM ) ta achieve thi s simul taneous matchi n9:
4.5
4.6
l
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where:
---If B. > 0 (i = 1, 2) ~ the minus sign is used, otherwise the plus
l
sign is used.
5ince:
and if RE (Zsm) and RE(ZlM) are always positive then
1 r sm 1 < 1 and 1 r LM 1 < l
From (4.5) and (4.6) the Rollet factor can be derived.
1 - 1 51" 1 2 - 1 5221 2 + 1 fJ 1
K = ---------
39
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, . ,
The necessary and suff1cient conditions for simultaneous matching of a two port networ~ are met when K i s greater than 1.
The network is called unconditionally stable when K 1s gréater than 1 and ~ < 1 , which means that the input and output impedances rerna1n positive for all passive load and 'source impedances. In th1s ~ase the maximum availab-1e power ga1~ 1s given by:
Plus si gn used when B1 < 0
Minus sign used
U nfortunate ly most of the power FET 1 S avail ab 1 e present KI s sma11er than 1 which means potential1y unstable devices, that 1s, the rea1 part of thefr input and output impedances can be negative for sorne source and load impedances.
T 0 guarantee stabi 11 ty for an amp 11 f1 er over a w1 de range of frequencfes when K 15 less than 1, severa1 methods are availab1e:
l
40
'.
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~ .. {
(
1 -
A) External Feedback
Through externa 1 feedback the network can be made uncond1t1onally stable. Due to the difficulties fn'lolved in . construct1ng .an MIe feedback network in- this frequency
l'
range this method was not adopted .
.., " B) Damping Resistance R
1
This method consists of coupling a resistance ,in parallel
with the u~stable port as shown below:
---- --------...,. J
" 1 r- ~ ---, ,- ---1 t l' 1 ' 1 1 1 ,. 1
1 1 1 1
1 1 1 l' 1
1 1 1 1 r:
,\ 1 ri :l r: ' +R'" :l :l
1 1 , 1 1 , 1 • -' Su Su S22 $11 ~2 S22
1 1 1 1 1 1
1 1 1 1
J 1 1 ----- 1 -----1;- - - ,1 - --- --------
41
2.
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( )
The 5 parameter of the res1stor can be eas11y ca1cu1ated as:
y
y + 2
2
y of: 2
The new 5-parameters of the comb 1 ned network then are:
S12521Si, Si', = 511 + ------
1 - 5225;1
512S;2 sn iii
12 1 - 5225;, 0
521 521 sn •
21 1 - 5225; 1
! , . , ~
42
/
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t
1 t
\
i
Solving the system for 522 satisfying the conditionthatl 5221
is less than 1 we can calculate the maximum value of R.
R < Z o
/
/
Any resistor satisfying this condition will 'guarantee the
amp 1 ifi er stabil i ty by mak i ng ,,/,S22 ''1 < 1 . - . -.-
C} Gain Reduction
This is the most straight-forward method of" achieving
stability for a conditionally stable -device, and was the
method used for this ampl ifier _design.
Basically this method consists of determi ning which source -
and load impedances provide stable operations. This can be
done by plotting the input and output stability circles whose
centers and radi i are expressed by:
cél\ter of the ;nptt plane: Cs = C~ ,
43
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, . " 1
\,
t \' f , ~
t ,
Center on the output plane = C., =
Radius on the output plane = rl =
where: C, = S11 - 6,~22 p
C2 = S22 6$, i "
Â' = S"S22 - 5'25 21
C* 2
8y pl otti ng these ci rc1es for many frequenci es in the bandwi dth of
i nterest. a very c1ear picture of the areas of safe operation in the
input and output plane is obtai ned.
The next step is ta design the input and output matchfng network
outsi de this area of instab111ty and optfmi ze them over the ban~i dth
of interest to make sure the amplifier'will meet the specification for
gain, input return 10ss, output return 10ss etc. ~ver the bandwi dth.
In Appendix B a brief description of the software too1 "RFOPT"
used for the stabil1ty analysis and circuitry optfmizat1on 1s given
and the stabfli ty analysis results for some of the devices are shown.
, 44
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f -r
(, ,
4.~3.2 FITA - 2
This section discusses the computational method and the assoc1ated computer prograllll\ing (FITA 2) which was developed by the author for the purpose of designfng the amplifier system. It was
essential 'ta, achieve a very accurate MIe. circuit, reproducing the optimized computer model 1n order to minimize. the trimming process. , . To a.chieve th1s, a program that generated'. a table- of microstrip 1 ine parameters was developed based on recent pub li shed works.
The microstrip line configuration shown in Figure 12 is a typical
microstrip line structure. where a strip conductor is separated
t
, . Strip
Conductor. ,
,
f
& , w \ ( 8.,
. •
~ H . . . , .. . . , . , . .. . . ,
, t ••• " .. ' : . . . .. T
Figure 12
45
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1· r
( from the ground plane by a dielectric substrate and since not a11 the
field lines are contained in the substrate, the propagating mode
along the strip is not purely T.LMoJ but a quasi-T..E.M.mode 'of
propagation can De assumed.
In this c.ase the phase velocfty in the microstrip 1s given by :
c
Where C is' the velocity of light, and Beff is the effective
dielectric constant of the substrate, material. The effective
dielectric constant is lower than the relative dielectic constant,
since it takes into account the external fiel ds.
The wave length 15 9iven by:
V À = --L =
9 f
\
The characteristic impedance 1s given by :
z = o
1
VpC
c
Where C is the capacitance per unit length of the line.
'.
46
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\
At lower microwave frequencies the evaluation of 6eff and C based
on the QUASI-TEM mode 15 fairly accurate. At higher frequencies, however the ratio of longitudinal-to-transverse electric field becomes
significant and can no longer be considered QUASI-TEM.
Since the avaflable numerical methods for the characterization of microstrip lfnes invo],ve extensive. computations, closed fonn
expressions that had been reported recently were used i nstead in the development of FITA2, resulting in a more efficient program.
For the microstrip impedance and. effective dielectric constant,
Hammerstad (48) expressions were used, which fall within + 1 percent of Wheeler l s numerical results.
for w/h < l
where:
z = o
for w/h > l
where:
z = o
60 ----.-(1 n (8h/w + 0.2 5w/h) (Eeff )
E + 1 r 2 +
Er - 1[(1 + l2h/w)-~ + 0.04(1-w/h)2]
2
w/h + 1.393 + 0.667Ln(w/h + 1.444)
E - 1 Er + 1 = ----:~- +
2 _r .....
2--(1 + 12h/w)-~
47
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Hammerstad's expressions for "th in terms of Zo and ér are:
" for w/h ~ 2
W _ BeA 11- e2A _ 2
for w/h > 2 E - '1 0.61
!! = f. [B - 1 - 1 n(2B - 1) + --=-r---l(Ln(B - 1) + 0.39 - ---)] h 11 2Er Er
where:
A ==
Z e: '+ l o ( r 2 )112+
60
377n B = ---...---
o 2Zo
(E:r)!
These expressions are accurate for a practica1 range of microstrip li nes:
0.05 ~ w/h ~ 20 and Er ~ 16
48
.... #,'
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i ....
!
i !
i
The formulas given above assume the strip thickness (t) equal to zero, these fonnulas ,can be modified to consider the strip thickness (t) by replacing the strip width (Iii) by an effective strip width 'iie). Expressi ons for we are
1 we w t 2h il = 'il + 1Th (1 + ln .t) for w/h ~
27r
2'71'
we w t 47TW 11 = h + 7Th (1 + Ln -t-)
for w/h <
Additional restrictions for applying the equations above is that:
t < h and t < W/2
The formulas presented for char~cteristlc impedance and effective dielectric constant are based on a QUASI-TEM mode of propagation which 15 a good static' approximation at lower frequencfes. As frequency 1 ncreases the effective di el ectri c constant and characteri stic i mpedance begi n to change due to the propagati on of hybr1 d modes making the transmission line dispersive. Fortunately, the changes of Seff and Zo are smal1. The frequency below which the dispersion effects May be neglected is give,n by Chudobiack (49) through the relation
fO(GHz) = 0.3 [Zo ! ] , . h(e: - 1) , . . r
(h in cm)
. ,
, . _. T.
, . - - . -( .' ,." , 6' ...... ':b ':. ~ .
49
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Analytical fO!IDUlas for di spersion whi ch agree closely w1th
experimental and numerica1 results have been presented by Gets1nger (50) and are given below:
.. Er - E:.eff
1 + G (f/fp)2
where:
G = 0.6 + 0.009Zo
(f in GHz' substrate thickness in cm)
~----- ----- -
From the Getsingerexpresion we can see that if fp» f,éeff Cf) =
E eff. In other words, high impedance li nes on th1n substrates are less
di spersive.
C10sed form expressions of the frequency de pende nt behavior of Zo . based on a parall el-plate model of mi crostrip 11 ne have been reported by Owens (51).
These eXp'ressi ons are:
377h
and he effect,i ve wi dth, weff (f) i s:
t
1 _--___ _ If) --L l
/
50
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\
\ , )
,.
! { t ~ f t l , \
Where Weff (0) 1s obtained from the expression of zoe f) when
f = O.
Another very important characteristi c of any transmi ssi on 11 ne 15 its attenuatfon constant. In a mfcrostrfp circuit there are two,
sources of dissipative lasses: conductar 10ss and substrate die1ectric 10ss.
Pucel (52) derived very accurate closed fonn expressions for the conductor losses as fol1ows:
For w/h < 1/21T -a.6aRs a = c 21TZoh
For 1/2IT < w/h ~ 2
Cl = c
For w/h ~ 2
a = C
a.6aRs 21TZoh
-:-.-.."':,
[1 h h ( 4.h . t)] P . + - + -- Ln-+-we 1TWe t w
. P. Q ,
• Q. [ we + :.. 1 n ( 21Te (...!L + 0.94» r 2
h 'lT 2h
we/1Th we .(-+---
-h we + 0.94
2h
51
1
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1
Where
( we 2
P = 1-) 4h
Q = 1 + J! + .lL (Ln 2h _ 1.) we nwe t h
1l :: free space <!'penneability o
~ :: conductivity of the microstrip material
From these fonnul as one can see that for a fi ~ed impedance the
conductor 10ss decreases inversely with substrate thickness and
increases with the square root of frequency.
Dielectric losses are norma11y v~ry smal1 eompared with conductor
1 osses» however they cannot be negl ected in mi crowa ve ci reu i ts
requi ri ng sma 11 attenuatf on. SChnei,der (53) deri ved the closed form
expressions for the attenuatfon constant given below.
where:
E:eff - 1
E:r - l
Ào = free space wavelength
tanô = l oss tangent
tanô dB/cm
52
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i t.
f 1 i
1 1
l 1
(
'" 1
The circuit quality factor can be related to the total lasses in
the 1 i ne by the express ion:
21T where: 13 1:
Since in many cases the MIe structure does not cons1st of straight microstrip structures on1y, and bended lines are required, there is a
need to minimize the bend effect on the circuitry.
R. Douvill and D. James (53) made an interesting experimental stuqy of symmetric mfcrostrip bends and the influence of mitering was
" i nvestigated.
A useful empirical expression for' the optimum percentage of miter ~
M for a right angle bend, as shown in figure 13, was deve10ped. t.
~,' x '~ ________ ~ ,
,--"--____ ..J
M itre=Lx100% d
,
•
. Bend Equivalent Circuit
Fi g. 13
53
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( )
!
The optimum percentage miter M that guaranteed a: VSWR less than
or equal to 1.1 was faund ta be the one g1ven by:
-1.35w M = 52 + 65.e 11
- ~- -.--:r:::~ - .. _ 0
for w/h ~ 0.25 and ,Er ~ 25
It is interesting to note that the optimum miter increases
exponentially with the decrease of line width.
The software tool (FITA2) that was· developed to accurately
reproduce the MIe structure based on this work is presented in Appt:mdix c.
54
'"
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1
(
, 1
\
1 , ,
55
4.3.3. MATCHING NETWORKS
. In order to achieve 500 MHz bandwidtt~ for the overall amplifier,
, ,.t
care was taken dur1ng the design of ,the ',individual modules to make
certain that the bandwidth speciftcation would be met by the amplifier both in the analo9 (saturated mode) and digital application (linear 0, mode) •
However. since the relative -amplifier B.W. is· less than 10~,
simplified matching networks (TT-Type. L-Type t T-Type etc.) were used
for each module during the initial circuit design stage and then
optimized through RFOPT (Compact) across the band (5.9 -~''": 6.4 GHz). <> # ";-..
The optimized roodul es are shown in Appendix 0 -
4.3.3.1 L-TYPE MATCHING NETWORK
An L-Type matching network like the one shown 1n Figure 14 can
have three basic configurations: Q ,
1 - Open stub 1
2 - Short ci rcuit 'stub 3 - Stub terminated with an impedance Z
Fi g. 14
,.
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56
( Graphical solution tt\rough the Smith chart for this Idnd of "
matching structure i5 trivi al as shown in F1g. 15.
The analytical solution 1s given by the following
expressions:
A - Fi rst location of stub (Pl)
Al - Open stub {12 open)
A2.'- Short cir:cuit stub (12 short) - .. ~- - ..
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j
f ' f r t ~
1 ~. ~ f 1
!
'
1 ~ ..
. \
",
", \ . "
?
8 - Second location of stub (P2)
81 - Open Stub (1'2 open)
tan ((3L20> = [{(l -
where:
Analyzing this" structurès frequency response one can verity
that the bandwi dth of the match1 ng network can be reasonab ly wi de •
(lO~) when the < transmission line length invo1ved in the matching
networ~ can-be made smal'. ", >"
For. thi s r~ason the matchi n9 network shoul d start as close as
possible to the FET. If, for instance, "the stub 1engt,h needed is still,
too long, one can place two stubs in para11e1 instead of one. To make
it shorter, the stub impedance can be lI!ade smaller as well. Special
éare shoul d be takeri to keep the stub l ength longer than the li ne
width. otherwise the stub acts as a transformer.
l
"
·f
57
1
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.,Jo.-
(.
j l' 1
' f r
\ , \
<-
r 1
!
1 i
t 1
,( )
'~wr-
# r ZI
Zin l -
,\
L>W
(a) (b)
Figure 16
If, for instance, one side of the parallel stub shown in
figure 16(b) is short c1rcuited and the other side remaini open. this structure will present an interesting characteristic given bel~w.
~--~--
where:
v !vo " Isc
I:J.t = t R, oc - sc
\' ' The effect1veness of this structure can be ~asl1y ver1f1ed for the case loc = lsc = 10/2
"
'II'f y c_ 2Y 0 cotan ---
2fo
f = o
58
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,
~ f
1 1
t
)
!e !
1 l "
1
( i /
Vp = phase velocity of propagat1ng wave
We can see in this case that the characteristic admittance is doubled and the open-short circuit shunt resonates at 10 = ÀgJ4
4.3.3.2 TI - TYPE MATCHING NETWORK
The TI-Type matching network (Fig 17) 1s s ome,ti mes more convenient to achieve matching depending on the fmpedanceqL'
rL. <1
" (Z2
L2 (Za
la
j B3 jBt
r j (Z3 (Z1 1 Zl r' lin
La L1 Zin -=- .....
Figure 17
The graphical solution for this matching network is a1so -trivial as il1ustrated in Figure 18. \
Fig- 18
59
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(
•
4.3.3.3 TRANSMISSION LINE TRANSFORMERS
The short step Chebyshev 'Impedance transformer is used quite often in matching the complex fmpedance of a transistor to a f'lxed 1 i ne impedance. This type of transfonner 15 rather more complicated and its main advantage 15 its large bandwidth wh1ch can be ach1eved when a number of sections are used. However, in many cases li simple
transformer, util1zing a specifie length of transmission line, is much simpler and quite Adequate. Figure 19 shows a real-to -complex
, impedance match.
Transmission Lin. • Z
/'
Figure 19
It 1s known that if Xo .. 0 good matchfng can be achieved when:
and
e 900 L =
60
._1
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() '-'
When Xo ~ 0 the values of ZL and 6l can be obtained from
Z (R - R ) el = arctan[ l pi]
- R.X , 0
The absolute value of the reflection coefficient at any frequency can
be expressed as •
[R;(ZL - XoAtan9) - ZLRo] + j[ZL(XoA + ZLtane ) - R;Rotan9]
[Ri(Z~ - XoAtane) + ZLRo] + j[ZL(XoA + ZLtane ) + R;Rotan9]
since a/el = flfc where f 1s the frequency of interest and fc 1s the center frequency. The reactance multiplier A is a/el for i nducthe reactance and E\/e for capacit1ve reactance.
In many cases this simple structure prov1des excellent
matchi n9 over reasonab ly wi de bandwi dths (l5~) dependi n9 on the impedance that will have to be matched.
61
j
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. l i
f . (.
\' , ;,
. i f ,
f , , -1 , , \
..
" "e
. "'
x
4.3.3.4 MICROSTRIP STEPS
It 1s "nown that various phys1cal d1scont1nu1ties in the center conductor of the strip11ne such as steps, holes and bends will act as reactive structures. These' reactive structures can be used for
,
matching purposes, in other words, semi-lumped elements can be realized in this way to achieve the desired matching •
Figure.20 shows two possible steps and their equivalent'circuits:
/
1 Z1;, Î' Z"
f ' 1 Zl 1
"
Z2 ~ Zl
.'~ --.--t2
"
-..-....--il
jX!2 jX/2 jX
:Bh 1 ,.,..,..,..
1 Q " e T T
, ., -= x '" ZlSINBt1 ~ïl~1î 1
.. IJ; 1 st1 < :;
. . _-_~~ = __ Y2SINBt2 ~ ~2Bt21
' ,'. 'B12 < i-Figure 20 ~ - ~,
.....
------_ .. _----~ - - ___ A.~_. __ _ .
62
0
jB/2
•
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,",
( , i
(
o
. "
, .
As can be seen, a short length of high =Zo line terminated at both ends by a relatively low impedance has an effect equivalent to that of a series inductance having a value of L = ZOl/l.gf (henries), while
----a short lengt~ of lowZo l1netenninated by a relatively high impedance line presents a shunt capacitive effect havfng a value of C= , I(Zo ').gf) (farads)
,----
" ----- -
63
----..!....
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4.3.4 COMPARATIVE STUDY OF A PUSH-PULL CL S AB AMPLIFIER AND OF A
HYBRID COUPLED CLASS A POWER AMPLIFIER.
This comparative study was initia ed for the purpose of determining the suitability of push-pull clas AB power amplifiers for radio 1 ink application.The same devi~e was sed in all the circuits
and the modes of operation analyzed to avoid misinterpretation due to
different device characteristics.
~
The active device used in this exploratory work, the Nippon i
Hi gh Powe.r Fi el d Effect transi stor NEC868296 has agate length of 1 pm
with a total gate wi dth of 2800 }lm, which provi des 29 dBm of output for 22 dBm input power at 6 GHz.
. stage class A and AB high power In designing the single
ampl Hier, the large signal
me<~sur~ at Pin = 22 dBm and Ids s-parameters Sn and 522 were
= 370 ma and 100 ma respectively.
The small-signal S12 and S21 were used in the design
for both the class A and class AB amplifier, since these parameters change very little with the drive level. For class A the devices
operated at Ids = 1/2 Idss and for Class AB operated at Ids = 1/10 Idss approximately. The set-up for the large-signal s-parameter is
shown in Appêndix A •
. "RFOPT II was used for the optimization of the matching networks
across "the band 5.7 to "6.7 GHz. Open stubs combined with microstrip
line transformers was used as matching structures for the amplifiers.
The teflon fiberglass material was used for the matching networks.
64
L
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[
"
\ ! .
)
1 -
4.3.4.1 BIASING CONSIDERATIONS
One of the maJor aspects of microwave power amplifiers design
- ysing FET's is related to the effect of bias point selection on the
ampli fier performance. A group of static DC ,curves from which the
designer chooses a quiescent operating point is ,given in Figure 21.
~------_______ Idss [Vgs:OV]
Vgs = - 2Y
O.1Idss'
Fi gure 21 ,
Points A to 0 correspond to four typirn- applications of GaAs
FET's: low noise. low power (Al" low noise. higher gain (B). class A
power (C) and class AB power (0).
In this exploratory work a comparative study is made between
hybrid coupled class A (point Cl biased in two different modes,
cons tant gate val tage and constant drain cu rrent wf th the push pull
class AB {point Dl biased with constant gate voltage.
.1
\
65
,-
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1
1 1
J
1 J i
Î - -- --9,-
A s1mplif1ed circuit diagram for the two bias modes used is given in Figure 22.
,
Conat. Conat. (a) Gate Voltage (b) Dra; n Current
F1gure 22 .,
An efficient way of supplying DC bias to the FEl over 500 MHz bandw~dth was needed. Since th~ biasïng must be achieved w1th minimum RF energy 10ss to the bias line and DC source. the input impedance of the bi as H ne shoul d be an RF open-ci rcuit over the entire bandw1 dth. Bias lines constructed us1ng lumped elements or sections of quarter wavelength transmission line have narrow bandwidth and therefore are i nappropr1 ate.
66
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"
1
L
To meet this broad bandwidth requirement, the circular stub or uhalf-moon u structure (Fig. 23), plus a high impedance quarter wavelength transformer was used. (Fig. 23)
-, ~.
C
A 1 1 1
50n LINE
1 1 AI
Figure 23
The input; -impedance of an open circuited microstr1p radial element with vertex angle ~ radians 1s ( 54 ) •
, given by
. hZo(Kri)cos[e(Kri ) - ~(KrL)] ~ ZT = J
ar;sj"[1/J(Kri ) - 1/J(KrL)] (rt~inner radius)
where Zo 1 s the wave impedance of a rad; al transm; 5S; on li ne
67 \
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c
J
()
wheré: Jn is the Bissel function of the first kind of order n
Nn is the Bessel function of the second kind of order n
Er - dielectric permittivity )
1/2 k = W(ErEo~o) is the wave number
a = "half moon" angle
e(Kr) = ~rctan [No(Kr) / Jo(Kr)]
V(lI} = arctan [-J l (Kr) / Nl (K~)]
~ , (tige effects of the microstrip elements which are not taken
,
into account by this model can be approximat~ly taken into account by \
replacing the relative pennittivity Er by the effective permittivity
~eff.
The resulting reflection coefficient rA at the plane A-A'
can 'show unit magni tude and 180· phase, arg ( rA ), at the center
frequency of operation fo by choosing the proper radi~s ri • The
angle oIJ is detennined from bandwidth requirements. 'As the angle ~
i s made bi gger the vari ation of arg ( rA ) with frequency i s reduc.ed.
68
" - ~
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69 !,l
", This bi as ' structure was optimized through "RFOPT u and the o
resu1ting optimized dimensions are shown .in Figure 24 along with thê o
computer print out. ~ {
ll~ Variable (1)- to be opt1mized ~
12= Variable (2) to be optimized p
CIRCUIT OPTIHIZATION IJITH - 2 VARIA~LES ~
• INITIAL CIRCUI'T ANALYSIS
INPUT REFl.p COEF. ANI' VSWR IN SO. OHM SYSTEM IJITH 50.(1 OHM Lor.I'
FCMH:!'. RHO C I1AGN. <A'lGLE:) VSWR RrT L/G (f.r.) z (r~+JX) m:t1
THE ~CST [O[,E -IS AN APPROXIMATION < •
5S00.000 .996 ' 6.9 -4-49.35:1 -.04 30.02 829.91-5900.00(\ .999 2.4 3226.51:1 _°.01 34.01 234J.43 k15O • OQO 1.000 -.3 -.00 .46 -21197.:'1 6400.000 .999 -3.0 3650.42
0: 1 -.00 20 ... 0 -1928.83
6Bp-0.OOO - .996 -7.3 549.88:1 -.03 22.3" -781.8:;
OPTIMl~AT'ION IrEGINS &.lITA FDLlOUING VARIAIrlES ANlt GRAr'IENTS .
VARJAIclES 'L (1): 372.00 ml,·
C :!): 220.18 mil. ERR. .r • .& !' 1. 00.. .
----tt't:~----
GRAItIENTS C 1): • 77133f-O" ( 2): -.94456[-02
HOW MANY ITERATJONS ~EFORE NEXT STOP '0' RESUlTS_ IN FINAL ANALYSIS. LIANT INTEF"I1EI'JATE PRINTS (YES=1,NCl=0)1 TYPE TIJO IIUI1r:ERS: I.J l' 2,0
( 1): . 371.9ft ( 1): .42977r:-03 ÇlI '1 (:!>: 22'.S6 (2): .30671E-(\2
ERR. f'. z: 1 • 00,( -;---t.t*t----'
GRAI'IENT TERHItMTJON
!iSOO.O"o 5900.000 1-1S0.000 6400.000
) 6"800 .. 000
" -
.996 1.000 1.000 .999 .996
LlIT:' A JiOVE VALUES. rIlIAL
. 1..S 520.08:1 2.3 -....
-3.2 2598.28:1 -7.4 ~BO.81-:1
Fig. 24
AtIAl YSIS FOLlOWS l'j
",;
'" -.03 27.4t· 8-43.16 -.00 24.~9 24~::;.71
-.00 'l 40.:!i. -15397.63 -.02 26.8'- o-186th~5
-.04 24~90 -771.79
(1
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, l, (
i 1
1
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..
.".,
4.3.4.2 CIRCUIT ANAlYSIS _
The hybrid coupled class A power amplifier 1s. shown 1né
figure
25 and consists of two identical s1ngle-ended gain stages, 1ncluding
input and output matc~ing networks connected together with two 3 dB / quadrature hybr1~s.
IIp Hybrid
IDENTICAL GAIN STAGES WITH PARAMETERS
[ Sll Sl2]
S21 S22
Fi gure 25
OIT.! Hybrid
500
OUT '---. ~
/
The design of the hybrid coupled class A power amplifier was t.t,~,"::;:
then d1v1ded into three tasks:
1/ Design and optim1zatton of the quadrature hybr1ds.
2J Design an~ opt1mization of the single ended gain"modules.
3/ 1 ntegrat1 on of the s 1 ngl e ended ampli fi er with the i np'ut
and output hybr1ds. (
70
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~. ".
~ 1'1. ••
le,. , '00
~.' CI ('., ", .. : . ' •• ~ , 1
.. : ~,' . . f ,;' . , .~o\~, (
of., " \t.
o .t o i'
o •
,
!--
The ~brid schematic and MIe real1zation is shown in
Figure 26 •
Zo A.81/90·+9
CD ,...----, ... ---' ® . "
0' ~'~'
B.a11..!!:. ®
son
Figure ?6
Ideally" there would be equal power division) perfect phase quadrature between the, two outputs' and rio internal losses or
reflections. 1 n tenns of fi gure 26 thi s woul cl imply A = B_= 1/-/2 ,1
6= 0 and complete isolation.
châractéf'l sti cs:'
" \ The MIC ~bfid presented at 6.2 GHz the ~11owi~9 '\ _.
'r,.
Coup1.1 n9 Factor· C2 1: C4 D 3.4 dB Retum loss @ port 1 1: 25 dB
Isolation = 22 dB., ...
1 ..
71
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, J
t t \
c'
. ,
..
72
(
The S-parameters of the overall amp1i\fier can be written as follows:
rJ
,
Assuming the FE-Ts ident1cal·· and the hybrids ideal woul d
resu1 t: lb . \ / 511 / :: / S22 / :: 0 ! .'"
or /
1 521 / IS 21 (A) 1 &: IS 21 {B)/ .
= ...
/5 l2 / :: '/S12(A)1 = / S 12 (B >1
e
The optimfzed circuit on teflon fiberglass (ér :: 2.2, h = .031 11) is
shown in Fi gure 27 •
;, --........ ~- -
Figure 27
1 ---"-
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f 1
! J t f,
1 , 1
f
-,
The push-pull class AB power amplifier 1S shown in Figure 28,
consisting of two identical single endèd qain stages f.ed 180· out of phase through a RAT RACE coupler and having, 1ts output combined
through another RAT-RACE coupler. \ ,"
f u IN
INPUT RAT-RACE IDENTICAL GAIN STAGES OUTPUT RAT-RACE ....:.;.---
~--- - ~ ....
Figure 28 •
The MIC realization of the RAT-RACE coupl er 1s shown in Fi gure 29. ,
0 " 0 )
é, Zo A.a~9o·+e
o Zr
o Zo
Figure 29
--~-----'--------~~-----
73
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1 K ,
~ , 1 ,
1
1 )
( ')
l t was found experimentally that the optimum perfonnance for ~
the RAT-RACE coupl er was achieved when the impedance of the ri ng was 5
t above the nominal i.e. Zr=1.0sJ2.Zo.
Ideally there would be equal power division, perfect 180· "-phase sh1ft,· between ports 2 and 4 and no internal losses or '.'
reflections, thfs could imply A = B = 1f/2 ,fJ • 0 and
complete i solaUon. \
The MIC RAT RACE presented at 6.2 GHz the fo1lowing
character1 stics.
Coupling factor '= C2 = C 4 = 3.3 dB
Return loss @ port 1 greater than 24 dB
1 solat1on = 25 dB
The S- parameter magnitudes of the overall amplifier ar~~ the same as
the ones given for the overa 11 hybri d coup 1 ed amp l1ifi er wi th the additional advantage of eancelling out the second harmonie as
discussed further.
The optimized circuit for the push pull class AB amplifier on teflon
fiberglass (E'r = 2.2, h • .031") 15 shown in Figure 30.
Figure 3q
74
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t - ,
t.
il i' 1 1
j'
>.
AMPLIFIER PERFORMANCE
Two prototypes based on ~he computer optimized ~1rcu1ts for the hybri d coupl e~. cl ass A and push-pull .cl ass AB were bui lt and 'Optimi zed 'on the bench'. for maximum power output and then c~aracterizeë:l with
. respect to bandwidth, fnput- pow~r vs output powel". drain current vs
input po~er, efficiency vs ~nput power-'and AM to ~M conversion. The measurement arrangement utilited is shown in appendix E. The measured
results are shown in Figures 31 ,32, 33, 33, 34 and 35 respectively.
'" With respect to bandw1dth, a sl1ght'reduction was observed in the
push-pull class AB configuration in comparison with the nybrid coupled class A amplifier. For this class A ampHfier the same bandwidth was obtained in the two bias modes configuration. Higher power output can be obtained from a class A FET power amplifier whenlds 15 maintained constant instead of Vgs constant but a deteriorat1on in the AM/PM convers; on becomes very si gnif; cant after 22 dBm input power.
The same power output (31.5 dBm) for 25 dBm input drive level was ,measured for the push-pull class AB and constant current class A
" hybrid coupled amplifier, but- as was expected the best
effi ci ency /power compromi se was obtai ned ~;for the push-pull cl ass AB power amplifier. The efficiency 1s improv&d from'19% Cclass A) to 35~ (class AB) with a sacrifice of AM to PM conversion that reaches a peak of 6.5°/db, which 1s 1° worse than that of a hybrid coupled class A
(const. current mode), and 3.5 0 worse than that of a hybr!d coupled class A (VGS constant mode). At the same time it is interesting to note that the phase distortion on the push-pull class AB configuration i s better than the other. two modes of operation up to 23 dBm input power level (29.7 dBm O/P) and Just then it deteriorates much faster.
75
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1 , 1 1 l,
\
>
J
(
( )
On the other hand, the second order hannon1c iS-:...lower for the push pull class AB: -55 dBc was measured for the push-pull and -45 dBc was measured "for the hybrid coupled class A (const VGS mode and
constant Ids mode). These measurements· were made at 25 dBm RF drive level. The push-pull class AB amplifier offered the highest saturated output power but the linear gain was about 0.5 dB worse than ~hat obtafned by the class A.
From this co~parative study the following conclusions are reached:
1 - The parallel hybrid coupled class A amplifier configuration is the one that offers the minimum AM to PM distortion with the drawback of sacrificing the amplifier efficiency, 2nd order harmonie
response and power output. Incl ass A operation, constant drai n current bias mode, as in the one used, 1s preferable to constant
gate bias mode, if power output is the main concerne When AM to PM
requirements are very tight the 'constant gate voltage mode s~ould b~ chosen instead.
2- The pu~.-pull class AB amplifier ;s an obvious choice when efficiency is the most important parameter. In addition it offers
'sl1ghtly higher output power and 10\iier 2nd order harmonie response.
Hybrid coupled class A constant current mode of operation was chosen to be used in the power stage for the power amplifier line-up due to the fact that for this particular ampl ifier the bi/PM conversion specified at -10 dB AM/PM point is 4 degrees maximum, and this would be much more diff1cult to achieve with the push-pull configuration
based on the results obtained from this exploratory work.
76
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,. :r
" 1
f
1
! \ 1
f . I,W.
(dB.
31
30
2t
21
24
22
20
, "
~ Puah-Pull Ch •• AI
e e Parall.l Coupl.d Cla •• À
5~ U U AD 82 8A sa U fMC! (OHz)
Fig.3i - Amplifier Bandwidth
10 12
Fig.32
14 18
--6-- Pu.h-Pull Cl ....... ---- Parallel Cl ••• 4
C (v.. CODat.) ~ Parallel Cl ••• 4
(Ida Cout.)
" 20 22 24 28
Output Power vs Input Power
77
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•
•
()
,. 2.
;
--..- Pusb-Pull Cl •• s AB --e- Pal'.11.1 Cl ••• > A
•• (V,. COllat..)
Pan (dam)
Fig.33 Drain Current vs Input Power
- Pusb-Pull Cl ••• AB ----- 'ar.11.1 Cl ••• A
('f,. COllat.) --e-- '.1'811.1 Cl ••• A
" (Id. CODat.)
,. 21
Fig .~34 Efficiency vs Input Power
'6ft (dBIft)
78
,;/
• 0
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i , 1 '. , 1 i !,
[ l t, ~ f' <'
t r 1 , i 1
1 1
1
-2
n
.'
- ... - Push-Pull Class AB -ee- Parallel Clasa A
(VS. CODS t.) --Oe-- Parallel Cla.s A
(Ida Coust.)
tO 14 t' 22
-Fig.35 - Phase Distortion vs Input Power
79
Phi . (dBm)
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-( )
4.4 FINAL P.A. LINE-UP AND RESULTS
To reaeh the fi nal P .A. li ne-up the following tasks had to be
accomp li shed: ------ -
1 - Deviee evaluation
2 - Modul e optimizati on
3 - Intègration
To evaluate the deviees, matehing networks (L-type, n-type etc)
were initial1y designed using the Smith chart based on the small
signal s-parametersfrom the manufacturer data-sheet and then optimized
for gain (MSC~8000, MSC88001, MSC88002, MSC88004) and power (MSC88012
and NEC868898-6).
Every device was then evaluated in its own optimized test jig such as the one represented in Fi gure 36.
/~--------------~
Fi g. 36
80
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-.,. (
\
After the evaluation of the devices with the results summarized as in Table III, the following decisions were tak:en:
- MSC 88000 was removed from use in the input module because it
did not meet the specifications for gain and bandwidth and the MSC 88001 was substituted for it.
- MSC 88012 was removed from use in the output module because in addition to not being able to meet the requirements for linear gain and power output it 1s ~an extremely difficult device to obtain the optimum matching network:" due mainly to its extremely
} low input impedance. The NEC868898-6 was used instead.
A fter the <tevi ces had been eva l uated and wi th the modi fi èat ions shown before, the input, driver and output modules were then designed . ~
fo1lowing the standard procedure shown previously and then tested individually be,fore integration. the input and driver modules were optimized for maximum gain, and thJ;! output module was optimized for maximum power output. The optimized modules are shown in Appendix D.
The final line up is shown in figure 37. A drop in isolator (M/A 54446) was inserted in front of the input module to guarantee an input return 10ss better than 9 dB. Between the driver and output module it was not necessary t~ use a "drop i n" isolator to faci1i tate integration since the 3 dB hybrid coupler provides a good input return 10ss (greater than 20 dB) for the output module. To meet the output return 10ss an isodaptor (SMA coax to waveguide) was specified.
'1'"
81
_1
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r~l
MSC88000 (Vds ': 8.0 V I..ds =80 MA)
1 i.
MCS~Ol (Vds :: 8.0 V Ille = 8- MA) MCS88002 (Vds = H.U V Ide: = 15- MA) MSC88004 (Vds = 9.0 V ILle: ': 300 MA) MSC88012 (Vds = 10 V . Ide = lA)
NEC868898-6 " (Vds :: 10 V
lds :: lA) . "
----~--... - ~ .. • _r ....... _:;!,_ -~r
Gain @ 1 dB Power @ 1 dB B.W. 1>
Ripple C.P. (dB) C.P. (dBm) (MHz) (dB)
MSG SPEC. MEAS. SPEC. MEAS. SPEC. MEAS. MEAS.
11.0 9.0 8.0 18 12 500 300 2 .
., ,
11.9 8.0 9-:5 24.3 22.4 500 800 .5
. lJ.l H.U 9.U 'l.7 • 'l. 26.2 500 750 " .4
"
13.7 8.0 8.2 29.8 29.0 500 600 "" .5 "
7.8 6.8 5.2 35.2 34.6 500 500 1
, ,-
11.9 6.5 6.3 34.8' 34.7 500 600 .5 0
1 ~ ,
Table III - Deviees Evaluation
!
-~---
'YI!@. 1 dB-C.P.
MEAS.
2.5 1
' i
26.4
33.3
25.0
20.1
22
-
"...,...., )
"
<Xl N
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• , ,. 1 i
1-
--.. 1
...J
INPUT MODULE /"
TCHPEllATUII.Jl COHPESATIOH MODULE
."~-'''''''--~- - - ~-v
DUVER HODULE OUTPUT MODULE COUPLER
~--~-----------i---------_·_·_--~----------------'-----------~----------------,------------, '1 '2 '3 14 '5 1 : , l ' : t
~. \-
, ' 1 ~,I l , 1 MIe HEC : RF mon.
, tOOA .-. -~. LJ il
, 1 1 1 1
'---------
'-
1 1 ..
s c ,
--------~-------~~-
> ~ ,
~
1 o ~
c > a ,
POWER CONDITIONEA
~
} ".6 ~
+
~ • ,
oc E
1 ft
oC
1-t _L __ _
1 z ft
1 o ., ..
> > > > ::! .. ~ .. , ,
Fiaure 37 - FINAL P .A. - LIME-UP
r-
.' ~~
c :J
c :l
:. > > :. • 0 • 0 ..... ." .. , 1
o
MAc: OUT
1
---_.'- c-------
OC mon.
"
HODULBS l, land 4 - SBB APPJtRDXlr D
HODULJt8 2 and li - 'ln: pa. 14
l'
)~
CD W
X-
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l 1
\ 1 \ t
1 l
1
r ~
The temperatUre compensation module* shown in "ffgure 38 presents
the minimum in~ertion 10s5 (-3.2 dB) when the' pin diode is Qiased at
30 ma. This module presented two major drawbacks. Ffr5tly the . l '
excessive minimum insertion 10ss 15 no~ desfrable and secondly the - excessive slope of the insertion 10ss across' the band(5.9 -6.4 GHz),
wh en the pin diode fs biased at lower current to achieve higher
attenuation, is very diff1cult ta compensate. The pin diode attenuator
was integrated to the ampl1ffer biased for minfmum insertion 10ss.
;; " 0" \.,
'.
,IN OUT
.----r 1
) f
1
\ Figure, 3~
J
The RF and OC power monitoring i s made through the ~IC coupler
placed after the output stage. T~e~coup'er fs shown in f1gure 39.
IN
RF mon. r-----------~~--~~----~ --'-
OUT ..
\ ~ .
~ Supplfed by Bell NO/~lIern Research (BNR) <
, . .r ,_
"
i .... ~~/ 1
84
.,
6
~ f
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"
r ( Every module has its own power conditioning board, simplifyfng
fntegratfon and maintenance process. The voltages are fed into the mo"dules through fil tercons (1500 pF) with ferrite beads to provi de
resfstive 1 oadi n9 at low frequencf es and ensure s tabil f ty at low frequenci~s.
In des1gning the carrier plate for each module, special care was
talcen to prevent non-TEM modes from propagating laterally an"d causing ~ unwanted feedbaclc and instabil1ties. For that reason the carrier wfdth
was chosen to be less than ~/4 for the h1ghest operatfng frequency (6.45 GHz).
.. -
The carrier plates were made of brass and the housing of aluminum. The housing was desfgned to accomodate a 10 watt amplifier as well and . talces i nto account the fact that i t woul d be located on the top of an equfpment baye The housing'including the heat sfnk has approximately the followf ng dimensions: 1 = 33 cm, w = 16 cm. h = 11 cm.
Table IV shows the measured results for this amplifier. Followfng integration into the amplifier housing. sorne additional tuning was requfred to achfeve the overall amplifier specifications. This is due primarily to the small mfsmatches introduced by the flexible module
interconnections and due to evanescent radiation modes which tend to degrade performance by mutual coupl i ng between modules.
85
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86
POWER N'S> SPEC REgUIREMENT ACTUAL MEASUREMENT -- -
Frequency Range 5925-6425 MHz Saturated!5800-6550
Linear:5900-6475
Minimum -lOdB AM/AM Pout 37 dBm
Maximum-lO dB AM/AM Pout 40 dBm 38.2 dBm ~
r Small Signal Gain 42 + 3 dB 46.0 dB -•
< Small Signal Gain + 0.5 dB -1.2 dB (without
Stabil ity with time & tempo compensation)
temperature
"Pout @ -lOdB AM/AMI! + 0.5 dB -.3 dB(25 to 7d·C) - ; s tabi li ty wi th time &
-~emperature
• ~ 1
AM/PM @ "-10 dB AM/AM 4 deg/dB 1.8/dB worst case
Pout" ,
Noise Figure ~
than 7 dB 8 dB -Less
,
~
"t
o lEQUI1BHBNT VS MEASURED RESULTS
TABLE IV c_
._-- _ __ 1 \f-
Q /
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~ \ \ 1
1
1 ,1 i
, . "
1
1
Output return .1 oss 26 dB Not: Hea.sured
l
I "put return loss 9 dB . 20 dB -,
Sma11 signal slope 0.25 dB /30 MHz 0.2 dB/30 MHz ,
Saturated slope 0.1, dB /30 MHz 0.1 dB/30 MHz
" , ,
Harmoni c O"utput -20 dBc , -40 dBc . 5 Watt PA Power .
,
Dissipation 40 watt 36 'w -
Output Moni tor -30 dBc + 2 dB -33.5 dBc @ 38.2 dBm -( ca11br:ated)
,
-Output Detector -.5 volt @ + 30 dBm -0.75~V @ + 38 dBm
MTBF at T = 2S·C 200,000 hrs.
1
REQUIREMENT vs MEASURED RESULTS
TABLE IV (CONT.)
, 1
8'7 .
~.
~
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t \ \ 1 f f l
(
o
•
5. CONCLUSIONS
From the results presented on Table IV we see that exéellent
performance was achieved 'with the. solid state FET power amplifier, and
the main requirements for analog and digital radio transmission
application were met.
In particular, this FET power amplifier presents the fol1ow-ing
advantages over any ex; stfng comparabl e travell ing wave tube
amp 1 Hi èl"rs :
- LOWER NOISE FIG~RE, 8 dB was achieved with this amplifier
compared to approximately 25 dB for the T .W. T.
HIGHER LI NEARITY, for digital application this amplffier has to
be backed off only 5 dB wh; l e any ex; sti n9 T. W. T. woul d have ta
be backed off at least 8 dB due to its inherent non-linear characteristic. "
- SMALLER SIZE, this amplifier is 20" smaller in size than a
T~W.T. including power supply and heatsink and requires 10w
supply vOltages(+ 10.5 V and -5 V) compared with a few kilovolts
for i ts tube counterpart.
- HIGHER RELIABILITY, 400,000 hrs MTBF is expected for. this
amplifier whfle the state of the art TWT has ~n estimated MTBF of 200,000 hrs.
- LOWER CO ST • this ampl Hier has an estimated cost of $2000.00
while its tube counterpart acost is $3500.00.
,
88 l'
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: i 1
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) . ..
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, Based on these advantages and the flct that the relative cost of a
travelling wave tube amplifier on radio linlc transponders °1s
approximately 30' of the transponder cost makes the FEl power ., amplifier an appropriate substitute and improvement over exist1ng
travelling wave tube ampl1fiers •.
The computer aided design t90ls RFOPT <Compact Engineering) and •
the FITA-2 developed during this project were used extens1vely in the
development of the amplifier and proved to be very efficient aids in
reducing substantially the fine tuning necess~.
Thics amplifier can be improved still further and the two maJor
a reas 1 eft for improvemerft are:
1 - Substitute a dual gate FET for the first stage and achieve
temperature compensation by varying the gain of this stage
instead.of using the pin diode attenuator.
Advantages:
- reduce the number of modules
very f'l at frequency response
2 - Substitute for the power stage a linearized push-pull' class AB
power ampl Hier. This possibility 'lias left open for further
investigation .. The author bel i'eves that the use of other
biasing circuits such as a broadband emitter follower plus an
optimization of the output matching network for the lowest
thi rd order i ntennodul ation product and maximum power , s1multaneously. would malee the push-pull class AB pow~r
amplifier feasible for radio l1nk applications.
89
--'"
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APPENDIX A - S-PARAMETER MEASURH1E!lT
, 1 n this appendix. the S-parameter arrangements for measuring
SU, \ 522 (large signal), 'and S12~ S21 (small signal) 1s pre'sented and the measured resul ts are shown. •
. obs: The measured results 1s nct an average result of maAy'devices but
just a single device measured result.
( ~
\
FREQ. . COUNTER
,
.
POWER SUPPLY
D.U.T
SWEEP 'REFL TRANS GENERATORr-_____________________________ ~ TEST UNIT
HP8620 '" HP8743A DIRECTIONAL
. COUPLER
..
Flg.A1
REF TEST CONVERTER
NETWORK ANALYZER HP8410
SKALL SIGNAL S-PAAAMETER MEASUREMENT ARRANGEMENT·
•
A1
•
, !
i \ ,
,
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,
,-SWEEP ,
GENERATOR HP 8620 ,
.~.,4> . . , .
~ . ..
LINEAR P. A. r
. -.."
1 3d! l . 1 l'
. ~
, ,
. ,
.
.
. .'
,
, \
1 ~ . ~ 4
- .. , J
. ,
Il
REFLECTION UNIT -
HP 8742A
J ;. 110 dB 1
20dB 120dB
1 REF TESy
CONVERTER .'
NETWORK . ANALYZER
HP8410
1
1
>;
Figure A.2
" .
.
. .
POWER SUPPLY
.
,~ ,
'D. q,.dT.'
.
-,
LARGE SIGNAL S-PARAMgTER ARRANGEMENT
-.t',
. .
.
, .
~
'\
._-~,
,
HPOWERl-PAD
"
V
: 4
-,
,
.
.
POW'ER METER -
.
6 1
.
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" ~ ,
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" 1-.J,
.~
Devie, MSCatlOOl Vds· 9;0 volts, 1ds ~ 100 "" A4 ~ • •
# ' Vgs• 2..§5 volts .
(~ , Freq. Sl1 521 S12, S22 (GHz] Mag Phase Mag Phase Mag Phase Mag Phase
, 1
I,t 5.7 .92 -148· 1.3 sé· .064 5· .81 -114·
, t,
, 5.9 .90 -153· 1.3 56· .064 4· .82 -117·
6.1 '.92 -155·"" 1.4 54· .067 3· . .83 -117·
:'158· 48· 3° . ,
-119· 6.3 .90 -1.3 .065 .84
6.5 .87 -163° 1.4 44° .067 2· .81 _126° 1
, 6.7 .84 -171· 1.45 39· .070 1° .78 -123° '" , 1
0
D-evice - MSC88002 Vds = 9.0 volts, ~ds = 200 'mA Vgs ' = 0.21
.. volts ,.,
Freq. Sl1 S21 S12 S22
(GHz) Mag Phase Mag Phase Mag Phase Mag Phase
.. 5.7 .90 175 0 1.8 44 11
" .039 28° .65 _129 D
5.9 .88 112" 1.8 42" .039 30° .66 -132' ,
6.1 .88 168" 1.9 39° .040 30· .68 _132° ~
6.3 .85 163" 1.8 ( 340 .042 32 0 .68 -134' "
"
6.5 .83 155" 1.8 29' .042 .28' .66 _135°
'~
6.7 .82 156 0 1.7 25° Ji};t .042 26· .64 _140'
0 , ,)
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...
;;lI Deviee - MSC880~ Yds· 9.0 volts, Ids • 400 mA AS
" Ygs• 2.15 volts
(. Freq. S11 $l1 S12
LO
S22
, (GHz) Mag Phase Mag Phase Mag Phas~ Mag Phase
t 5.7 .92 168· 1.3 37· .046 30· .62 -153·
/IP 5.9 .92 164· 1.3 35· .048 30· .63 -156· ,
, !
6.1 .93 163· 1.4 30· .050 28· .64 -156·
; , 157· 26· 30· -158· 6.3 .• 90 1.3 .052, .62
6.5 .89 151· . 1.3 20· .052 26· .60 -162°'
/y ,
6.7 .88 142· 1.3 ' 16° .052 24· .58 -169· 1 4
f'
Deviee NEC8682 Vds = 9.0 volts, Ids" 275 mA Ygs .. 2.0 volts
~~. t , '
Freq S11 521 ·SIZ· o SZ2
(GHz) Mag Phase Mag Phase Mag 'Phase ' Mag Phase .. '"
S".7 .75 122° 1.3 Z· .07 15° .55 -lSO·
1
. j 6.2 .56 87· tJ 1.6 -19· .08 O· , .60 178·
6.7 .46 20· 1.6 -42° .08 -12· .60 1650
"
<:!> .' ...
. i 1
1 1 • 1
J
t \
(~) ,...-/
. , \ . ""'i
'v k !
1 j
._---,-_.-- ----- - "",--.. _ .. __ l .. .
..
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---~ --- ---- - -~- -
" , ' "
~,
• f
Deviee '- NEG8682 Vds. 9.0 vol ts AS
Pin JE + 20 dSm
f,
\ ~
, s111 f
Ids 5.J ~Hz 6.2 GHz 6.7 GHz (MA)
~
10 ( .9 162- .9 155- .8 148-..
. .,.. . 100 .8 148-
, ,.7' 140- , .5 125- &.
f •
/.
200 .71 132- .5 118- .22 85-. ..
300 .,7 120· .5 85- .4 15-
Ir . .
t(t?
,,~
, , 1 S22' 1
Ids 5.7 GHz 6.2 GHz 6.7 GHz hM) J '
~
10 .82 172- .9 ' ,,164- ' .8 152-
100 .75 173- '.8 165· .52 15S·
, 200 .45 '160- .65 155- .60 145·
J
, 300 .55 162- .55 160· .53 150·
The devf ces. MSC88012 and the internally matched deviee. 1
NEC868898 ~ere not character1~ed.
i',
0
. (
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• \ )
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~~~---,---
J
APPENDIX B - RFOPT AND STABILITY ANALYSIS
1 - RFOPT (Comp~ct) (75)
RFOP.T 1s an optimization program ideal for des~9ning
microwave circu1ts)deve1oped by Compact Engineering lnc.
This program uses a gradient optirnization technique which , . computes the result of changing each var; able element on the overa11 error function which is defined by the, ·user. The error
function is the weighted difference between the calculated circuit responsè and speciffed target response and may be
'" specified to place more emphasis on certain parameters such as gain, noise fi~ure, and less on others suc~ as input and output retur~ """1 oss. The magni tu de of the gradi ent prov; des a measure of the sensitivity of the various e1ements to the overa1l circuit response and can be used by the desi9'1er ta reduce aptimization
'" time and hel p insure a reproducl ble desi gn. The program accepts active and' passive elements as well as complex interconnections.
The three'rnain functions of RFOPT are:
1 Circuit.analysis and stability analysi s
2 Sensitivity analysis
3 Op'timization (up to 15 variables)
This program was intensively used for the stabflfty analysis of the Field Effect Transistors and for the optimization of vthe matching networks of the ampl i fier modules.
The sta~i1ity analysis result of some of the FErs used 1n the power ampli fier i s shown below.
81
-'
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_pqr ...... M %... S" ..... ____ ~ _ • . ,.......,..- __ l'tll!~'f'q~
........ -' ~
.. DEVICE ; MSC88002
PO~A~ s-rAR~HCTERS IN 5(\.0 OHM ~YST~H r:-REO. 511 S:!1 512. ~22 521 1:
C"~GN~AHGL) ( HAGN<ANGL) ( H~GN<AHGL) (HAGN<ANGl) nll rACT. . 500.(\0 8.,· ... . .. ... -78 3.11< 130.1 .O1~< -i3.:i .32< -7~ lO.l-i 2.2S
1000.00 .86< -90 3.10< 11 S. 0 .046< J-i.O .38< -7<f 9.~J .60 ZOOO.OO .88< -112 1.80< 96.0 .07:!< 16.0 .-i5< -93 9.9-i .2' 301:'0.00 .90< -13-4 =.50< 7~.0 .(19(1< -~.O .S2< -11:'7 7.96 .0' ;<
-iOOO.OO .86< -15-i 2.1-i< 55.0 .095< -19.0 .52< -117 6.61 .2' S500.00 .81<: 17-4 1.75< 28.2 .088< -3~.' .-i6< -138 -i.S" .:'0
.10""0. (\0 .83< 163 1.63< 16.0 '.080< --43.0 .-46< -1"7 -4.2-i .73 . (,5(\(\ 100 .OS< 1 r ., .J_ 1.56< 6.4 .081< -49.0 .-46< -156 3.88 .~,
71:'('(\.(\(' • PS" 1~:! 1.49': -1.0 .081.-:' -~4.0 .-i8~ -166 3.-i6 .~2 8(\00. ("Ct" .92< 1:!7 1.:!S< -1~.0 .067< -63.0 ·.SS{ 177 ,;!..1 .9-4 ....
o ...
ST~8ILITY CIRClC LOCATIONS
t-------IUPUT PLANE-------~ *------OUTPUT PLANE-------* GHAX· r LOCATION STA~LE LClCATION STAEtLE OF\ MSG
11HZ M,~G~ M·IG~ r::'Pll1S r,'EG ION MAGN ANGL R~Drus r,EGION (Dr.~
500.0 l .20< 79 • : 0 OllTSlflE ., ..,..,,, C:! 1. 13 (lUTSI [Ir 16. J - . - .
1000.0 1. 1" < 93 :': 1 OUTSUI( ~.33-: 101 lo5~ (lUTSIDC 18.3 2000.0 !.11< 117 .29 OUTS HIE :'.14< 1:!1 1.1oS OUTSI [lE JS.S' 3000.0 1.('9( 1-1! .36 OUTSlr.E ':.01/ 1<4(' 1.66 OllTSl Dr 14. J .,C'oo.o 1.14< ,~! .33 OUTSII'E l • 98' 1 .. ;' 1. -44 OllTS J f'E 13.~ 5500.0 1.20(-17(\ .., ..
'-' OUTS IllE ~.OO< l!i7 1.16 OUTSI]IE l~.O 6000.1J l . 1 7 < - ! l> (. .,~ ourSIN: 1.93-: 16S 1.07 OUTSlf'C 13.1 '--65"~. 0 1.!3<-148 · :! c· OLll SI rl~ 1.S~~ 17< 1 .0:; (lUl SI fIE 12.' 7000.0 1.(\9(-:-4(\ · l r- OUTSJ rl~ l .6t-/-17~ .97 OUTSI [le Jf" BOoo.O 1.0~<-1:(, " • '3 OUlSIflf 1 • .34~-16;, .71 nUT!; 1 [lE J2.7
CD w
1-
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..... r:..~~.II. .. ~s t!,.~, ... 't'>~j .. ""
,.,"'::· .... 4"ir~~ ... ~I_.' ... -,. ~''''~~.....,....,.,..-- , --------- ~
~
DEVICE : MSC88004
1 POLAR S-PARAHETERS IN 50.0 OH" SYSTEM
F~[Q. Sil S21 SJ~ S~2 S21 ,.. ( HAGH<ANGl. , ( HAGH,"AHGL) ( HAGH<~NGL) (HAGH<ANGL) Il Il FACT.
500.0(· .7-4< -10-4 3.-47< 11-4.9 .O:!1<' 26.9 .23< -112 10.80 2.12 1000. OC .• 77< -117 3.17< 103.0 .• 04~< 20.0 .31< -117 10.02 1.12 ~OOO.OC .79< -136 :!.70< 82.0 .060< 8.0 .~O< -126 9.63 .7-1 3000.00 .81< -155 2.23< 61.0 .07S< --4.0 .-49< -135 6.97 • 18 4000.00 .80< -167 1.90< 43.(\ .(\70::- -14.0 .50< -143 ~.:58 .71 :5500.00 .83< 174 1.48< 18.8 .C~~< -27.1 .49( -162 3.-41 .S4
- 6000~ 00 .86< 166 1.37< 11.0 .O!;8( -32.0 .52< -169 ~.7J .91 6500.00 .87< 159 1.28< 1.2 .0:;9< -l6.0 .5-4< -175 2.1-4 .6' 7000.00 .88< 15~ 1.19< -9.0 .061< -39.0 .56< 17'1 1.:51 .~ .. eooo.ob .89< 140 .99< -:!6.0 .0-49( -45.0 '.57< 169 -.09 ·'1
STABJLJTY Cl~Cl[ LOCATIONS
J-------INPUT PLAHE-------c ~ - -----Ol'TPUT f'LANE-------t GI'1,'\)';. • F' LOCATION STAttLE lOCATIOtJ STAttLE OR ,.,~c:
HH:! hAGN AHGL RAnlUS REGION ftAGH ANGL RADIUS REGION (f'l: )
:500.0 2.33< 10~ .17 OUTSIr'( 3.70-: 113 2.02 OUTSJ{IE' 15.9 1000.(' 1.~7~ 117 .:!~ OllTSll1E ~.7:!: 118 1 .6" (lUTSIfI~ J6.3 ~OOO.(\ 1.:!1< 137 .27 (lUT~ HIE" ~.OS< J30 I.ZO. OUfSl"rl~ 16.5 3000.(1 J.16< 157 .:!8 OUTSIflE 1.0.04< 1.041 .91 QUTSIllE 14.7 -4000.0 1.18~ 169 .2-4 OUTSHIE 1.6"'( 1~1 .77 OUTSIn~ l .: . 3 5:S00.~ 1.1-4<-172 .16 OlITSH'E ]'6~< 169 .69 OUTSIfiE 23.6 6000.0 1.11<-165 • 1 J OUTSIflE J • Sl .... 175 .60 OUTS] [lE 13.7 6!iOO.C' 2. 09('-1 ~8 .13 OUTS HIE 1.~~"'-179 ,57 (IUTSIfiE J:":.3 7000.0 1.09<-1~1 .l~ (lUTSIr.E 1.41 :-l73 .55 ours IllE ":'.~
80(\0.0 1.(\e<-13~ .08 (lUTSI [lE 1."3~-16) .... ) OUTSIl'( l :? • J
CD ....
1 1-
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~' (
APPENDIX C - FIT~2
This appendix describes the program FITA2 deve10ped by the author to ca1culate m1crostrip li ne parameters-; based on the recent works presented in Chapter 4
For a certain micros trip \lne structure as the one shown below:
T
Gi ven the INPUT DATA: (FREE FORMAT)
ZO - Initial Impedance ZINC - Impedance Increment ZFIN - Final Impedance F - Fr e que n c y ( G.B z ) H - Dielectric Thickness EPSR - Material Dielectric Relative Constant T - Copper Thickness DL - Dielectric Loss CL - Cop~er Loss
The program generates a table of:
H
IMPEDANCE vs STRIP WIDTH vs WAVELENGTH vs ATTENUATION/CM vs
QUALITY FACTOR vs MITRE%
OBS: MITRE%, means the amount of microstrip line that should be mitered of a right angle bend to guarantee a VSWR Iess than 1.1
The p~ogramming language used Is FORTRAN IV aed
conversationa1 mode i8 used to ~nter the data
C1
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(')
/
FITA2 BASIC STRUCTURE
CALCULATl
w, :\,ar.,Q,M
L.T.1/2'11'
CALCULATI
W/H
CALCULATI
W, :\, .. ,Q,M
C2
G.T. 1.
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1 ! ,
1 , \
(
APPENDlX D - QPTIMIZED MODULES
This appendix shows the input. driver and output IIOdules
after be1ng optim1zed.
INPUT HODULE
DRIVER MO:OULE
MseaaOOI
OUTPUT HOPULE.
MSCI8OO4
/
use.'OOI
uscaa002
NEC ....
[".n. .17"
raOA l·lI"
.01
1
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.- "'. 1 1 ; ....;
•
, .... ;:~
, 1-
____________ . ..,.._ .2 4.;; "'f'MIi 0&4= QAt il: .~''O .,14(, .l'11' .. 4 .... ;pç;tii!t$l 5!! 1""'~" fo"'t'iM;' lA 4;;'lO!41Q."
..
SWEEP GENERATOR HP8350A
... 1
~OWE~ h 1 SUPPLY
DIRECTIONAL ,COUPLER
/
POWER DETECTOR
SCALAR NETWOR~
ANALYZER PMI038 NIO
SET UP 1
P.A.
~
~
SPECTRUK ANAI.YZER HP8S69A
aGdB
POWER DETECTOR
~
(PIN vs POUT,B.W.,HARHONIC OUTPUT,EPPICIENCY,SMALL SIGNAL GAIN and SMALL SIGNAL SLOPH)
'1
.. ~
> ." ." ,.., :z c .... >< ,..,
i1\ ~ c: ;:0
~ Z -4
-CIl PI -4 1
~ CIl -
m ...
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--------------------------------------------------. , 1
!~ ..
.~
SWEEP GENERATOR HP8350A
..
J
DIRECTIONAL COUPLER
..
<Ii
POWER SUPPLY
PIA.
REl TEST CONVERTER
NETWORK ANALYZER HP8410A
SBT UP· 2 (AM PM MEASUREM&NT)
i ______ _
.......... ....--.-tf:"ll\liO ~ ~,..,.,...... W"f'~ .. '"'--~ ... il"f
DIRECTIONAL .--..-.,. ..... COUPLER
1J---+-JVQ\,.o""'l------t 30 dB
AIR LIRE
'{
"
POWER DETECTOR.
POWER ME'TER
:;;
~ ~.":l'
~.~,
m ~
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REFERENCES
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( 25. H. Morkog, T.J. Drummond , M.Omori t "GaAs I€SFETs by Molecular Bean Epitaxy n, IEEE-Trans on Elect. Dev. volt ED-29, No 2 'pp 222-224, Feb. 1982. \ ,
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72. R.S. Carson,;"High Frequency Amplifiers'" Wl1ey-Interscfence Publfcati on, Joh~ Wfley & Sons Il 1975.
73 "J.V. Dflorenzo and 0.0. Khandelwal in uGaAs FET Principle-. and Technol ogy ll Artech House, Inc., 1982.
74. K.C. Gupta. R. Garg and R.Ghadha, l'Computer Aided Design of Microwa:ve Circufts ll
• Artech House, Inc. 1981. ""
7S.RFOPT user manual,version S.l,August 1979,Compllct
Engineering Inc.
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