chapter 3 experimental set up and...

Chapter 3

Experimental Set up and

Diagnostics

This chapter1 contains the description of parts of experimental set-up designed to

produce plasma using Helicon breakdown technique and different diagnostic tools

used to characterize and observe different physical phenomena in plasma.

3.1 Introduction

The geometrical configuration of the experimental system is an important issue

to be addressed in radio frequency (RF) and microwave experiments. Since the

modes excited in such systems depends on the launched wavelength, the size of the

experimental system must be decided by the wavelength of the launched electro

magnetic(EM) wave. Large sized plasma chambers are also designed to neglect the

boundary effects because the system dimension is much larger than the vacuum

wavelength of the launched EM wave in this case. Experiments in Helicon wave

generated plasmas have been performed in both the linear or cylindrical systems

and the toroidal shaped systems according to the demand of physical phenomena

studied. Since, understanding the mechanisms which determine the confinement of

1 Parts of this chapter are published in:

Physics of Plasmas, 12, June 2005, pp.062510-17

Measurement, Science and Technology, 18, July 2007, pp.2673-2680

35

Chapter 3: Experimental Set-up ... 36

energy and particles in toroidal plasmas, is one of the most prominent and ambigu

ous objectives of fusion research, so the plasma confined in a toroidal magnetic field

as produced by tokamak experiments still retain their importance for the study of

plasma confinement and plasma heating. Moreover, toroidal bounded whistlers or

Helicon wave sustained plasmas in low aspect ratio confinement, show significant

deviations in their physical characteristics which have already been well explored

in cylindrical configurations. An excellent opportunity to study such behavioural

changes in Helicon plasma at very high frequency in such a parameter regime, is

provided by the small aspect ratio toroidal system used here. In our experiment,

antenna design takes care of the normal wave modes of this cavity in such a way so

that effect of boundaries on the wave field pattern could be realized. The integral

toroidal mode number(s) of the wave excited in the very high frequency regime, are

also taken care by the helical antenna designed.

The chapter is organized in the following manner. In section 3.2, the main exper

imental device, pumping system and toroidal magnetic field system are discussed.

Section 3.3 is dedicated to the Radio Frequency (RF) systems such as the RF gen

erators used in our operational regime, transmission lines used and the impedance

matching networks designed and constructed for our experiments. The helical an

tenna design and RF power feeder system design are elucidated in section 3.4. Sec

tion 3.5 contains the elaborated description of the diagnostics designed to probe the

Helicon wave sustained plasma at very high frequency. Essential operational elec

tronic circuits designed for plasma diagnosis are also described in this section. In

section 3.6, the data-acquisition arrangement is mentioned alongwith data transfer

up to the PC. The grounding and shielding schemes of our experimental system are

discussed in section 3.7.

3.2 Experimental Set up

The experimental system is aided with various auxiliary sub-systems. These sub

systems comprise of the main vacuum chamber, vacuum system, the magnetic field

arrangement, RF-power system, different diagnostics, operational electronics, data


acquisition, etc. All these sub-systems are shown together as a block diagram in

figure 3.1.

Figure 3.1: Block diagram of the experimental setup for toroidal Helicon discharge.

3.2.1 The Vacuum chamber

The experimental chamber is a toroidal, stainless steel vacuum vessel with minor

radius(r) of 10.5cm and major radius(R) of 30cm. The vacuum vessel consists of

eight radial ports at four equidistant toroidal positions, each port located at 50°

inclination with the equatorial/horizontal plane. The inner diameter of each port is

9cm and outer diameter for the flanges is 15cm. Locations of antenna and diagnos

tics are indicated in figure 3.2. All the ports of the vacuum chamber are equipped

with conflat (CF) flanges with viton '0'-rings.


-Call

Figure 3.2: Schematic of the experimental setup for Helicon discharge in a conducting toroidal vacuum vessel of small aspect ratio. Radial ports, probe positions, pump ports and helical antenna locations are shown in the figure. Direction of toroidal ambient magnetic field (BT) is also indicated.

3.2.2 Pumping system

The vacuum vessel of volume 68lit. is pumped by two 2" diffusion pumps placed at

90° toroidal locations to the right and left of the antenna, at two additional large

diameter ports. A rotary pump of 200lit.fmin pumping speed is used to back these

diffusion pumps. A base pressure of 1 x w-s mbar is regularly achieved and measured

using Pirani and Penning gauges. The gas feed point is placed diametrically opposite

to the helical antenna location. Discharge is produced primarily using argon gas

filled at a pressure of 2 x w-3 mbar by the aid of a mass flow controller. At some

stages hydrogen gas is also used instead of argon gas at a fill pressure of (8-6) x 10-3

mbar.

3.2.3 Magnetic field system

Magnetic field in plasma systems is essential to excite Helicon waves as well as it

improves plasma confinement. The magnetic field in this system is produced by

four copper cables wound on each of the quadrants of the vacuum vessel. These


coils are connected in parallel to each other to minimize their net resistance. A

pulsed current for 60mB (full width at half maxima) is allowed to flow from a power

supply producing a toroidal magnetic field (B0 ) of 1kGauss(max.) on the axis. For

each plasma shot, the capacitor bank is initially charged to desired value and then

discharged through these coils to achieve required magnetic field value, as indicated

by the calibration figure 3.5. Though, due to the presence of large diameter ports,

there exists a ripple of~ 10% at the outboard side in the toroidal magnetic field,

shown in figure 3.4, the effect on the measured density profile in the intermediate

region between two ports is observed to be negligible.

o.s u..o~ ........... 2~-'-':-4~ ........... 6~-'-':-a ~ ..... 1 ......... o Radial Position (em)

Figure 3.3: A typical radial variation of the toroidal ambient magnetic field measured using radial Hall probe attached with a F.W. BELL Series 9900 Gaussmeter. Geometric centre of the device is located at 5cm.

The toroidal magnetic coil and pulse power supply parameters are given in the table

3.1. During the experiments, at the desired gas pressure, the pulsed magnetic field

triggers the discharge for 50ms, via efficient wave coupling.

3.3 RF System

The RF system of our experimental set-up consists of RF generator, transmission

lines and impedance matching networks. Description of each of these subsystems

are provided in subsequent sections.

Chapter 3: Experimental Set-up ...

i 1.0 ::s ca

(!) .II: -aro.9

2

TowonlaAntenna=o

Port

4 6 8 10 Toroidal Position (em)

40

Figure 3.4: Toroidal/ Axial variation of magnetic field measured near a radial port using axial Hall probe attached with a F.W. BELL Series 9900 Gaussmeter. The port location and antenna position are indicated in the figure.

1. Coil material Enamelled copper (12 mm dia.)

2. No. of layers in coils 5

3. No. of turns in each layer 47

4. Peak current in coils 1.8 kAmpere

5. Capacitor rating 100 mF, 50 V

6. No of capacitors 30

Table 3.1: Specifications of the toroidal ambient magnetic field BT system, designed to produce a peak BT = 1kGauss for ~ 60mS {FWHM).

3.3.1 RF-Generators

RF breakdown in pulsed mode provide immense opportunity to study the discharge

equilibrium related issues. Our Helicon breakdown studies have been discussed in

chapter 3 of this thesis. Our prime requirement of stable and unmodulated output

RF power for requisite duration of plasma discharge is sufficed by a high power RF

generator. It consists of a colpitts type oscillator that operates in the frequency

range of 20M Hz to 40MHz. +2dBm(max) RF power is delivered by the oscillator

stage (Agilent 8648A) to the low power amplifier stage. +47dBm(max.) of RF

power is delivered by the low power amplifier stage (Mini-Circuit LZY- 1) which is

fed to the high power 2kW amplifier stage. The high power amplifier stage consists


1.2

1.0

i 0.8 :I ~ 0.6 .II: --:. 0.4 m

0.2

o.o~ ........................ ........._...........,.o.u.J............._j .......................................... 0 5 10 15 20 25 30 35 40 45

Capacitor Charge (Volts)

41

Figure 3.5: Calibration of the ambient toroidal magnetic field (Br) with charging voltage of capacitors, measured at the centre of antenna port, shown in figure 3.2, using radial Hall probe.

of an air cooled triode, configured in class B grid mode. The plate voltage is varied

in the range of (2- 3.6kV) during our experiment. During the initial breakdown

phase of the discharge, the RF generator has to withstand the reflected power for

~ lOOJ.LS because the time interval is scanty to tune the impedance matching unit.

So, a vacuum tube based RF generator is used in our experiment that can handle

2kW of reflected RF power. This RF generator can provide up to 500W of output

RF power in continuous rhode and 2kW of output RF power in pulsed mode. A

4k V, lAmp DC supply is used to provide the necessary input power to the genera

tor. Output power of this generator is produced in unbalanced mode. Input (A.C.)

voltage of the supply can be varied within (0- 415)V using power controller. A

step-up transformer with two secondary output sections, star and delta, is used to

reduce the ripple in the output D.C. voltage.

The pulse width of our RF operation is set to be ~ 50mB, synchronized with the

ambient magnetic field duration. Initially, low RF power in CW mode is applied

to produce plasma and optimize the matching unit. This is a procedure regularly

followed to avoid any large amount of reflected RF power during high power op

erations. This also improves the surface condition of vacuum vessel which reduces

the probability of arcs inside the system. After these steps, RF power is increased

gradually to the desired level for high RF power operations in a pulse width of


~ 50mS.

3.3.2 Transmission Line

The Radio Frequency Generators are designed to drive a specific characteristic

impedance, 500. This is the characteristic impedance of the output cable and allows

the generator to deliver full power at maximum efficiency. If the output impedance

varies from 500 the protection circuits in the generator reduce the output power

in order to protect the generator from overload. During our experiments, 1.8kW

of maximum RF power is fed to the helical antenna in pulsed mode for ~ 50mS

duration. For the transmission of RF power in unbalanced mode, RG- 213 coaxial

cables are used between the high power amplifier and impedance matching network,

up to power feeders. The forward and reflected power in this mode of transmission

are regularly measured with a Bircfl'M power meter, kept in a shielded enclosure

during experiments, to avoid any electrostatic interference generated by the power

meter. For data transfer from various diagnostics to the acquisition system, RG58

coaxial cables are used in our experiments [1, 2].

3.3.3 Impedance Matching Networks

The term impedance is used to convey the concept of the opposition of a system to

an energy source; particularly one that varies with time. Impedance matching for

minimizing reflections and maximizing power transfer over a large bandwidth (also

called reflectionless matching or broadband matching) is the most commonly used.

To prevent reflections of the signal back to the source, the load, which must be to

tally resistive, must be matched exactly to the source impedance, which again must

be totally resistive. The maximum power transfer theorem, RF systems and the

matching networks designed for our experiments, are elucidated in the forthcoming

sections.

3.3.3.1. Theory

In general, impedance is a complex number, which means that loads generally have

a resistance to the source that is in phase with the source signal and a reactance to


the source that is in quadrature to the phase of the source. The complex impedance

is the vector sum of the resistance and the reactance. Whenever a source of power

operates into a load, the greatest power is delivered to the load when the impedance

of the load (load impedance) is equal to the complex conjugate of the impedance of

the source (that is, its internal impedance). Impedance matching is necessary for

loss minimization in transmission lines, signal to noise ratio maximization in input

stages, minimization of signal distortion in transmission lines, avoiding wavefront

reflections and pulse superposition along with voltage and current amplification or

attenuation. For two impedances to be complex conjugates, their resistances must

be equal, and their reactances must be equal in magnitude but of opposite signs. If

the signals are kept within the narrow frequency range designed for by the matching

network, then reflections (in this narrow frequency band only) are also minimized.

At high radio frequencies, the spurious elements (like wire inductances, interlayer

capacitances, and conductor resistances) have a significant yet unpredictable impact

on the matching network. Above a few tens of megahertz (MHz), theoretical calcu

lations and simulations are often insufficient. The computational values are required

to set up the type of structure and target component values. Smith chart is one of

the basic tool used for determining transmission-line impedances for RF impedance

matching. It is used extensively during impedance matching to determine proper

arrangement of impedance matching circuit elements.

3.3.3.2. Networks used

Various matching network schemes are described in this section. Resistive networks

are used for the purpose of extremely wide bandwidth matching. Inductive and ca

pacitive matching are preferred for narrow bandwidth and high efficiency operations,

such as our experiment. A tunable matching network is necessary to compensate

for any change in the impedance to prevent the reflection of RF power back to the

amplifier. Generally, in a helicon discharge, the antenna and plasma are strongly

inductive. In our case, the helical antenna being radially mounted in the toroidal

vacuum chamber, it is enclosed by the S S chamber and in very high frequency

regime, its capacitive impedance is pronounced. The antenna impedance, measured


by impedance analyzer, turns out to be Zantenna = (3.5- j66), at an operating fre

quency of 32M Hz. The RF generator has a standard 500 output impedance. This

creates a severe impedance mismatch which prevents the RF power from being de

livered to the antenna because the power launched by the RF generator is reflected

at the load. Since the objective our experiments is to maximize the density in a

helicon plasma, minimizing RF power losses was critical. For our narrow bandwidth

operational regime (30- 35)M Hz, networks designed with inductive and capacitive

elements sufficed the purpose of impedance matching. Adjusting the capacitors in

the matching network each time a plasma parameter is change minimizes the re

flected power. Description of two types of impedance matching networks designed

for our experiments follows.

L-type Matching Network

This simple and most widely used matching network receives its name from the

orientation of components in this circuit. One reactance is in parallel with the

source (or load) and the other is in series with the load (or source). In our case,

reactance is in parallel with the source, so that the effective network matches from

high impedance to low impedance [3]. The series impedance-matching element then

resonates with or cancels any reactive component present, thus leaving the source

driving an apparently equal load for optimum power transfer. The components are

mounted in a high-pass configuration. The £-section is inherently a narrowband

matching network. The capacitors of matching network are adjusted during the

experiments to keep the reflected power minimum. When any system is driven by

RF drive, it's resonant behavior depends strongly on the quality factor (Q). The

L-type network has a high quality factor. A system with a high Q resonates with

a greater amplitude than one with a low Q factor, and it's response falls off more

rapidly as the frequency moves away from resonance. Though a high Q network is

difficult to tune with the necessary precision, maximum power transfer is possible

in such a circuit. The values of the components to be incorporated in the L-type

matching circuit are indicated in the smith chart, shown in figure 3.6.

The values of the components of matching circuit is obtained analytically by fol-


(SO- 250) pF ~ = s = = = ~ -.. = ~ "'':t = = I ~ ~ .... ~ N = ~ < '-'

Figure 3.6: Schematic of a L-type impedance matching network designed to obtain highQ matching during helicon discharge at an operating frequency 32M Hz.

lowing equations. For maximum power transfer using L-type matching circuit, for

a real source impedance R8 , real load impedance RL and a reactance X in parallel

with R8 , the real part is

When solved for X,

X= RLR; Rs -RL

In our case, Rs = 500 and RL = 40. So, for Rs >> RL, above equation can be

approximated as

Our L-type impedance matching circuit, depicted in figure, consists of a variable air

core inductor of (1.2- 4)~-tH and a pair of tunable air core capacitors, each having

a tuning range of (25- 125)pF. Up to 1.8kW of RF power is coupled at an oper

ating frequency 32M Hz). In our experiment, (90- 95)% of RF power transmission

has been achieved with the Pi-type matching network. However, demand of broad

operational bandwidth for the parametric study of plasma current drive lead to the

design of a Pi-type network for our latter experiments.

Pi-Type Matching Network

The impedance matching network can be realized as two "back-to-hack" L-type


matching circuits, both configured to match the load and the source to a virtual

resistance located at the junction between the two networks. Ease of tuning, con

tributed by more number of adjustable components and enhanced harmonic filtration

are helpful in this circuit. The loaded Q of this network can be defined as

where,

RH = largest terminating impedance of series or load resistance,

R = virtual resistance.

300nH .············.

I <

Figure 3. 7: Schematic of a Pi-type impedance matching network designed for broadband impedance matching during helicon discharge in the operating frequency range (27- 33) MHz.

The maximum bandwidth (minimum Q) available from this network is obtained the

virtual resistance can be expressed as

Our Pi-type impedance matching circuit, depicted in figure 3.7, consists of an air

core inductor of 300nh and two tunable air core capacitors. The capacitor parallel

to the source is Jennings CV DD- 100 tunable from 10- lOOpF and the capacitor

mounted parallel to the load is Jennings CVCJ- 1000 tunable over the range of

7- lOOOpF. In our experiment, up to 85% of RF power transmission has been

achieved with the Pi-type matching network.


3.4 Helical antenna and RF power feedthrough

Ionization sustained by characteristic waves excited by radio frequency fields in

plasma is an efficient process provided the length of the plasma column is several

absorption scale-lengths. Special antenna structures are used for exciting Helicon

waves in plasma. For antennas placed inside the plasma vessel, the coil is immersed

in the plasma medium where the wavelength is a few tens of centimeters and hence

couples RF power very well into the Helicon waves. In contrast, when located out

side the plasma vessel, the radiation efficiency of the structure is poor for the coil

size being negligible in comparison to the free-space RF wavelengths and the cou

pling takes place by transformer action.

Apart from most commonly used Nagoya III antenna, right helical antenna is also

popular for efficient Helicon wave excitation in plasma. This type of antenna con

verts an inductive (electromagnetic) field into electrostatic field. The legs of the

antenna parallel to the axial/toroidal magnetic field play the vital role. As the an

tenna current rises in these legs, an induced electric field (EM) gives rise to space

charge effect till the electrostatic field from these charges cancel the effect of in

duced electric field (EM). The space charge is of opposite sign at the opposite sides

of the antenna diameter due tom= +1 symmetry. This effect gives rise to a trans

verse electrostatic field at the center of the plasma column. This field pattern has a

considerable overlap with the electric field of the wave excited [4].

3.4.1 Design and construction of the Right Helical antenna

The dimension of the helical antenna are optimized by considering the dispersive

paramater a which is defined as

a= w J-Loeno R l Br

where l is the toroidal mode number, related to k11 as

(3.1)

(3.2)

----------------------------1


and other variables w, J-to, n0 , e, Br, Rare operating frequency, permeability, av

erage electron density, electronic charge, ambient magnetic field and major radius

respectively, all in SI units. The Br profile, as shown in figure 3.8, is computed [5J at

the plasma boundary as a function of a for a given set of poloidal(m) and toroidal(!)

mode numbers. Since the boundary condition for an infinitely conducting wall sug

gest Br = 0 at r = a, so the sequence of minima in figure 3.8 corresponds to the

radial eigen modes of resonant cavity.

1.2 •

0.8 -ca l!. -..

!!!. 0.4

4 8 a,

12

-•-m=O -•-m=+1 -•-m=-1

16

Figure 3.8: For a particular value of m, when l = 4, the minima obtained in the simulated Br(r = a) profile represent the allowed discrete values of a in our experimental device. The Br profile clearly shows that the antenna does not support any mode other then m = + 1 at present parameter regime.

Lowest radial eigenmode, being most appropriate for maximum power deposition,

was initially chosen for antenna construction. However, the antenna dimensions

constructed according to a = 4.1 could not be inserted in device due to mechanical

constraints. So, the final right helical antenna is designed for the second radial

eigenmode a= 7.8, for which the toroidal mode number turns out to bel= 4, for

m = +1 helicon mode. The value of a obtained from equations 3.1 and 3.2, suggests

a toroidal length >.11/4 = 11 em and phase velocity V<t> = 1.4 x 107 m/sec in this case.

Once the toroidal length of antenna is decided, the gain factor for this antenna is

evaluated as the square root of the antenna gain, given as

G = 7.8>.11 21ra

(3.3)


where a = helical antenna radius.

For above parameters, the antenna gain is G2 ~ 83. Convergence of Brad, for this

antenna dimension, at the radial boundary of the vessel is verified numerically us

ing a finite difference multi-dimensional shooting program [5]. An optimum radius

(a) of 6cm is chosen for the antenna to minimize the capacitive effect between the

antenna and the vessel wall. A right helical antenna made of copper strip of 3mm

width and 2mm thickness is used to excite Helicon wave of m = + 1 mode. The

antenna is provided proper electrical isolation with a small cut at the midpoint of

the longer strap of the antenna for RF power feed.

3.4.2 Design of RF power feedthrough

The RF power is delivered from the generator to the helical antenna through impedance

matching network, using RG- 213 coaxial cable. During low power experiments,

a distributed feeder system is used. The longer straps of the helical antenna are

oriented towards coaxial connectors. The central conductors of these connectors

are connected to these straps with the aid of 6" long M4 bolts to make proper

RF contact with straps. The bolts are isolated from the brass limiter by ceramic

fittings. These coaxial connectors are connected through the central conductors,

between the antenna straps and another set of N IF to N IF adapters connected

on a perspex flange. These adapters are then connected to the output of matching

network through coaxial cables. The antenna support structure, which is a hollow

brass cylinder, shown in figure 3.9, is supported onto the flange by a 4" long M6

bolt. Using this type of power feeder system along withaL-type matching network,

~ 800W of RF power could be coupled to the antenna.

However, during high power operations (2: 1kW), this type of distributed RF

power feeder design leads to sever arcing and signal distortion. So, a parallel RF

feedthrough is specifically designed for the transmission of Radio Frequency power

in to the vacuum environment during high power experiments, shown in figure 3.9.

High conductivity, nonmagnetic materials should be utilized in the RF feedthroughs,


Figure 3.9: Schematic of low power (left) and high power (right) feedthrough, designed and used in our experiment. The right helical antenna designed for our frequency regime is also shown here.

as they are less susceptible to the effects of current induction. Our feedthrough is

constructed entirely of copper which enhance their performance in RF induction

fields and designed for high voltages. Such feedthroughs are used primarily in high

power and high frequency applications. Since the operation is pulsed for 50mS, it is

observed that the characteristic skin effect of RF at these power and frequency levels

doesnot generate much heat to damage the flange to metal vacuum seals. Two cop

per plugs, slitted for proper RF contact, are welded on the antenna straps. These

plugs are pushed in to hollow copper cylinders. Two parallel 12cm long, hollow

copper cylinders, each of lcm diameter, are fitted on a 14mm thick perspex flange

using 14mm thick blind threaded copper caps. The power cable is connected on the

other side of these copper caps. This high power RF feedthrough, tested for~ 2kW

RF power transmission, is used successfully to couple up to 1.6kW of RF power to

the antenna.

3.5 Diagnostics used

In this section, various diagnostics used to measure plasma and system parameters

are discussed. The different probes used are given in the table 3.2.


S. No. Probe Plasma/System Parameter

1. Hall probe DC Magnetic field B0

2. Langmuir probe Electron density ne,

Electron temperature Te,

Floating potential V1

3. Emissive probe Plasma potential Yp

4. Electric dipole probe Wave electric field Ewave

5. Magnetic loop probe Wave magnetic field components Br,O,<P

6. Rogowski coils Plasma current Ip

Table 3.2: Various diagnostics designed and used to probe the helicon wave sustained plasma in our toroidal device at present parameter regime.

3.5.1 Hall probe

Axial and radial hall probes are used to measure the magnetic field distribution in

the system in absence of plasma. Measuring range of these probes are 300(±0.l)J.LT-

30(±1)mT. It works on the principle of Hall effect. These probes are attached with

a F.W BELL Series 9900 Gaussmeter.

3.5.2 Langmuir Probe

Use of a metallic electrostatic probe to diagnose plasma was proposed first by Lang

muir [6]. Since then it is the most widely used diagnostics for characterization of low

temperature, magnetized plasmas [7] as well as non-Maxwellian plasmas [8]. It re

mains the most easiest and accurate way to make local measurement of density (ne),

temperature (Te) and to obtain information about the type and nature of the dis

tribution functions of the charged species present in the plasma. In the such types

of plasmas as mentioned above, usual configurations of Langmuir probes such as

cylindrical, shown in figure 3.10, spherical and disc shaped probes are used [9]. The

general theory for Langmuir probe have been discussed and elaborated by many au

thors [8, lQ-13] in context of Maxwellian as well as non-Maxwellian plasmas. Since,

the Langmuir probe technique involves exposing a metallic collector to the plasma,

~----··--


the probes are generally made of refractory materials such as tungsten, tantalum,

graphite, molybdenum, etc. or of materials which are chemically inert such as plat

inum and gold.

The primary use of a Langmuir probe diagnostics in our experiment is to measure

the plasma parameters and then to measure the electron distribution (EEDF) in

different operating regimes. Objective of the EEDF measurement is to establish

our operating regime suitable for the investigation of nonresonant current drive by

helicon waves.

SStube Glass cover

:::::::::::::::::::;::::;:::::::::::::::::::::;::::::

4mm

:::::::::::::::::::::::::::::::::::::

7cm

Seml-rlgldco-axlal cable

Figure 3.10: Schematic of an ordinary or uncompensated Langmuir probe, used to measure the plasma parameters during helicon discharge experiments.

However, few modifications must be made in the usual configuration of this probe

to make measurements in strong magnetic and RF fields. Due to oscillating plasma

potential in RF environment, it is always difficult to get the correct values of Te

and floating potential with the probe characteristics of usual Langmuir probe [14].

In strong RF environment, oscillating current in the antenna structure generates

oscillating magnetic field in the vacuum and plasma. Owing to the plasma potential

oscillation with applied RF field, the probe characteristic shifts back and forth along

the voltage axis in accordance to the magnitude of amplitude of oscillation. This

lead to a shift in the floating potential and the usual graphical methods of analyzing

electron currents overestimate the electron temperature. The driven RF field inside

a plasma can also get spread in frequency when there are low frequency fluctuations

in plasma potential.


For conventional Langmuir probes, electron current collected by probe (Je) at a

given bias voltage (Vb) is given by

Ie = Ioexp(Vb - V, + V,. 1) (3.4)

Here, Vrt is the magnitude of RF oscillation over plasma potential (Yp) and / 0 con

tains rest of the physical parameters, held constant for a set of electron temperature

measurement. Due to this non-linear relation of Ie with v,.1, use of low pass fil

ter is not sufficient for the elimination of errors introduced by rf fluctuations. To

compensate for such errors occurring in probe measurement in rf environment, spe

cial arrangements are made in two ways [15]; one approach is entirely passive, the

other uses external active components. Comparative study of both the methods of

compensations [15] indicates that the distortion in the average characteristic due

to potential excursion, is maximum in case of active approach. The characteristic

of the passive probe remains unaffected by the plasma RF, owing to its broadband

response. In presence of non-sinusoidal potentials, the passive design is superior in

its ability to follow the plasma fluctuations.

Factors affecting Langmuir probe measurements

The use of standard Langmuir probe theory in Helicon discharges is constrained

by many factors: sheath effect, which depends on the probe size, a strong de mag

netic field which distorts the I - V characteristic of the probe, RF oscillations of

the plasma potential, plasma temperature which determine if collisions in the probe

sheath have to be taken into account and the probe orientation in presence of de

magnetic field. However, couple of these issues have been taken care during previous

discussions. Some of these factors, relevant for our experimental setup, which can

affect the measurement of plasma parameters with a Langmuir probe in RF plasmas,

formed in a magnetic field, are listed below.

1. Sheath effects : Since, the collection area of a Langmuir probe is approx

imately equal to the area of the sheath, the sheath thickness can be assumed to


be approximately equal to the De bye length for an unbiased probe. However, this

assumption can be an underestimate of the sheath thickness for a probe which is

applied a large negative bias, due to which electrons are expelled from a larger region

due to the sheath. An estimate of the sheath thickness can be obtained from the

Child-Langmuir relation together with the Bohm current as

~ = 8J27rEo fe y3/2

l 9 V-:;;;, Prf32 (3.5)

where (3 is a constant proportional to Pr/ Sn the ratio of probe and sheath radii,

l is the probe length, V is the voltage across the sheath and i is the ion current

collected by the probe. The ion flux, j at the sheath edge can be determined from

the ion acoustic speed given as

. i m J = 21rSrl = 0.6enoy--:;;;,- (3.6)

here it is assumed that the Bohm criterion holds good. Equations (3.5) and (3.6)

gives

S {32- 1.05\2 1.5 r - ADTJ

Pr (3.7)

where TJ is defined in terms of the probe bias Vs and the plasma potential Vp as

_ -e(Vs- Vp) TJ- kTe (3.8)

For cylindrical probes, sheath effect can be neglected if the ratio of probe ( Pr) and

sheath (Sr) radii is less then 3 [12,16]. The sheath radius can be calculated, in terms

of basic plasma parameters derived from the above formulae, as

500V.314

s- b r- 1/2 3/4

ne Te (3.9)

where 'Vb is the bias voltage on the probe, ne is the plasma density and Te is the

electron temperature. Sheath radius (Sr) turns out to be O.lmm and the Pr is

0.5mm, in our case. So,

Sr Pr = 0.2 « 3 (3.10)

which is small enough to neglect the sheath effect in our case. The fact that we

are in the collisionless thin sheath regime also means that we can use Druyvesteyn's


formula to analyze the Langmuir probe data. The effect on the Langmuir probes

due to the applied magnetic field is negligible when the probe dimensions are much

smaller than the Larmour radius (rL) of the collected plasma species. In present

operational conditions, rL turned out to be 0.1mm which is much smaller in com

parison to the probe dimensions.

2. Magnetic field effects: The effect of ambient magnetic field on Langmuir

probes is significant only when the probe dimensions are much bigger than the

Larmour radius of the collected plasma species. The ion collection is usually not

severely affected due to relatively large mass and large Larmour radii. Once the bias

of the probe become small or positive, the probe collects low energy electrons which

have a small Larmour radius. The Larmour radius is given by

ffiV_i

T£ = JqJB (3.11)

Electrons with Larmour radius smaller than the dimensions of the probe can either

be collected through cross-field diffusion or from flux tubes intersecting the probe.

It has been shown that the electron saturation current is reduced by a factor [17]

8 = 16,\ei [D.ill/2 1 + (1i/Te) D11 21ra

(3.12)

where Aei is the mean free path for electron-ion collisions and D .l is the diffusion

coefficient across the magnetic field lines given by

D.1 = D11 (1 + w~) (3.13) vet

where Wee is the electron angular cyclotron frequency, Vei is the electron-ion collision

rate and D11 is the diffusion coefficient along the magnetic field lines given by

Te Du=-

vei (3.14)

Before experiment, an electron has a temperature of 1 e V and an ion has a temper

ature of 0.025 eV in the neutral gas. Taking the minor radius of the experimental

system a as 10.5 em, for a typical argon plasma, the reduction factor 8 is in the order

of 0.1 and can be neglected. Then for the parameter of our discharge, Vei ~ 1068-1

and Wee = 3 x 1098-1. For electrons ~ ~ 103 and the electron radial diffusion llei


is strongly inhibited. Since for ions, ~ ex (!!!G.M·), the ion diffusion is not severely Vet '

limited and the usual standard expressions for the electron and ion probe currents

can be used with appropriate corrections for the effective probe area.

An important consideration for temperature measurement is the effect of the plasma

potential on the probe current particularly at small biases. The electron tempera

ture should be measured in the region where the probe bias is less than the floating

potential [18, 19]. The reason lies in the simple fact that near the plasma potential,

cold electrons with small gyroradii are predominant and therefore the saturation

electron current is difficult to reach since it is diffusion limited. Thus, the magnetic

field has no effect on the parameters measured in the Helicon plasma with a Lang

muir probe.

3. Probe orientation: Before insertion of Langmuir probe in a de magnetic

field aided plasma such a Helicon wave sustained plasma, it has to be ensured that

the probe is capable enough to sample the complete distribution function of the

plasma constituents. This implies that the measured value of plasma parameters

are not affected by the cyclotron motion of plasma particles in presence of ambient

magnetic field which can induce kinetic effects.

Moreover, in presence of an ambient magnetic field, such as in our experimental

device, the tip of Langmuir probe should be aligned perpendicular to the magnetic

field so as to eliminate any discrimination of low energy particles by the probe. For

a cylindrical Langmuir probe, aligned along the magnetic field, the electric field(E)

between plasma and the probe acts perpendicular to the magnetic field(B) lines

which results in changing the direction of E x B drift of the guiding center of the

plasma particles in azimuthal direction with respect to the axis of the probe. Thus,

the low energy plasma constituents, owing to a smaller Larmour radius than the

probe sheath thickness, are restricted to reach the probe due to the superposition of

cyclotron motion on the guiding center motion. The resulting discrimination of low

energy particles by the probe leads to a higher measured value than the actual value

of temperature. This effect is eliminated by orienting the probe tip perpendicular

to the magnetic field so that the orbit effects are taken off the scenario by keeping


Ex B zero. In all the experiments discussed in this thesis, the Langmuir probe tip

is aligned perpendicular to the magnetic field.

RF compensated Langmuir probe design

Due to simplicity of construction and better performance at all harmonics of rf

(15], the passive compensation technique is used in the present experimental set

up [20, 21]. The design is similar in principle to one suggested by Gagne and

Cantin [22], improvised by Dilecce et al (23] and Chatterton et al (24]. In order

to overcome the problem of existence of RF structure on the plasma potential(Vp),

the probe potential is allowed to follow the variation in Vp so that the DC probe the

ory can be followed even during our present analysis. The variation in Vp is sensed

by a large remote floating electrode which is capacitively coupled to the probe tip.

The probe tip is isolated from the ground by means of tuned chokes which provide

high impedance(100k0) at the tuned harmonic frequencies. The capacitor joined

between the floating electrode and probe tip is chosen so that it's capacitance is

greater then that of the floating electrode at the operational frequency( w). The LC

chokes, used for the construction of tuned chokes in the RF compensated Langmuir

probe, provide an attenuation of -35dB at 30M Hz. To obtain very high impedance

at tuned frequencies, the condition w2C L ~ 100 is imposed on the sheath potential.

During this calculation, Vp ~ 100V is assumed, so as to achieve a compensation

within 1% of plasma potential. So after the floating electrode/ guard ring, which

surrounds the probe, two stages of LC chokes/filters, which consist of two pairs of

ceramic capacitor and miniature inductor, are used to suppress the rf harmonics,

as shown in figure 3.11. Such an arrangement provide a stable range of frequency

compensation components to use for the modified Langmuir probe. To enhance the

immunity of the probe against oscillating RF structures on plasma potential, the

potential at the tip of LC-chokes should be greater then Vp so as to block any RF

structure riding on Vp in the probe signal. The simple theory and calculations neces

sary for the construction of the RF compensated Langmuir probe for our operational

regime, are provided below.

------------------------~---------- --~-


Schematic of the probe is shown in figure 3.11. For simplicity, only the first harmonic

is considered, as for higher harmonics, value of (3 becomes negligibly small. So, the

harmonics considered in the plasma potential are

Vp = Vpsinwt + (3Vpsin2wt

where (3 is the fraction of first harmonic present in Vp .

The total impedance of the probe is

Z 1 jw£1 jwL2

r=--+ +-----jwC 1- w2Cr1L1 1- w2Cr2L2

(3.15)

(3.16)

where Cis the effective sheath capacitance, CTl, Cr2 are the parallel tuning ca

pacitors and £ 1 , L2 are the inductors for the first and second harmonics respectively.

The capacitance of the floating electrode to the plasma is

C= EoA s

8 = >. Jv- Vp D. kTe

(3.17)

(3.18)

where Vis the probe bias voltage, Vp is the plasma potential, S is the sheath thickness

and A is its area. Now,

Vo -w2CL - = ---=-=--=-Vp 1-w2CL

(3.19)

where Vo is the potential at the junction of the probe tip and the LC-circuit for first

harmonic rejection and

L = £1 + L2 1 - w2CT1L1 1 - w2Cr2L2

Hence, the voltage across the sheath is

Vp Vp- Vo = 1- w2CL

(3.20)

(3.21)

In order to achieve compensation within 1 %of Vp, it is required that w2CL ~ 100.

Capacitance for a cylindrical guard ring of 2cm diameter and 10mm length, with a

dielectric (Teflon) cylinder to isolate the probe tip from guard ring, is 300pF for our

operational regime. The 2nd term = 3rd term = Vp, in equation 3.16, is considered

to make sheath drop equal to 1% of probe potential and the harmonic rejection


SStube RF hannonlc rejection sections

-~· 0 C-~=·~'=·r -~ 0·"-'=·~, =· Semi-rigid SS shield e~tended

co-axial cable from coax1al cable

4mm

Floating electrode/ Guard ring

Biasing capacitor

20mm

Figure 3.11: Schematic of the RF-Compensated Langmuir Probe (right), designed for a range of operating frequency (27- 34)MHz, used for plasma parameter measurements during helicon discharge experiments.

components are calculated.

Measurement techniques

The arrangement for the measurement consists of a small conducting wire shielded

throughout its length with its end tip exposed to plasma and the voltage at the

tip is measured with respect to vacuum vessel with a high input resistance oscil

loscope [25]. The plasma is diagnosed by a set of ordinary and RF compensated

Langmuir probes (RFCLP), placed opposite to each other, toroidally besides the an

tenna to observe the radial variation of plasma parameters. Two different floating

power supplies has been used to bias the probes for the comparative measurement

of plasma temperature. The ion saturation current is measured from the tip of the

probe which is connected to the oscilloscope through a lOkn resistor and a power

supply. The probe tips of both the probes consists of a tungsten wire 0.5mm in

diameter and 4mm in length. The probes are connected to the external connectors

using a coaxial cable passing through a movable stainless steel probe shaft. The bi

asing circuit and the oscilloscope are provided a clean and common reference ground

connection with the vacuum vessel. A typical IV-characteristics obtained with the

RFCLP during Helicon breakdown, is shown in figure 3.12.


200

160

- 120 -1 - 80 -;; ... -

40

0

-40 -75 -50 -25 0 25 50 75 100 125

Bias Voltage (Volts)

Figure 3.12: A typical probe characteristics obtained during plasma temperature measurement, during helicon discharge experiments, using RF-compensated Langmuir probe.

Probe cleaning by resistive heating of the probe tip biased to collect electron sat

uration are attempted during continuous operation to improve the reproducibility

of the probe characteristic curves. Comparative study of both the probes in our

system clearly demonstrates the capability of rejection of potential excursion by

the RF compensated probe designed. This part of the experiment and the results

obtained are discussed in detail in Chapter4. Measurements are done with the com

pensated probe mounted in radial ports at 50° inclination from the horizontal plane.

Perhaps the most obvious problem in hot and dense plasmas, is that the probe must

survive the heat load. This places a limitation on the plasmas that can be studied.

The solution is to implement a host of measures which in their totality will bring

the heat load to a minimum. One is to build big probes. Probes in large tokamaks

are often bulky and are made from graphite or other heat resistant materials. In our

case, tungsten probe suffice this purpose. A second step is to avoid drawing large

electron currents, which can cause much greater heating than ion currents. This is

done by keeping the probe biased to roughly floating potential or below. It causes no

great loss of information because in magnetized plasmas the electron collection re

gion is unreliable for diagnosis because of perturbations to the ideal Boltzmann law.

Therefore fitting to the characteristic much above the floating potential is avoided.


A third step is to limit the time duration of the probe's exposure. However, dur

ing our pulsed operations of 50mB duration, no such heating of probe tip is observed.

A valid application of the Druyvesteyn formula [26] requires collisionless electron

motion near the probe. This needs the Debye length (Ad) to be much less than the

electron mean free path (Ae)· Along with this condition, it is also necessary that

probe radius must be smaller than Ae to be non-intrusive effectively. In the present

case, maximum Ad is 0.1mm, minimum Ae is 10mm and the probe radius is 0.5mm.

So, the necessary condition to allow this probe for electron energy distribution de

termination is sufficed. The plasma potential value obtained by emissive probe is

(80 ± 5)V and the floating potential value obtained by RFCLP is (18 ± 2)V. Elec

tron Energy Distribution Function (EEDF) is determined using the RFCLP from the

second derivative of the IV characteristic that requires collisionless electron motion

near the probe [26].

e3/2 A 1oo Ie = to= (U + Vb - Vp)F(U)dU

v8m Vp-Vb (3.22)

die e_3/2 A 1oo dV, = . to= F("\lb - Vp)d"\lb

b v8m Vp-Vb (3.23)

d2 Ie e312 A d~2 = vrsmF(Vp- "Vb) (3.24)

where A is probe area, U is kinetic energy and F(U) is the normalized isotropic

component of EEDF. By setting U = (Vp- "\!b), F can be written in terms of the

kinetic energy, as vrsm d2Ie

F(U) = e3/2 A dV.2 (3.25) b

This approach needs the Debye length (Ad) to be much less than the electron mean

free path (Ae) and the probe radius smaller than Ae to make the probe non-intrusive

effectively. Although determination of EEDF by Langmuir probes in RF plasmas, is

subject to misinterpretation, cautious design and proper operating procedure results

in a very reliable and versatile experimental tool.


3.5.3 Emissive Probe

Though local measurements of basic plasma parameters can be achieved easily us

ing a Langmuir probe, with an accuracy up to some extent, errors can incur in the

measurement of plasma potential. It is well known that the 'knee' of electron sat

uration region in the probe (I-V) characteristics of a Langmuir probe can give the

estimate of plasma potential [27,28]. However, this "knee" is generally not obtained

in RF and microwave experiments. So, for such measurements, hot probe, instead

of cold probe, is used. The working principle of plasma potential measurements

using hot probe with in the plasma, relies on the dependence of the emitted electron

current on the plasma to filament potential difference [29]. Emissive probe [30, 31]

is used for the plasma potential measurements in synthesized plasma streams [29].

An emissive probe, made of tungsten tip, is used in our experiment to measure the

plasma potential which is essential for electron distribution estimations. The typical

construction of this probe is depicted in figure 3.13.

2.5cm

~ Filament wire

·copper pipe

Ceranmic tube

......---- Connecting wire

6mm

1.8mm

Figure 3.13: Schematic of a emissive probe used for plasma potential measurement during helicon discharge experiments.

The "droop" method is applied for the measurement of plasma potential using emis

sive probe in our experiment. This method is applicable in steady-state experiments

in which Te >> T1, where Te and T1 are the electron and filament temperatures, re

spectively [32]. This method is based on the observation that time-varying filament

temperature causes the floating potential to vary and the proportionality constant

between the two is extremely sensitive to the difference between the biasing voltage

and plasma potential.


An emissive probe is a hot wire probe immersed in plasma to measure plasma po

tential directly. It is generally made of tungsten which is heated enough to allow

thermionic emission of electrons. The emissive probe used to measure the plasma

potential in this system is made of tungsten wire of 0.01 mm diameter and 5 mm

length. A maximum current of 2A can be fed to this filament to heat it up. This

filament is enclosed in a machineable ceramic enclosure with two wire slots as shown

in figure 3.13. This whole setup is fitted with a probe shaft and a flange from the

front radial port.

..... :z: ~ ---------a COLO PROBE

!INCREASING Tw

BIAS VOLTAGE

Figure 3.14: The effect of increasing the filament temperature on emiSSive probe characteristics and a cold probe characteristics, are shown here. (M.A.Makowski and G.A.Emmert, 1983)

When the probe bias voltage is more negative than the plasma potential, the probe

inject electrons in the plasma generating an effective ion current. The emitted

electrons can change the cold probe characteristics only below floating potential

because in this regime, emitted electrons have no significant potential barrier to

overcome in leaving the probe. When the probe bias voltage is positive with respect

to the local plasma potential, electrons ejected from the probe return back to the

probe and the current collected by the probe increases sharply. This effect leads

to a break point which demarcates the emissive probe trace from that of a cold

probe, illustrated in figure 3.14. The voltage at which this break point occurs, is


considered as the space potential. This process does not perturb the plasma because

it depends directly on plasma potential rather than electron kinetic energy (as in

Langmuir probe). It is less sensitive to probe surface contamination when heated

surface provide electron emission. Typical plasma potential measured in our toroidal

Helicon plasma device during Helicon discharge is 80V.

Method of measurements and analysis of data are chosen in such a way so that the

plasma potential is minimum affected by RF fluctuations. The RF noise is filtered by

averaging many probe data samples in order to reduce the overestimation of plasma

potential. Although an overestimation of ± 10% of plasma potential is expected

in our experimental conditions, it does not lead to any serious problem during the

derivation of electron energy distribution from the RF compensated Langmuir probe

characteristics using Druyvesteyn formula, detailed in Chapter 4.

3.5.4 Dipole Probe

A dipolar double probe [33] is used for in situ measurement of the time-varying

wave electric field the Helicon wave sustained plasma in our operational regime.

An electrically insulated linear dipole has two shielded ends, made of tungsten,

floating in plasma to pickup time varying electric field signals. It works on the

principle of developing a small open loop voltage across the dipole terminals and then

transfer the information from an in situ dipole to an external acquisition system.

The equivalent electrical circuit of a dipole can be represented by a single small

capacitor in series with an AC voltage source. The capacitance per unit length, C

of the dipole is given by 7r Eo

C = 2ln(l/d) (3.26)

where lis the electric dipole length and dis diameter of the wire. The dipole probe

used in our experiments is as shown in figure 3.15.

This probe has an l of 20 mm and d of 2 mm. This gives the dipole probe capac

itance, C = 6 pF fm. The relation between the output voltage, Vout of the electric

dipole and the Helicon wave electric field is given by

(3.27)


2.5 em

20mm .. ... ~ Dipoletip

Copper pipe

Ceranmlc tube

Connecting wire _.,.---

65

Figure 3.15: Schematic of a dipole probe used for wave electric field profile measurement during helicon discharge experiments.

here, w = 21r f and Rr is the terminal resistance of the coaxial cables connected

to the probe. In these experiments, the source frequency, f is 32 ± 0.2 M Hz. For

Rr =50 n, C = 6 pF/m, Vout comes to be

~t = 1.2 x 10-5 m (3.28)

The very small V / E ratio is because of the low capacitance of the probe or ( equiv

alent to) its high internal resistance.

z = -1- = 42 kn

wCl (3.29)

The electric field can be calculated from the difference in the voltage signals received

at the two arms of the electric dipole as

V2-Vi Ewave = V/m

8 (3.30)

here, Vi, V2 are the two voltage signals and s is the tip separation of electric dipole.

A dipole probe made of tungsten and insulated by Teflon jacket, is used in our

experiment. Both the tips are connected to a pair of coaxial cables and the signals .

are observed on the oscilloscope. The electric field is calculated from the difference

between the voltage signals received at the two arms of the electric dipole. The

electric field measured by dipole probe is cross-checked with the magnetic loop

probe signals during our experiments.


3.5.5 Magnetic probes

Magnetic loop/B-dot Probes

Small wire loop probes have routinely been used in plasmas to measure the time

varying magnetic flux density B(t). Such small wire loop probes, generally known

as B-dot or ~~ probes, produce a voltage that is proportional to the component of

the time varying magnetic flux ( rv loop area x ~~) that is directed normal to the

plane of the loop. For a sinusoidal fluctuation, B(t) = B0sinwt, the induced voltage

is

Uind = -nBoAwcoswt, (3.31)

i.e., Uind ex w. From the B-dot voltage and relative phase, spatially dependent elec

tric field and the current density can also be determined through Maxwells equa

tions (34]. Though analogous study is also possible by many advanced techniques

like soft X - ray emission [35] or motional stark effect [36] etc., but the simplic

ity and wide variety of magnetic loop probe diagnostics [37] available still remain

unmatched. For this reason the magnetic loop probe diagnostics remain an indis

pensable for the study of wave magnetic field components in Helicon discharges.

However, for frequencies above 100kHz relatively small magnetic loops can be con

structed to find a compromise between high spatial resolution and reasonably large

induction signal. Magnetic loop probes have been used in many different plasma

experiments, e.g., fusion related devices [38, 39], mirror machines [40, 41], induc

tively coupled plasma experiments [42], plasma flow generators [43], and helicon

plasma sources (44, 45]. In high density and high temperature plasmas, magnetic

loop probes are generally located at the outside of the plasma, which requires addi

tional considerations for data evaluation. Under less demanding plasma conditions,

magnetic loopprobes can be introduced directly into the plasma for a truly local

B-measurement.

Two types of B-dot probes are used in our experiment. During low power dis

charge, three movable mutually orthogonal magnetic probes, are used to measure

the three wave magnetic field components (Br,O,I/J) in plasma. Five turns of 1mm

thick enamelled copper are made on a former of 5mm diameter to make the mag-


netic probes [46,47] which are mounted on the torus through the radial port at 50°

inclination from the horizontal plane, shown in figure 3.16.

Figure 3.16: Schematic of a multiturn B-dot probe designed to measure the wave magnetic field profiles during helicon discharge experiments.

In this design, a multi-turn floating B-dot probe is used in a differential arrange

ment, to measure the wave magnetic field components. However, due to presence of

significant RF fluctuations during high power helicon discharge, same probes cannot

be used and electrostatic pickup rejection technique need to be invoked. So, center

tapped B-dot probes with bifilar winding [37, 48] are used for precise measurement

of Br,O,tf> components during helicity current drive experiments at high RF power,

shown in figure 3.17.

- - - - - - - .... Oscilloscope

Coaxial Cable

Glass Cover ' -------------------------------------------~

Figure 3.17: Schematic of a bifilar center-tapped B-dot probe, designed to ensure proper measurement of helicon wave magnetic field profiles, inherently eliminating capacitively coupled components.

In this type of arrangement, one winding of 50 turns is wound in opposite direction,

bifilar with another winding of 50 turns, using enamelled copper wire of 0.1mm

diameter, on a Teflon former of 4mm diameter. The design remains effectively a

center-tapped with a subtraction transformer. The unidirectional current induced

by any gradient in the local plasma potential, in both the windings get cancelled

upon being added, reducing capacitive pickup more effectively in comparison to the


single winding. Measurements are done by mounting three mutually orthogonal

center-tapped bifilar magnetic probes in radial ports, shown in Fig. 3.2, at both

sides of the helical antenna. The plasma is radially scanned by these probes. The

probes are inserted inside a tight-fitting glass tube, inserted into the vacuum vessel

through a vacuum seal.

Sensitivity measurement using Helmholtz coils

A Helmholtz pair consists of two identical circular magnetic coils (radius, R) that

are placed symmetrically one on each side of the experimental area along a common

axis, and separated by a distance equal to the radius of the coil. Each coil, with N

number of turns, carries an equal electrical current (I) flowing in the same direction.

It is observed that a cylindrical region extending between the centers of the two coils

and approximately i of their diameter will have a nearly spatially uniform magnetic

field, given by

In the center of the coils, x = ~,

3

B ( z) = ( ~) a JL~l

Sometimes, a slightly larger separation (> ~) improves the field uniformity. A

Helmholtz coil pair of R = lOcm and N = 25 are used. A magnetic field of

2Gauss/Ampere(max.) produced is measured in this coil configuration using Hall

probe. The sensitivity for multiturn probe and center-tapped bifilar probe, obtained

from the slopes of the linear fit of the calibration graph, shown in figure 3.18, are

~ 0.2V/Gauss and~ 2V /Gauss respectively. A poloidal array of four center-tapped

probes are used to measure the poloidal variation of Br,O,rp and an axial array of six

center-tapped probes are used to measure the axial variation of Br,O,rp·

Frequency response measurement

For frequency calibration of B-dot probes, a signal generator (Agilent 8648A) is used

in the Helmholtz coil configuration. The frequency range is lOOkHz-lOOOMHz and


10

-rn 8 -0 ~ 6 -::;, a.

4 -::;, 0 G) 2 .c e

D.. 0

1.0 1.5 Magnetic field in Helmholtz

Coils {Gauss)

69

2.0

Figure 3.18: Linear fit to the calibration plots of multiturn B-dot probe (circles) and center-tapped B-dot probe (box) reveal better performance of the latter.

the output of the generator is +lOdB(max.). However, impedance of Helmholtz coils

vary with frequency and hence the magnetic field produced by it also varies with

frequency. Therefore, for frequency calibration of B-dot probes, transmission line is

preferred Helmholtz coil because the impedance of transmission line is independent

of frequency. The coaxial transmission line used for determination of frequency

response of B-dot probes has the outer radius (b = 15.2cm) to inner radius (6.6cm)

ratio ~ = 2.3 and a characteristic impedance of 50ft

The 30cm long transmission line has 170nH inductance per unit length and

67pF capacitance per unit length. The line is terminated at the ends so as to main

tain constant ~ ratio (49, 50]. The B-dot probes are mounted in the intermediate

region of inner and outer cylinders of the transmission line so as to intercept the

azimuthal magnetic field. The line is energized with the signal generator (Agilent

8648A) through an appropriate power amplifier at one end, while the other end is

terminated with a 50n impedance load. The frequency response of single turn B-dot

probe, multiturn B-dot probe and center-tapped bifilar B-dot probe, obtained from

transmission line measurement, are shown in figure 3.19. As evident from the figure

3.19, center-tapped bifilar B-dot probe excelled in performance and hence, preferred

for high power helicity current drive experiments.


Frequency (MHz)

Figure 3.19: Above figure represents the frequency response of the center-tapped B-dot probe, designed and calibrated for measurement of wave magnetic field profiles in present operational regime.

Dual Rogowski coil system

Use of a toroidal air core solenoid around a conductor to measure the voltage induced

by the transient current was first proposed by Chattock in 1887 [51], later elaborated

by Rogowski and Steinhaus in 1912 [52]. Since then it has remained the most easiest

and widely used diagnostics for transient current measurement as it is independent

of the manner in which main current threads the opening of Rogowski coil. If N is

the number of turns of the Rogowski coil, R is the major radius and r is the minor

radius, then the voltage difference at the terminals is given as

V = J.-loN1rr 2 dip= Kdlp 21rR dt dt

(3.32)

Ip(t) = ~ 1t V(t)dt (3.33)

A numerical or analog integration of the measured voltage V gives the total plasma

current. It's capability of measuring wide range of current, tunable risetime [53],

non-saturation by over current, very high band width, absence of heating due to

galvanic isolation and very good linearity attracted it's use in measuring plasma

current in fusion plasma experiments [54-57). In tokamaks, Rogowski coils are in

stalled either inside or outside the vacuum vessel to measure total plasma current.

In large tokamaks like JET, TCV, instead of continuous Rogowski coils, appropriate

linear combination of magnetic probes, mounted outside the plasma current channel,


are used to estimate total plasma current. But such estimations usually incur large

measurement error. However, Rogowski coils mounted outside the vacuum vessel

are generally affected by eddy current and skin effect. For inner Rogowski coils, the

vacuum compatibility of coil materials and its installation are the important issues.

In some early tokamaks (e.g. TEXT, TEXTOR) had partial Rogowski coils (usually

called sinO, cos(} coils) for measuring the plasma position. These coils are named so

because the coils are made in such a way that its winding varies as sinO or cosO.

Since the prime objective of our experiment is to investgate current drive in plasma,

the Rogowski coil is the prime diagnostics of our investigation. The Rogowski coil

is intended to measure the average plasma current in our device during different

parameter regimes. We also intend to accomplish the parametric dependence of the

plasma current generated in our operational regime.

However, to achieve ideal properties in a practical coil, considerable care is to be

taken in its design, construction and pickup rejection due to stray fields. Rogowski

coil is a balanced probe since the two ends of the coil are held at equal potential, but

opposite in polarity, with respect to the ground during transient current measure

ments, where as in a coaxial cable both the conductors have unequal impedance to

ground and hence is an unbalanced system. However, during plasma experiments,

very often a Rogowski coil is terminated with a coaxial cable. This balance to un

balance transition makes this diagnostics prone to undesirable spurious electrostatic

pickup along with the desired inductive voltage pickup during transient current mea

surements. Though in case of very high magnitude of current (~ lOOOAmps.) this

problem does not lead to any discernible error, but low frequency, low magnitude

current measurements (:S lOOAmps.), suffer largely due to interference from the un

balance currents flowing through the shield [58]. The transition from unbalanced to

balanced configuration can be done using a ferrite core transformer in between the

twisted wires connected to the coil and the coaxial cable that enhance the mutual

inductance of coil and hence increase the voltage induced in the coil. But limited

bandwidth of a transformer restricts this method to a particular frequency range.

Moreover, in strong magnetic and RF fields, magnetic materials like ferrite need


a special care for shielding and the whole arrangement need to kept very near to

the Rogowski coil arrangement which makes it more cumbersome. Same can also

be achieved by using double Rogowski coils in tandem but this require an exactly

symmetric pair of coaxial cables between the coils and a differential amplifier with

high common mode rejection ratio. Other disadvantages of this configuration that

shows up only during low magnitude current measurements, will be elucidated later.

Use of bifilar coils can also serve the purpose to some extent but in big geometry this

arrangement is prone to error because it is hard to restrain affection of the toroidal

magnetic field due to the difficulty of firm winding of wire on the core.

We have introduced a method to measure very low plasma current (~ lOOA) with

two identical Rogowski coils, used in a manner that makes this system equivalent

to a center-tapped magnetic fluctuation probe. To distinguish it from the usual

double coil system, we will refer our coil configuration as dual Rogowski coil sys

tem. Since any electromagnetically coupled signal is a superposition of an electro

static( capacitive) and inductive signal, in present case, the dual Rogowski coils itself

work like a differential amplifier with a high common mode voltage rejection. Ca

pacitive pickups, being common mode type signal, has same magnitude and polarity

(as well as phase) on both the wires for a particular configuration whereas inductive

signals are differential mode signals in which the signal on one wire is same in mag

nitude but opposite in polarity on the other. So, two identical but oppositely wound

coils can be used to cancel out the common mode part in the signal and retain the

genuine inductive part only. Few advantages of this system over the existing double

coil arrangement are listed below:

1. This system does not require an instrumentation differential amplifier but an

amplifier with suitable gain for capacitive pickup minimization, and

2. Coil system can be coupled balanced to unbalanced coaxial transmission line

of the integrator circuit without any discernible phase error. This has been

verified during the tests carried out, elaborated later.

Rogowski coil construction


Figure 3.20: Schematic of the Rogowski coil system installed in THPD for plasma current measurement. The coils are surrounded by an electrostatic shield to prevent capacitive noise coupling, due to large voltage fluctuations.

The requirement for insertion of the Rogowski coil system into THPD constrains

the design of the diagnostics. Coils are designed so as to encircle the plasma column

inside vacuum vessel. Detailed design features of the coil system are depicted in fig

ure 3.20. Each coil is of Scm major radius with 700 turns wound on a Teflon former

of 3mm radius. Wire of 0.5mm diameter has been chosen to make the Rogowski

coil to minimize the coil capacitance. The inductance of a coil is 13J..LH/m and its

resistance is 800mnjm. Both the butted ends of each coil are tied together closely

to maintain the right alignment and to avoid cross-talk or pickup from neighbouring

conductor or stray sources of magnetic fields in close proximity. Two identical oppo

sitely wound Rogowski coils are mounted on a metallic ring, with 15mm 'separation

in between them. The ring is placed in a coaxial toroidal brass housing of 9cm major

diameter and 2cm minor diameter to provide electrical and thermal insulation to

the coils. To avoid any common mode current due to rapid voltage changes during

plasma shots, a toroidal S 8304 foil of 0.5mm wall thickness has been placed around

the coil inside the housing [61]. The minimum thickness of the housing wall is 3mm


and a 0.5mm wide toroidal opening has been made in the inner wall for flux linkage.

Vacuum sealing is taken care of by a Wilson feedthrough co-axially fitted on the

SS tube. Since the coaxial cable has a characteristic impedance of 50!1 which is

significantly different from coil impedance, it is not correct to terminate the coil at

the integrator end. In present case a lkO resistor is connected at the twisted ends

of each coil. Output of the detection coil is transmitted by the twisted ends of coils

through a metallic shield, inside a stainless steel(SS) tube of 5mm diameter, to the

BNC connectors fitted on the probe cup.

Integrator circuit design and construction

The operational amplifier used for integration is chosen on the basis of low offset cur

rent. Though the electrostatic pickup signal is weakend by the dual Rogowski coil,

to enhance the immunity and accuracy of the integrator circuit, a 1.5kw resistance

is incorporated across the input signal. This resistance aids in further minimization

of any weak electrostatic part (if any) still present in the input signal along with a

minimum diminution of the main signal. The frequency of plasma current, estimated

from the expected current rise time, is ~ lkH z. In order to overcome the problem

of high frequency rings ("' 1M Hz), low pass filter stage is also used in the circuit, as

shown in figure 1 in appendix. This filter minimizes the high frequency components

by -6dB and the input signal is diminished by ~ -40dB. Since, an integrator have

a low pass filtering effect, one need to be extremely careful regarding the choice of

low pass filters to be used prior to integrator circuit to avoid any interference.

The IC chosen for differential amplification (INA110) has suitable bandwidth, high

common mode rejection ratio, low gain drift and low offset drift. The differential

amplifier is used only for a double coil configuration. The amplified output signal is

fed to the integrator circuit. Amplification of a fast signal prior to integration is not

preferred in instrumentation provided fastness of signal is not so fast. In present

case, fastest rise time of plasma current signal is expected "' lmS. Though due

to 20dB amplification, an analogous decrease of gain bandwidth of the amplifier is

expected, use of an amplifier with a bandwidth 2:: lOOj.tS will suffice our purpose.


The amplifiers used in present system has a gain bandwidth :2: 1M Hz and hence

the passing signal will not suffer any distortion. Other methods, such as use of

multiple cascaded buffer stages so as to strengthen such fast signals, are useful but

it will make the present circuit cumbersome. An integrator will have a low pass

filtering effect but when given an offset, it will accumulate a value building it until

it reaches a limit of the system or overflows. So, a suitable op-amplifier with very

low noise, drift and offset current (~ lOpA) is used in the integrator circuit. While

using this diagnostic during pulsed plasma experiments, an isolation amplifier with

galvanic isolation is used to isolate the integrator circuit, which is on instrumenta

tion ground, from the data acquisition (oscilloscope) system ground. Bandwidth of

the isolation amplifier chosen, is 20kHz, which is higher then the integration time

constant and it is used with unity gain. The integrated signal (~ lkH z), before and

after isolation, has been verified for absolutely distorsion free output. A floating de

supply is used for the entire circuitry. Since any transient inherently contains fre

quency components lower than the fundamental frequency of the current of interest

and the offsets present take a long time to decay, they give rise to a DC offset in

the integrated output along with high frequency jitters. This effect can be reduced

by adding appropriate terminations as well as low pass filters in the Rogowski coil

system and integrator circuitry respectively. The lkO termination is chosen exper

imentally for impedance matching as well as it serves to define the output voltage

reference. Since Rogowski coils do not suffer from saturation and the mutual in

ductance is independent of the current being measured, they can be calibrated at

any convenient current level and the calibration remains valid for almost all currents.

Calibration and electrostatic pickup rejection

A quantitative comparison of electrostatic pickup for the different configurations

of Rogowski coil has been done in a test field. A schematic of the test setup is

shown in figure 3.21. To test each of the designs for capacitive pickup rejection,

apart from charging the capacitor to different values, the inductive load impedance

is also changed for each capacitor discharge. Conclusions drawn from the relative

electrostatic pickup measurements are provided below:


+ 3F

1:1 pulse Transfonner

-----------------------------.

Load 30mOhrnll00uH

Isolation Barrier I

76

Data

Figure 3.21: Schematic of the setup to test various coil configurations. For each case, the capacitor in the test circuit is charged to desired value and dissipated through the inductive load of known impedance.

The most commonly used single coil configuration, which is a single-ended-ground

referenced probe, turns out to be prone to common mode electrostatic pickup

(CMEP) due to a long unbalanced transmission line from the probe cup to the

integrator circuitry.

Coil Housing

Terminating Resistors lkOhm

t Probe cup with

insulated top cover with BNC connectors

Differential Amplifier

Low Pass Filter

Data

DAC Ground

_ ! Jnslrllmentation - 1 Ground

Isolation Amplifier

Figure 3.22: Schematic of the floating double Rogowski coil system designed and tested for electrostatic pickup rejection during transient current measurement in the test setup. In this figure, two isolated coils, oppositely wound, are placed in the metallic housing.

To compensate the error induced by unbalanced current in single coil configura

tion, a double Rogowski coil system is designed, as shown in figures 3.22 and 3.23,

in which two identical, terminated, floating Rogowski coils with opposite turns is

used in tandem. The difference between two equal but opposite ~; signals obtained

in this configuration, is accomplished using a differential amplifier. Any asymme-


Amplifier

Low Pass

(ffi!Hffiffi£HffiffiH!IDJ~f==~~]~"ti4==~~ Filter

Coil Housing Probe cup with insulated top cover

with BNC connectors

Data

-= DAC Ground

77

_ ! Instrmentation - 1 Ground

Isolation AmpHfler

Figure 3.23: Schematic of the ground dual Rogowski coil system, designed and tested for electrostatic pickup rejection during transient current measurement in the test setup. In this figure the floating shorted terminals of the coils are grounded through the housing to the device ground, according to the scheme shown.

try between the transmission (Tx) lines may lead to a finite phase error. Though

such differential measurements are immune to common-mode electrostatic pickup,

noise still couple with the genuine inductive signal by magnetic induction into the

loop formed between the two parallel lines. This type of response due to signifi

cant unbalance current, arising due to magnetically induced voltage into a loop, are

remarkable when the total loop impedance is low. So, in the next step, both the

cables are twisted to reduce the enclosed magnetic flux and expose both the cables

of twisted pair equally to capacitive noise. Though this design minimizes the elec

trostatic pickup further, some finite error still persists due to distributed diagnostic

circuit. So, an inherent pickup rejection method is attempted by shorting the return

wires of both the floating coils in double coil arrangement. Our design, shown in

figure 3.23, curtails the urge of exactly symmetric or balanced transmission line up

to the integrator circuit. Since this configuration itself works in a differential mode

with high common mode rejection, use of an expensive instrumentation differential

amplifier (IN A110) is conveniently averted and instead of IN AllO IC, a low-cost

amplifier in the integration circuit suffice our purpose. Low magnitude differential

mode signals obtained from the dual Rogowski coil configurations, are amplified by

+20dB before integration. Moreover, the impedance of the input terminal to am

plifier in the integrator circuit used with our dual coil configuration, is high and

hence, chances of interference voltage to inject any significant signal into the input


loop could be diminished.

1.0

0.8

;r 0.8 ~

JDA 0.2

0.~1_...0 ........._0 -1..._0 -2~0.......,.,30........,.40~50:--" Tlme(mS)

~1~0~0~10~~~~~~~~~ Time(...,

u 1.0

.., ... ~ 0.1

- OA

u 1.0

..., 0.1

~ 0.1 I

- O.A

10 ~ ~ ~ 10 10

llloo(ftolll

~ ~ 40 10 10

llmo(mll

78

1.0

0.1

~ 0.1

Ju

10 20 ~ 40 10 10

llmo(mll

1.0

0.1

I 0.1

J O.A

0.2

o.o ._Y ·10 0 10 20 ~ 40 10 10

n .. ,...,

Figure 3.24: Comparison of the integrated signals obtained with various coil configurations, during electrostatic pickup rejection tests, performed in the test circuit. Current signals obtained by different coil configurations are normalized (!Normalized) (dotted line) with the standard current transformer signal (solid line). Dashed lines represent the !Normalized signals obtained after rotating different coils by 180°. (Left to Right, Top) Single Coil-Floating, Single Coil-Ground, Double Coils-Parallel Tx lines. (Left to Right, Bottom) Double Coils-Twisted Tx lines, Dual Coils-Floating, Dual Coils-Ground

Our dual Rogowski coil configuration minimize the error in low magnitude plasma

current measurement to a large extent. Further suppression of noise is achieved by

connecting the shorted return wires of both the coils in dual Rogowski coil configu

ration, to a clean ground along with the coil housing. Since electrostatic pickup is

a common mode signal, it is determined by comparing the current signals obtained

in normal position and in inverted (180° rotated) position of coil(s), along with

in situ measurement of current by a standard pulse current transformer( CT). The

relative electrostatic pickup could be obtained from the ratio [1Norma!-hnverted] and

!Normal

the phase error is obtained from the ratio [ Icx1~~coil J for each configuration. Peak

values are used to calculate the total error in each configuration. Results obtained

after a meticulous, quantitative, comparative study of the errors in different com

monly used Rogowski coil configurations and our dual Rogowski coil configuration

are elucidated in the table 3.3. For comparison purpose, the current (lp) signals,


shown in figure 3.24 are normalized with the CT signal.

Rogowski coil configuration Error (in dB)

1. Floating Single coil -15

2. Floating Double coil system -20

3. Floating Dual coil system -34

4. Ground Dual coil system -45

Table 3.3: Electrostatic pickup test results, presented here, revels the best performance of our ground dual Rogowski coil configuration.

Best result is obtained with the ground dual coil configuration, checked with an

lnfinium H P54825A oscilloscope. Unfortunately, in RF environment, ground or

single ended configuration is prone to unbalanced current that adds directly to

the signal. Thus, the floating dual coil configuration is used for plasma current

measurements after proper calibration of the entire assembly shown in Fig.l, for low

frequency, low magnitude current. The sensitivity of the floating dual coil system is

25mv /Amp., obtained from the calibration graph. The signals from the diagnostic

system to the integrator circuit and henceforth, to data acquisition system, are

transferred through twisted shielded pair of cable. The dual Rogowski coil system

so constructed is capable of electrostatic pickup rejection inherently better then

other commonly used configurations. Thus, high temporal resolution (:5 lOOJLS)

and higher signal-to-noise ratio obtained lead to a proper measurement of plasma

currents.

3.6 Data Acquisition

The signals from the diagnostic systems are mostly acquired by four channel Tek

tronix TDS224 and TDS2014 model oscilloscopes which has a sampling rate of

lGs/ sec and then transferred to a PC using RS232 cable and WAVESTAR soft-

ware.


3. 7 RF noise suppression and Grounding/Shielding

The noise-filled environment created by helicon discharge can result in an un

favourable signal-to-noise ratio, impeding data collection. There are many RF noise

sources in our experimental setup. The primary source of RF noise is the high

power 20 - 40M HZ RF generator. The other sources are the antenna feedthrough,

the plasma, the helical antenna ,the grounded vacuum chamber that saturates the

ground with RF noise and the impedance matching network. While tackling any

noise related problem, we need to categorize the noise generated as:

1. an electromagnetic induction from a remote RF source, coupled to the system

due to antenna effect,

2. an electrostatic pickup due to presence of large amplitude oscillations at close

proximity to the system,

3. a conductively coupled noise.

After proper categorization, a two step measure is invoked to minimize the perva

sive RF noise; choke back the noise at the source and further suppressing the noise

from the data signal by proper isolation at the data collection point. The precau

tions/measures taken to minimize RF noise are listed below :

1. RF low pass filters are used in the data-carrying cables placed at a close prox

imity to the noise sources listed above. Filters are also placed immediately before

the data acquisition point. RF chokes, created by tightly winding the data-carrying

coaxial cable around a ferrite core, are used to eliminate the common mode noise

on the outer conductor without disturbing the actual signal.

2. Profuse use of isolation amplifiers between the diagnostics and data acquisition

system helps in successful elimination of potential ground loops and protects the

data acquisition system from voltage spikes as well.


3. Lengths of the data-carrying cables are shortened in order to prevent the ac

cumulation of noise on the data transmission line. Electronic amplifiers (wherever

needed) are placed on the diagnostics to prevent amplification of any superfluous

noise.

Proper grounding and shielding scheme in RF or microwave experiments is crucial

for improvement of signal to noise ratio, minimization of electromagnetic interfer

ence and electrostatic pickups apart from providing a safe reference point for data

acquisition system. To eliminate any common impedance coupling, all the power

supplies are connected to the diagnostic systems, capacitor bank, data acquisition

system and vacuum chamber are separately connected to the main ground in the

"star" mode. To minimize the electromagnetic induction and electrostatic pickups,

the generator and the high voltage supply are kept at a distance of ~ 20meter from

the experimental device and connected to a separate ground point. For further re

duction in noise level, all the power supplies and RF systems are properly shielded.

All the openings of metallic enclosures are kept less then tenth part of free space

wavelength.

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chapter 3 experimental set up and...

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