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Chirped Amplitude Modulation Ladar for Range and Doppler Measurements, Part II: Experimental Results*

William Ruff, Barry Stann, Mark Giza, Brian Redman, William Lawler, Keith Aliberti, and William Potter

Army Research Laboratory, 2800 Powder Mill Road, Adelphi, MD 20783

ABSTRACT As described in the companion paper, Part I, the Army Research Laboratory (ARL) is involved in a joint Army-Navy program to develop a ladar primarily for cruise missile tracking out to 10 km standoff range, but also with an adjunct mission of 3D imaging for force protection/situational awareness. To accomplish both of these missions, ARL is modifying the Intevac short wavelength infrared (SWIR) electron bombarded active pixel sensor (EBAPS) to operate as the receiver in a direct detection, chirped amplitude modulation (AM) ladar (CAMeL) architecture. For the cruise missile tracking mission, ARL will use range-Doppler processing algorithms originally developed for moving target indicator (MTI) radars with a chirped AM waveform bandwidth of less than a MHz, and with a chirped AM waveform center frequency of several tens of MHz. This regime for the center frequency keeps the expected Doppler shift due to the target’s translational motion well within the bandwidth of the EBAPS readout circuitry. By modifying software-controlled chirp-waveform parameters and sub-array region-of-interest (ROI) settings for the EBAPS readout circuitry, the proposed ladar can be electronically switched from missile tracking mode to 3D imaging mode to perform the force protection/situational awareness mission without modifications to the ladar hardware. For the initial laboratory experiments, ARL modified an existing CAMeL with an InGaAs metal-semiconductor-metal (MSM) optoelectronic (OE) mixing detector-based receiver to demonstrate the capability of this direct detection ladar architecture to measure simultaneously range and velocity using the range-Doppler processing algorithm proposed for the joint Army-Navy ladar. The results of these experiments are presented in this paper. We also discuss the modified Intevac SWIR EBAPS receiver, the chirp generator, the diode laser seeded fiber amplifier transmitter, the transceiver optical system, the control software, and their integration into the ladar breadboard for this program. We present the laboratory characterization measurement results obtained so far for these components. Keywords: Ladar, laser radar, lidar, FM-CW, chirped AM, 3D imaging, Doppler, cruise missile, tracking

1.0 INTRODUCTION The mission scenarios and requirements, as well as the ladar design parameters required to accomplish the missions for the joint Army-Navy ladar, are discussed in the companion paper, “Chirped Amplitude Modulation Ladar for Range and Doppler Measurements, Part I: System Requirements, Design Parameters, and Performance Prediction.” The joint Army-Navy ladar development will be accomplished in two phases. Currently in Phase I, ARL is designing and building a breadboard ladar test system for risk reduction and proof-of-principle static platform field tests. In Phase II, ARL will build a brassboard ladar test system that will be designed and built to meet operational goals in shipboard testing against realistic targets. The ladar is based on ARL's patented chirped amplitude modulation (AM) technique [1]. The principles of operation for the CAMeL for range and velocity measurements, the ladar performance model, and the design parameters for the Phase I breadboard and the Phase II brassboard are discussed in Part I. This paper is divided into two major parts. The first part of the paper describes a laboratory experiment designed to verify the CAMeL’s ability to measure the range and velocity of a moving target. The second part of the paper discusses the ladar breadboard, and describes the preliminary test results on the ladar components and subsystems. The top-level configuration and component improvements for the brassboard ladar are briefly described at the end of this paper in a section on future work.

* This work was jointly sponsored by the Office of Naval Research (Program Officer: CDR Keith Krapels, USNR, Ph.D.) and ARL.

2.0 TRANSLATIONAL DOPPLER DETECTION USING ARL CAMeL ARCHITECTURE 2.1 THEORY

The operation of ARL’s ladar architecture is based on frequency-modulated continuous-wave (FM/CW) radar theory. The experimental data presented in this section is obtained with a triangular, linear, frequency-chirp modulation. However, since only the up or down chirp data is processed, this experimental data is nearly identical to data that will be collected with the joint Army-Navy ladar using a sawtooth waveform. A detailed explanation of the signal processing we are using to measure the target velocity and range can be found in most books that discuss radar theory [5]. Below, we briefly discuss some of the basic operating theory for clarity.

Tchirp

fstart

fstop

τ

∆F

Freq

uenc

y

Time

fIF

TransmittedSignal

ReceivedSignal

fDoppler

Chirp Waveform: Sawtooth

TchirpTchirp

fstart

fstop

τ

∆F

Freq

uenc

y

Time

fIF

TransmittedSignal

ReceivedSignal

TransmittedSignal

ReceivedSignal

fDoppler

Chirp Waveform: Sawtooth

Figure 1. Transmitted and received signal for a system using a sawtooth up-chirp waveform.

A frequency versus time diagram of the transmitted and received signals for the joint Army-Navy ladar is shown in Figure 1. The transmitted chirp waveform, which is also used as the local oscillator (LO) waveform, is shown as a solid line in the figure with the dotted line representing the chirp waveform received from the target. The received chirp waveform is nearly identical to the transmitted waveform except that it is delayed with respect to the transmitted waveform by the round-trip time between the ladar and the target, i.e., cR2=τ , where R is the target range and c is the speed of light, and except that it is shifted in frequency due to the Doppler frequency shift induced by the target motion along the ladar line-of-sight (LOS). These effects are represented by moving the received signal left and right in the figure by the round-trip delay time, and by moving the received signal up and down in figure by the Doppler frequency shift. For the case where the target is stationary, the intermediate frequency (IF) is given by

chirpchirpIF cT

FRT

Ff ∆=

∆=

2τ , (1)

which can be obtained graphically from Figure 1 by multiplying the slope of the line by the round-trip delay time, τ , which is related to the target range as described above. In the equation, ∆F is defined as the chirp bandwidth and equals fstop - fstart, and Tchirp is defined as the chirp period, which is the duration of a single up (or down) chirp. Solving equation (1) for R expresses the range as a function of the signal's intermediate frequency, fIF.

Taking the fast Fourier transform (FFT) of this signal over a chirp period allows the signal's intermediate frequency, or equivalently, the target range to be easily measured. The frequency resolution of this measurement is equal to chirpT1

which corresponds to a measured range resolution of FcR ∆=∆ 2 [6]. Each frequency bin or cell in the FFT is also referred to as a range cell. For the case of a moving target, the intermediate frequency will be different. The average Doppler frequency shift over the chirp waveform can be calculated using the center frequency of the waveform,

cf

f centerDoppler

v2= , (2)

where 2Fff startcenter ∆+= , c is the speed of light, and v is the target LOS velocity. The instantaneous frequency difference between the transmitted and received waveforms, or intermediate frequency IFf , can be obtained from the diagram as

Dopplerchirp

IF fT

Ff ±∆

= τ . (3)

If the IF signal is measured only over a single chirp, the measured frequency will be the result of both the unknown range and unknown Doppler frequency shift, and this single measurement cannot separately determine these two frequencies. Since the complex amplitude of the FFT used to measure fIF during each chirp will oscillate at the Doppler frequency over repeated chirps, simply taking a second FFT at each range cell across a number of repeated chirps, maps the data into a range-Doppler space with two frequency axes (one corresponding to range and one corresponding to Doppler frequency shift) in which the target's range and Doppler frequency shift can be measured simultaneously. Using this measurement technique, the Doppler frequency shift measurement resolution is limited by the duration of the measurement over multiple chirps, and is given by DopplerT1 or chirpNT1 , where N is the number of chirps used to sample the Doppler frequency. Note that the resolution of the range dependent frequency measurement over a single chirp is limited by the duration of a single chirp. 2.2 EXPERIMENTAL MEASUREMENT SETUP

Figure 2 is a basic electronic block diagram of the ladar test setup for the initial range-Doppler data collection experiments. A trigger signal generated from a field programmable gate array (FPGA) initiates the generation of a triangular chirp signal that serves as both the laser modulation and local oscillator (LO) signal. The chirp signal in our case is generated by a direct digital synthesizer (DDS) and is, therefore, programmable by the user. The chirp signal may have a start frequency and a stop frequency in the tens-to-hundreds of megahertz (an up-chirp is shown at the top of Figure 2). To modulate the laser intensity, the chirp signal is fed into a wideband RF power amplifier whose output is summed with a DC current in a bias tee to provide a modulated current drive for a semiconductor diode laser. For the experiments described in this paper, a 1.55 µm wavelength semiconductor laser operating at 1 mW average power was intensity modulated to 2 mW peak power by the RF amplifier. The divergent laser beam from the semiconductor laser is collected with a collimating lens and focused to a small spot that is aligned to intersect the detector field-of-view (FOV) at the target. A portion of the laser beam reflected from the target is collected by the receiver optics and imaged onto a single InGaAs metal-semiconductor-metal photo detector (MSM-PD) used as an optoelectronic mixer (OEM) [8]. The chirp signal is applied as an LO signal to modulate the responsivity of the MSM-PD. Since the output current of the MSM-PD is determined by the instantaneous responsivity multiplied by the instantaneous received laser power, the MSM-PD performs the same function as an RF mixer, multiplying the signals together. Therefore, the output current from the MSM-PD is modulated at the intermediate frequency (IF) as shown in Figure 1.

Figure 2. Block diagram of the chirped AM ladar setup used in the range-Doppler laboratory experiments.

The photocurrent IF waveform from the detector is converted into a voltage signal using a transimpedance amplifier (TZA). Following the TZA are two amplifiers, one with a fixed gain and the other with a computer programmable gain, that are used to set the frequency response of the receiver, and to buffer the signal for the subsequent analog-to-digital (A/D) converter. The amplifier chain has a computer-controlled equivalent impedance of 440 MΩ to 44 GΩ, and a computer controlled bandwidth of 50 kHz to 120 kHz with a cut-on frequency beginning at 2 kHz. The A/D samples the voltage from the detector, and transfers the voltage data into computer memory. The detector voltage is measured periodically over each up and down chirp at a rate adequate to sample the highest expected IF. The resulting data set contains the IF waveform that includes all of the magnitude, range, and phase information derived from the light reflected from all scatterers in that pixel [1]. The entire chirped AM ladar setup utilized in the range-Doppler laboratory experiments was previously fully characterized using a GaAs MSM-PD OEM linear detector array [7]. 2.3 EXPERIMENTAL RESULTS A white target board, mounted on a metallic carrier, was guided on a metal track to provide a target for the Doppler detection. A string tied to the front of the carrier was used to move the target towards the ladar, and a spring was used to retract the target to its original starting position. A photograph of this target apparatus is shown in Figure 3, with the surrounding background clutter faded for clarity. The initial target position was located approximately 8 m from the ladar, and the range of translational motion was 109 cm, defined by two c-clamp stops. The chirp generator was programmed to output 128 up-down chirps at an 800-MHz center frequency with chirp bandwidths (∆Fs) of 75 and 37.5 MHz at repetition rates of 49 and 98 Hz, respectively. The 800-MHz center frequency was chosen so that a number of Doppler cycles could be measured with the small length of target travel available in the laboratory. The target, when pulled the full length of travel, traverses approximately 6 Doppler cycles at 800 MHz. A 193-ns delay line was inserted between the chirp generator and the LO port on the MSM-PD OEM detector to artificially move the target out to a range far enough for the IF to be in the passband of the receiver electronics. The up and down chirp data were separated during the processing, and are nearly identical except for the change in the sign of the target’s Doppler shift. For brevity, only the down-chirp data is shown below.

Figure 3. Photograph of the target apparatus for the translational Doppler frequency shift measurements.

Figure 4. Time series of Hanning-weighted FFT of the IF

signal from a stationary target. Figure 5. Real part of Hanning-weighted FFT of the IF

signal from a stationary target.

Figure 4 is a plot of the time series data collected during a down-chirp for a stationary target, and Figure 5 is a plot of the real part of its corresponding Hanning-weighted FFT, collected at a ∆F of 75 MHz and a chirp repetition rate of 49 Hz. From Figures 4 and 5, the range to the target is between 10 and 11 range cells, found by counting the number of cycles in the time series data or by the frequency bins containing the signal in the FFT, which corresponds to an artificial range (including the added delay) of between 20 and 22 meters. Figure 6 is a range-Doppler image produced from 64 down-chirps collected over a 2.6 second interval by displaying the magnitude of the complex FFT taken at each range cell for all 64 down-chirps. This image simultaneously displays the target range and Doppler shift corresponding to the target's translational motion, which, for this stationary target, falls on the 0 Hz Doppler frequency line at the center of the image. Figure 7 is a plot of the real part of the Hanning-weighted FFT of the 10th range cell for each of the 64 down-chirps taken when the white target was pulled towards the ladar, and then allowed to retract back to its original position, with the chirp generator set to a ∆F of 75 MHz and a chirp repetition rate of 49 Hz. It was very difficult to move the target at a constant velocity using the pull string, so the target accelerated during the image collection. This is visible in the range-Doppler image, shown in Figure 8, as a smearing of the target signal in the Doppler dimension. The target moved the full 109 cm during the first 35 chirps and, from Figure 7, it is apparent that it transitioned (stopped and began retracting) in about 6 Doppler cycles (covering the full length of travel), as expected. The target transitioned only 3 Doppler cycles

before the end of the data collection, indicating that the target had not returned to its original position by the end of the data collection. The decrease in the amplitude of the oscillation as the target moved from the back to the front of the rail indicates that the target may have moved through the entire 2-m range cell and into the adjacent range cell.

To improve the previous experiment, we set the chirp generator to chirp at a faster 98 Hz repetition rate to sample the target’s more constant motion while it traveled through the middle portion of the rail. We also changed the chirp ∆F to 37.5 MHz, which corresponds to a ∆R of 4 m, to keep the moving target in one range cell. Doubling the length of the range resolution cell also has the effect of moving the target signal from the 10th range cell to the 5th range cell. Figure 9 is a plot of the real part of the Hanning-weighted FFT of the 5th range cell for each of the 64 down-chirps taken when the white target was pulled towards the ladar. The target appears to be accelerating for the first 10 chirps, and then moves at a more or less constant velocity for the duration of the measurement. This is indicated in the range-Doppler image, shown in Figure 10, where the target peak appears in the 5th range cell and in the +5 Hz Doppler frequency cell, with some smearing in the Doppler dimension due to the initial acceleration. Figure 11 is a plot of the real part of the Hanning-weighted FFT of the 5th range cell for each of the 64 down-chirps taken when the white target was retracted to its original position. As before, the target appears to accelerate in the beginning of the data collection and then moves at a more or less constant velocity until the end of the data collection, indicated in the range-Doppler image shown in Figure 12. Even though the real parts of the FFTs for both cases of target motion look similar, Figures 9 and 11, the complex FFT is able to distinguish differences between receding and approaching targets, Figures 10 and 12, as expected.

Figure 6. Range-Doppler image of stationary target.

Figure 7. Real part of Hanning-weighted FFT of Range

cell 10 for target moving back and forth. Figure 8. Range-Doppler image of target moving back

and forth.

Figure 9. Real part of the Hanning-weighted FFT of the

5th range cell for approaching target. Figure 10. Range-Doppler image of approaching target.

Figure 11. Real part of the Hanning-weighted FFT of the

5th range cell for receding target. Figure 12. Range-Doppler image of receding target.

3.0 BREADBOARD LADAR HARDWARE AND CONFIGURATION

The top-level diagram of the breadboard ladar transceiver is shown in Figure 13, and a photograph of its current implementation is shown in Figure 14. In the photograph, a Sensors Unlimited InGaAs PIN photodiode FPA camera is used in place of the Intevac EBAPS camera for optical characterization and alignment of both the laser and receiver optics while the EBAPS is in the RF modulation testbed. As seen in Figure 14, the RF-modulation circuitry has not yet been incorporated into the ladar transceiver assembly since it is still under development in the RF-modulation testbed.

Figure 13. Top-level diagram of the breadboard chirped AM ladar transceiver configuration.

Figure 14. Photograph of Army-Navy ladar transceiver breadboard as currently configured.

We plan to mount the breadboard ladar transceiver on a heavy-duty tripod for outdoor testing. As indicated in the diagram, we located the laser and fiber amplifier off of the transceiver breadboard to decrease its weight for mounting on the tripod. Currently, as shown in the photograph, the picomotor drivers and controllers are mounted on the transceiver breadboard, however, we may relocate them off of the breadboard before the outdoor testing to further decrease the weight. The JMI motorized focus controller, the operator station control computer/electronics, and the data capture, processing, storage, and display computer/electronics will all be located off of the transceiver breadboard. A discussion of each of the major subsystems is presented in the following sections.

ReceiverOptics

IntevacEBAPS

High VoltagePower Supply

LOModulator

DDS 1(Delayed)

MasterControl

DDS 2(Master)

TransmitterOptics

FiberAmp

(Manual control)

TemperatureAnd

Laser Power Control

SeedLaser

EthernetHub

Control Computer

Digital Compass

Ni DAQPad6508

Ni DAQPad6020E

HandControl

20 dB 20 dB

Det.

Focus Adjustment (I/O)

LVDSGate in

LM9638Control (I/O)(parallelport)EPIX Digital Data Out(68 pin SCSI cable)

RF in

High Voltage in

High Voltage out

High Voltage + RF out

Bias V Set inHV Set in

Bias monitor outHV monitor out

Address inData in

Address outData out

MasterClock

Clk inGate outVsync inEPIX trigger out

DDS 2 DDS2Program StartOut Out

DDS 1 DDS 1Program StartOut Out

Vsyncout

Clk inProgram inStart in

Clk inProgram inStart in

RF out

RF out

Data &Ctrl (I/O)(RS-232)

New Focus8752(Ethernet)Fiber in

Fiber out

Fiber in

Fiber out

RFin

TemperatureControl In

CurrentControl In

T out I out

USB

USBEPIX EPIXData CtrlIn In

ParallelPort

SerialPort

Ethernet

IEEE-488

IEEE488

Start Finished

Mclk/8out

To Display Computer

TP

TP TP

ReceiverOptics

IntevacEBAPS

High VoltagePower Supply

LOModulator

DDS 1(Delayed)

MasterControl

DDS 2(Master)

TransmitterOptics

FiberAmp

(Manual control)

TemperatureAnd

Laser Power Control

SeedLaser

EthernetHub

Control Computer

Digital Compass

Ni DAQPad6508

Ni DAQPad6020E

HandControl

20 dB20 dB 20 dB20 dB

Det.

Focus Adjustment (I/O)

LVDSGate in

LM9638Control (I/O)(parallelport)EPIX Digital Data Out(68 pin SCSI cable)

RF in

High Voltage in

High Voltage out

High Voltage + RF out

Bias V Set inHV Set in

Bias monitor outHV monitor out

Address inData in

Address outData out

MasterClock

Clk inGate outVsync inEPIX trigger out

DDS 2 DDS2Program StartOut Out

DDS 1 DDS 1Program StartOut Out

Vsyncout

Clk inProgram inStart in

Clk inProgram inStart in

RF out

RF out

Data &Ctrl (I/O)(RS-232)

New Focus8752(Ethernet)Fiber in

Fiber out

Fiber in

Fiber out

RFin

TemperatureControl In

CurrentControl In

T out I out

USB

USBEPIX EPIXData CtrlIn In

ParallelPort

SerialPort

Ethernet

IEEE-488

IEEE488

Start Finished

Mclk/8out

To Display Computer

TP

TP TP

Figure 15. Control diagram for the breadboard ladar.

3.1 FM CHIRP GENERATOR AND CONTROL ELECTRONICS Figure 15 is a connectivity diagram of the major components of the breadboard ladar along with the interconnecting control signals between the components. We are using a personal computer (PC) to setup and control the majority of the components during a data collection. We are using an FPGA to handle the critical timing signals, such as triggering the RF chirp generation, and triggering the gate for the EBAPS high-voltage and the RF-modulation. The FPGA will also be used to monitor the vertical sync signal from the EBAPS camera and the frame grabber status during a data collection. The PC will download the data collection parameters to the FPGA, and at the operator’s discretion, send a start signal to the FPGA to initiate a data collection. The steps required for a basic data collection are outlined in Figure 16. We are using two chirp generators to artificially move the center of a range swath in both the imaging and tracking modes of operation. This is necessary since the range swaths, limited by the maximum IF which can be sampled with the EBAPS read-out, for both the tracking and imaging modes of operation may be smaller than the maximum operational ranges. For the imaging mode in particular, the range swath is much smaller than the maximum operational range. By using two chirp generators, one to modulate the laser intensity and one as the LO, the start of the LO chirp signal can be delayed by some pre-set time after the start of the laser modulating chirp signal resulting in a shorter round-trip delay and, thus, a smaller IF. The starting phase, as well as the relative timing of the start of both waveforms, must be carefully controlled for the ladar to operate properly using this technique. Therefore, the two chirp generators are required to generate frequency-agile, sinusoidal wave signals with a known phase and time relationship between them. This has been accomplished by employing two Analog Devices AD9858 Direct Digital Synthesizer (DDS) chips synchronized together. This synchronization is implemented with the use of an FPGA that controls the frequency update signals to the DDSs. A 1GHz master clock is coincident to the pair of DDSs that in turn develop a 125 MHz clock to run the FPGA, which develops the synchronizing update signals. Various programmable registers and counters in the FPGA control the

start frequency, frequency step size, dwell time, and delay attributes of the DDS. These parameters are also downloaded from the PC. The FPGA further synchronizes the chirp generator with the EBAPS and the data acquisition system. Also provided are low pass filters and amplifiers on the outputs of each DDS, PIN diode switches for optional RF gating, and, as a system check, a mixer to combine the output of each DDS to form an intermediate frequency that is based on the relative delay and sweep time of the DDSs. We have also incorporated two PC-controlled, RF switches to allow the ladar to operate with only one chirp generator for testing purposes. A photograph of the two DDSs and the FPGA controller on a custom printed circuit board mounted in the chirp generator control box is shown in Figure 17.

Figure 16. Data collection control.

Figure 17. A photograph of the two DDSs and the FPGA controller on a custom printed circuit board mounted in the chirp generator controller box. (Note: the DFB diode seed laser (see the next subsection) in its temperature stabilization mount (the black box in the rear left corner with the yellow fiber pigtail) is also mounted in this box.)

3.2 TRANSMITTER We are using an IPG Photonics erbium-doped fiber amplifier (model EAD-10) as our laser source in the breadboard ladar. The EAD-10 has an optical power gain of 40 dB, and is capable of generating a maximum output power of 10 W at 1.55 µm. The IPG fiber amplifier has a fiber-optic pigtail with a collimating microlens at its output to provide a near diffraction limited beam with a 1.6 mm waist diameter and a beam divergence of 1.357 mrad. Our initial plan was to use a ~ 1.5 mW input from a fiber-coupled Mitsubishi Fabry-Perot (FP) diode laser (model ML976H6F) as a seed for the optical amplifier. We found, however, that when we intensity modulated this seed laser with the RF chirp signal, we also modulated the wavelength of the fiber amplifier's output outside of the receiver optical filter's passband (3 nm full-width at half-maximum (FWHM) planned for the brassboard ladar) as shown in Figure 18. Since this is not acceptable, we switched our seed source to a Mitsubishi ML925B11F distributed feedback (DFB) diode laser, and verified that the fiber amplifier's output maintained its wavelength spread within the 3 nm passband during the intensity modulation of this seed source as shown in Figure 19. The fiber amplifier's output with the DFB laser diode seed was able to attain a maximum modulation depth of around 70% over a frequency range of 25 to 1000 MHz before the onset of clipping as shown by the plot of a typical data set in Figure 20. This figure also shows that the sinusoidal waveform of the intensity modulation was much less noisy with the DFB laser seed than with the FP laser seed. The fiber amplifier output is routed from the amplifier to a fiber-positioner, located on the transceiver breadboard, and its output beam is directed to a Galilean beam expander using two turning mirrors. The two turning mirrors are mounted on New Focus model 8852 motorized corner mounts to provide both translational and angular alignment control using either a hand-held remote or the PC. The Galilean beam expander uses a -8 mm focal length, 8 mm diameter plano-concave input lens and a 500 mm focal length, 150 mm diameter plano-convex output lens to provide a beam expansion ratio of 62.5X that yields a 10.6-cm output beam diameter and a best case transmitter beam divergence of 21.7 µrad. The beam expander input lens is mounted on a picomotor equipped New Focus model 9064 XYZ translation stage to allow remote alignment and focus. The output lens is mounted on a height adjustable post and translation stage for initial coarse alignment. Both lenses are anti-reflection coated for operation at 1.55-µm wavelength.

Figure 18. Optical spectrum of fiber amplifier's output with the ML976H6F Fabry-Perot seed laser with (right) and

without (left) applied RF intensity modulation.

Figure 19. Optical spectrum of fiber amplifier's output with the ML925B11F DFB seed laser with (right) and without

(left) applied RF intensity modulation.

Fabry-Perot (FP) Laser Distributed Feed-Back (DFB) Laser

Ground (Zero) Ground (Zero)

Figure 20. Intensity of the fiber amplifier's output with the ML976H6F Fabry-Perot seed laser (left) and with the

ML925B11F DFB seed laser (right) with applied RF modulation.

We were able to test the transmitter using a water tower target located 1.7 km from our building at the ARL Adelphi Laboratory Center. Figure 21 is a photograph of the top of the water tower, containing a number of resolution test targets for imager characterization that are part of the ALOT facility [9]. Figure 22 is a montage of three images showing the focused laser spot on the tower, along with some undesirable white spots due to defective pixels in the Sensors Unlimited camera's FPA. The top and leftmost images show the laser focused to its smallest size at two different output power levels, and the right-most image shows the laser diverged to the size planned for the ladar 3D imaging mode. Note that the higher power focused spot appears much larger than that of the lower power focused spot only because the higher power spot image saturated a significant region of the FPA. The 1.5 m diameter sphere (overlayed with a resolution target) located on top of the water tower can be used to determine an approximate image scale. The picomotor mounts were not available for this test. Therefore, both of the lenses in the beam expander were mounted on manually adjustable XYZ translation stages which were moved to control the laser spot size and position.

Figure 21. Photograph of the ALOT target set located on top of the water tower target [9].

Figure 22. Laser spot measurements on the ALOT water tower at 1.7 km range. The laser spot in the top image is not saturated at 100 mW output power and subtends approximately 2x2 pixels which corresponds to a 0.134 m spot size at the target. This compares well with the calculated, diffraction-limited focused spot diameter for a Gaussian beam with a 10.6 cm beam waist of 0.138 m at 1.7 km range. The spot in the 3.5 W output power image appears much larger due to camera saturation so that it does not provide a useful measure of spot size. 3.3 RECEIVER The receiver optical system consists of a Meade LXD55 SN-10 Schmidt-Newtonian telescope with an input aperture of 25.4 cm (10") diameter with an 8 cm diameter central obstruction, and a 1.016 m focal length (F/4). A Schmidt corrector plate, anti-reflection coated for the visible wavelength region, covers the input aperture. The manual focusing tube for this telescope is replaced with a JMI model NGF55DM motorized focuser for remote control of the receiver focus. A choice of either a 10 nm or 50 nm FWHM bandwidth narrowband optical filter centered at 1550 nm is mounted in the focuser tube in the optical path before the receiver sensor for the ladar measurements. A visible black and white imaging sensor, National Semiconductor model LM9638, was mounted to the JMI focuser and used to acquire imagery to test the optical quality of the telescope. This imager is identical to the CMOS readout used by the EBAPS camera except that it is front side illuminated in this case rather than backside bombarded as in the EBAPS. The LM9638 can be set to read-out only specified pixels over a region of interest in the focal plane (windowing) at vastly increased frame rates. Adjacent pixels in the window can also be summed together (binned) to further increase the frame rate at a cost of decreased resolution. The sensor has 1032 x 1288, 6µm square, active pixels in normal operation which is decreased to 516 x 644, 12 µm square active pixels when the 2 x 2 on-chip binning mode is enabled. The window size and frame rates we will use for the breadboard ladar tests are listed in Table 1. Figure 23 is an image of a resolution target located on top of the water tower (see Figure 21) acquired over a window of 128x128 binned-pixels at a frame rate of approximately 50 Hz. The image was acquired about midday with no optical filter in place and with a long integration time to increase the pixel signal levels (which correspondingly decreased the camera frame rate). From a geometric optics approximation, the

expected pixel resolution at the range of the tower is calculated to be approximately 2 cm, which, from the figure, closely agrees with the smallest resolvable bar pattern in the image of about 2.3 cm.

Table 1. Breadboard ladar window sizes and frame rates.

Window Size Frame Rate (F/s) 1024x1024 21

512x512 65 256x256 200 128x128 540 64x64 1,300 32x32 2,900 20x20 4,900 16x16 6,200

Figure 23. Image of resolution target on top of water tower at 1.7 km using LM9638 sensor. The breadboard ladar receiver sensor is Intevac, Inc.'s Electron Bombarded Active Pixel Sensor (EBAPS) with a Transferred Electron (TE) InGaAs photocathode [10]. The EBAPS sensor provides good quantum efficiency (20% to 38%) in the short wavelength infrared (SWIR) band from 950 nm to 1650 nm at optical gains of up to 300. The EBAPS electron target/readout sensor is a silicon National Semiconductor LM9638 CMOS FPA that is electron bombarded on the backside to provide nearly 100% fill factor. The electron gain of the EBAPS tube is nominally 150 at a cathode-to-anode operating voltage of -2 kV, but it can be operated at lower and higher gains simply by changing the cathode-to-anode voltage. The photocathode also has a Schottky barrier deposited on the vacuum side of the photocathode. The photocathode quantum efficiency depends on the Schottky barrier bias voltage, which is nominally biased at a voltage of about +3 V to +6 V. By changing the Schottky barrier bias voltage by approximately 1 V, the EBAPS quantum efficiency can be changed from near zero to its maximum level, as shown in Figure 24. The LM9638's windowing function not only provides the high frame rates required for the chirped AM ladar, but also enables electronic alignment and adjustment. Since the pixel window can be placed at any location in the FPA, the receiver's FOV can be moved electronically by moving the pixel window in the image plane. This allows electronic control of the alignment to center the FOV on a target. It also enables electronic adjustment of the inclination angle between the transmitter's optical axis and the receiver's effective optical axis defined by the location of the center of the

pixel window. This inclination angle adjustment is necessary to compensate for the parallax between the transmitter and the receiver as the target moves in range. For the breadboard tests, the operator will manually point the tripod mounted transceiver to place the laser spot on an extended target. Then, electronically moving the pixel window and controlling the JMI focus adjuster on the receiver enables electronic fine parallax compensation and focus control.

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A key to using the EBAPS sensor in the ARL ladar architecture is our ability to efficiently multiply the incoming intensity modulated light with the LO chirp signal. Changing the response of the EBAPS tube, either by changing the Schottky barrier voltage and/or by changing the cathode-to-anode voltage, can be used to perform this multiplication. In Phase I, ARL has been characterizing the modulation of the tube response at the RF frequencies using both of these methods. Under a Cooperative Research and Development Agreement (CRDA) between ARL and Intevac, Intevac is providing EBAPS sensors and supporting hardware to ARL for laboratory modulation testing, and for use in the chirped AM ladar breadboard. ARL is replacing the low repetition rate, gateable high-voltage power supply that Intevac supplied with the EBAPS sensor with a GBS Micro Power Supply gateable high-voltage power supply capable of gating the cathode-to-anode voltage at up to 7 kHz gate repetition rate. Figure 25 shows a block diagram and a photograph of the RF-modulation testbed used to characterize the EBAPS under both types of RF-modulation. We are using two phase-locked RF generators, set to frequencies that are offset from each other by a few Hz to a few tens of Hz, to modulate both the response of the EBAPS sensor and the intensity of a 1.55 µm wavelength diode laser. The intensity modulated laser illumination is collimated and projected onto a target approximately 3 meters from the setup. The laser light scattered from the target is imaged onto selected regions of the EBAPS sensor. The cathode-to-anode voltage applied to the EBAPS sensor is also gated with a pulse generator synchronized to the vertical sync signal output from the CMOS readout. The camera output is captured using an Epix, Inc. frame grabber. The data is stored to disk for analysis. This setup allows great flexibility in characterizing the EBAPS sensor at a variety of modulation and offset frequencies, and at different applied voltage biases and gate times. An RF switch was added prior to the RF amplifier in the setup (Figure 25 (a.)) to alleviate data capture problems due to RF coupling into the CMOS readout. Figure 26 is a schematic of the circuit used to characterize the Schottky barrier modulation. The purpose of this circuit is to modulate the quantum efficiency of the EBAPS sensor by modulating the applied Schottky barrier bias voltage. From DC characterization data provided by Intevac and verified at ARL (Figure 24), modulating the barrier bias by approximately 1 V will effectively modulate the quantum efficiency by nearly 100%. The only apparent drawback to modulating the barrier voltage is that the measured capacitance of Schottky barrier is on the order of a few thousand pico-Farads (pF), thus presenting a difficult RF impedance. To drive the Schottky barrier, we use a low-output impedance field-effect-transistor (FET) coupled to a low-impedance coaxial cable that is then attached to the EBAPS cathode and Schottky barrier contacts. The coaxial cable is attached to the EBAPS as a transformer, thereby effectively lowering the FET output impedance even further to enhance the match with the Schottky barrier impedance. We have

operated this experiment over a number of frequencies spanning 20 MHz to 400 MHz, and our initial results indicate that modulating the barrier voltage does not effectively modulate the response of the tube. In fact, this modulation may be detrimental to the tube lifetime. During testing, dark areas in the interiors of focused laser spots producing annular spot shapes were observed. These dark areas were in regions corresponding to areas of high laser irradiance, and are indicative of “electron starvation” in the photocathode [11]. Since the irradiance threshold for the onset of the “electron starvation” phenomenon decreases with decreasing Schottky barrier bias voltage, we hypothesize that the time constant for the photocathode to recover from the “electron starvation” acts as a low-pass filter when attempting to modulate the Schottky barrier voltage at high frequencies.

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Figure 25. (a.) Block diagram and (b.) photograph of EBAPS RF-modulation testbed.

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Figure 26. Schottky barrier modulation circuit.

Based on these results, we decided to modulate the gain of the tube rather than the photocathode response. Figure 27 is a schematic of the circuit used to modulate the gain of the EBAPS sensor by modulating the cathode-to-anode voltage, similar to the method we reported in 1999 with a conventional GaAs photocathode image intensifier tube based ladar receiver [12]. The difficulty with this modulation technique is that swings of hundreds of volts are required to modulate the gain with a high modulation index. Fortunately, the cathode-to-anode capacitance is only 15 pF, and is much easier to drive with conventional RF amplifiers than is the Schottky barrier. As shown in the Figure 27, the RF modulation is coupled from a high power amplifier to the cathode and Schottky contacts on the EBAPS using a standard coaxial transmission line.

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Figure 27. Cathode-to-anode high-voltage modulation circuit.

We have collected cathode-to-anode high-voltage RF modulation data on the EBAPS camera over a frequency range of 100 to 400 MHz. A sample data set from these experiments is shown in Figures 28 through 30. Figure 28 is an intensity plot of a 64x64 pixel window of the laser spot taken with the EBAPS sensor during a modulation experiment. During the data acquisition, the two RF generators were set to a center frequency of 125 MHz with an offset frequency of 20 Hz. The gate width of the high-voltage pulse was adjusted until the highest intensity pixel was near saturation. Figure 29 is a plot of the intensity of a pixel located in the laser spot taken over multiple contiguous frames with the camera operating at a frame rate of 100 Hz. The pixel intensity clearly oscillates at the offset frequency as expected, but with some variation in amplitude. Figure 30 is a plot of the magnitude of the FFT of the intensity data shown in figure 29. Again, the peak of the oscillation is clearly visible in the FFT data at the offset frequency. The noise floor also appears fairly uniform, with no major interference spikes present. To date, we have experimented with several high-power RF amplifiers up to 12 W, and have achieved from 15% to 22% modulation index over a frequency range of 100 to 250 MHz. We have ordered a 50 W commercial RF amplifier, which operates over a frequency range of 20 to 400 MHz, to enable higher modulation depths.

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Figure 28. Intensity plot of 64x64 pixel window showing laser spot.

Figure 29. Pixel intensity variation over contiguous frames.

Figure 30. Magnitude of the FFT of the data from figure 29.

4.0 CONCLUSION In this paper, we have presented experimental verification of simultaneous range and line-of-sight translational velocity measurement using a direct-detection, chirped amplitude modulation ladar (CAMeL). We are currently building a breadboard ladar based on the ARL CAMeL using an Intevac EBAPS sensor as the receiver. This breadboard ladar will be used in proof-of-principle field tests at extended ranges to demonstrate the ability of ARL's CAMeL to measure target velocity and range for the cruise missile tracking mission in a joint Army-Navy program. This breadboard ladar will also operate in an imaging mode to form 128x128 pixel, 3D imagery of targets to demonstrate the capability of ARL's CAMeL to perform the force-protection/situational awareness mission. Performance modeling results presented in the companion paper (Part I), indicate that the breadboard performance will be adequate to measure the range and velocity of a cruise missile size target at ranges up to 3 km, and to produce 128x128 pixel, 3D imagery at ranges up to 1 km. The Phase I field tests to demonstrate these capabilities are planned for this fall and winter. We have completed construction of the breadboard laser transmitter, and have demonstrated near diffraction-limited operation with a beam divergence of about 22 µrad. We have completed construction of the breadboard chirp generator capable of generating two frequency-agile, sinusoidal wave signals with a known phase and time relationship between them. We have performed experiments

on modulating the response of the EBAPS at RF frequencies which have shown modulation depths over the frequency band of interest of up to 22%.

5.0 FUTURE WORK The EBAPS RF modulation experiments are continuing in order to improve the modulation depth and uniformity of response over increasing bandwidths. Integration of all of the components into the Phase I ladar breadboard is expected to be completed in time to begin characterization and local outdoor testing at the ARL Adelphi Laboratory Center (ALC) in the early fall. Longer range field tests are planned for the late-fall/early winter, culminating with a demonstration for Office of Naval Research (ONR) program manager and the Naval Research Laboratory (NRL) project manager in the late winter. Lessons learned in the development and testing of the Phase I breadboard ladar will be incorporated into the design, development, and testing of the Phase II brassboard ladar. Figure 31 is a conceptual diagram of the ARL CAMeL ladar brassboard that we will construct in the Phase II program. The brassboard will use a fiber-amplified, pulsed-laser transmitter capable of 1 kW peak power to increase the maximum effective range of the breadboard against cruise missile targets to at least 10 km. The new transmitter will also increase the maximum effective range for 3D imagery to at least 3 km. We plan to continue the modulation studies on the EBAPS, and expect to increase the modulation depth to at least 30%. We are also investigating the use of an electron bombarded metal-semiconductor-metal (EB-MSM) focal plane array to achieve simultaneously wideband mixing, high modulation index, high readout rates, and low-noise gain.

Figure 31. Conceptual diagram of the Phase II brassboard ladar prototype.

6.0 ACKNOWLEDGMENTS The authors gratefully acknowledge the support received for this project from the ONR program manager, CDR Keith Krapels, Ph.D., not only in the form of financial support, but also in the form of information on the missions and engagement scenarios. We also acknowledge the contributions of the NRL project manager, Dr. James Waterman, in providing technical information on the operation of the Distributed Aperture System Infrared Search and Track (DAS-IRST) sensor, which will be the acquisition and cueing sensor for the brassboard ladar in Phase II. In addition, we thank Dr. Waterman and his group at NRL for the technical feedback on the ladar conceptual design, the optical environment in the marine boundary layer, and the performance model results. The authors are also grateful to Daniel Hosek of the Army's Night Vision and Electronic Sensors Directorate (NVESD) for loaning us an EBAPS test set so that we could become familiar with the control and readout interfaces while waiting for Intevac to build our EBAPS test set. We also thank Dr. Bradley Schilling of NVESD for providing technical information on the EBAPS test set and tubes. We are grateful to Dr. Ronald Driggers of NVESD for his technical and programmatic suggestions as a reviewer of this project for ONR. We thank Verle Aebi and Phillip Arcuni of Intevac, Inc. for providing the EBAPS test set and spare EBAPS tubes for our modulation experiments and for the breadboard receiver. The authors also gratefully acknowledge Phillip Arcuni for providing technical information on the EBAPS test set and tubes, and for answering our many questions. We also thank Charles Kozlowski of CSK Consulting for his role as liaison between ARL and Intevac. We thank Dr. Neelam

Gupta of ARL for loaning us the Sensors Unlimited SWIR band camera for the optical alignment and characterization of the breadboard ladar transceiver.

7.0 REFERENCES 1. B. L. Stann, W. C. Ruff, Z. G. Sztankay, “Intensity-modulated diode laser radar using frequency

modulation/continuous wave ranging techniques,” Optical Engineering, Vol. 35, no. 11, 1996, pp. 3270-3278. 2. W. Ruff, J. Bruno, S. Kennerly, K. Ritter, P. Shen, B. Stann, M. Stead, G. Sztankay, M. Tobin, “Self-mixing

detector candidates for an FM/cw LADAR architecture,” Proc. SPIE, Vol. 4035, 2000, pp. 152-162. 3. B. L. Stann, et al., “Component research for the MDSS LADAR,” Proc. of Advanced Sensors Consortium, 2000, pp.

25-32. 4. P. H. Shen, et al., “Interdigitated finger semiconductor photodetector for optoelectronic mixing,” Proc. SPIE, vol.

4028, 2000, pp. 426-435. 5. Jerry L. Eaves and Edward K. Reedy, Principles of Modern Radar, New York: Van Nostrand Reinhold Company,

1987. 6. E. Oran Brigham, The Fast Fourier Transform and Its Applications, New Jersey: Prentice Hall, 1988. 7. W. Ruff, K. Aliberti, J. Damman, M. Giza, P. Shen, and B. Stann, “Performance of an FM/cw prototype ladar using

a 32-element linear self-mixing detector array”, Proc. SPIE, Vol. 5086, 2003, pp. 58-69. 8. K. Aliberti, H. Shen, W. Ruff, B. Stann, P. Newman, F. Semendy, M. Stead, R. Mehandru, and F. Ren, “Improved

optoelectronic mixing of InAlAs/InGaAs interdigitated-finger metal-semiconductor-metal photodetectors,” Proc. of IEEE International Conference on Sensors, 2003, pp. 1021-1025.

9. Mikhail A. Vorontsov, Gary W. Carhart, Matt Banta, Thomas Weyrauch, John Gowens II, and John C. Carrano “Atmospheric laser optics testbed (A_LOT): atmospheric propagation characterization, beam control, and imaging results”, Proc. SPIE, Vol. 5162, pp. 37-48,

10. Kenneth A. Costello, Verle W. Aebi, Gary A. Davis, Ross A. La Rue, and Robert E. Weiss, "Transferred electron photocathode with greater than 20% quantum efficiency beyond 1 micron," Proc. SPIE Photodetectors and Power Meters II, Vol. 2550, 1995.

11. George J. Yates, Thomas E. McDonald Jr., David E. Bliss, Stewart M. Cameron, Kenneth Greives, and Fred J. Zutavern, “Near infrared (NIR) imaging techniques using lasers and nonlinear crystal optical parametric oscillator/amplifier (OPO/OPA) imaging and transferred electron (TE) photocathode image intensifiers”, Proc. SPIE High-Speed Imaging and Sequence Analysis III, Vol. 4308, April 2001.

12. Charles Garvin, Maj. R. Chris Deluca, Arnold Goldberg, Steven Kennerly, Gary Green, Herb Pollehn, Greg Sztankay, Barry Stann, Bill Ruff, Dale Robinson, Mark Giza, Scott Sarama, Herb Schaake, Gerry Padgett, Paul Kisselberg, Lynn Mears, Dan Haver, Aaron Castillo, Cindy Pettitt, Neelam Gupta, Milton Gottlieb, Louis Denes, Boris Kaminsky, and Peter Metes, "Multi-Domain Smart Sensor Testbed Field Operation at Ft. A.P. Hill," Proc. 3rd Annual Fedlab Symposium, Advanced Sensors Consortium, 1999, pp. 15-21.