design of a multiband microstrip differential phase ... · 2-6 phase shifter designs ......

Design of a Multiband Microstrip

Differential Phase Shifter for Wireless Systems

Ali Reza Aboofazeli

A thesis presented to Ottawa-Carleton Institute for

Electrical and Computer Engineering

in partial fulfillment to the thesis requirement for the degree of

MASTER OF APPLIED SCIENCE

in

ELECTRICAL ENGINEERING

University of Ottawa

Ottawa, Ontario, Canada

October 2016

© Ali Reza Aboofazeli, Ottawa, Canada, 2016

ii

Abstract

This thesis investigated the design of a compact multiband differential reflective phase

shifter based on slot-coupled coupler layout. Such phase shifter configuration is formed

of a hybrid coupler with the coupled and transmission ports ended with elliptically

shaped microstrip loads. By optimizing the ending ports of the coupler, we achieved a

1.2-6.0 GHz differential phase from - 90o to + 90o, a frequency range that covers most

commercial, satellite and personal mobile communication bands.

Furthermore, we can obtain different phase ranges by terminating the coupler with

suitable reactance loads. Indeed, the simulations showed that the proposed design can

achieve ± 30o, ± 45o, and ± 90o differential phase shifts with deviation less than 3o, as

well as return loss and insertion loss better than 10 dB and 1 dB, respectively. The total

size of the designed phase shifter is 25 x 60 mm2. Measurements results agreed well

with the simulations, thus demonstrating the proposed design approach.

iii

Acknowledgments

I would first like to thank my supervisor Professor Mustapha C.E. Yagoub who has

supported me throughout my thesis with his patience and knowledge.

I attribute the level of my Master’s degree to his encouragement and effort and without

him this thesis, too, would not have been completed or written. One simply could not

wish for a better or friendlier supervisor.

Last but the most I would like to thank my family and my friends for their support and

encouragements during this thesis work.

iv

Abstract ......................................................................................................... ii

List of variables ........................................................................................... ix

List of Acronyms .......................................................................................... x

Chapter One ................................................................................................. 1

Introduction .................................................................................................. 11-1 Motivations .............................................................................................................. 11-2 Thesis contributions ............................................................................................... 31-3 Thesis Organization ............................................................................................... 3

Chapter Two ................................................................................................. 4

Phase Shifters Review .................................................................................. 42-1 Definitions ...................................................................................................................... 42-2 Design Parameters ........................................................................................................ 52-3 Different Types of Phase Shifters ................................................................................ 62-4 Phase Shifters Design Topologies ................................................................................ 9

2-4-1 switched line phase shifters ..................................................................................... 92-4-2 Hybrid-coupled phase shifter ................................................................................ 102-4-3 Loaded-line phase shifters ..................................................................................... 102-4-4 High-pass-low-pass phase shifters ........................................................................ 112-4-5 Discussion .............................................................................................................. 12

2-5 Planar Couplers ........................................................................................................... 132-5-1 Edge coupling configuration ................................................................................. 142-5-2 Broadside slot coupling configuration ................................................................... 14

2-6 Phase Shifter Designs .................................................................................................. 152-6-1 Schifman’s phase shifter ........................................................................................ 162-6-2 Mosko-Shelton’s constant phase shifter and directional coupler .......................... 172-6-3 Abbosh wideband phase shifter ............................................................................. 20

2-7 Conclusions .................................................................................................................. 22

Chapter Three ............................................................................................ 24

Coupler Design ........................................................................................... 243-1 Design Approach ......................................................................................................... 24

3-1-1 Odd and even mode analysis ................................................................................. 26

v

3-1-2 Coupler analytical formulation .............................................................................. 273-2 Designing the hybrid coupler ..................................................................................... 323-3 Manufactured coupler ................................................................................................ 353-3 Conclusion .................................................................................................................... 37

Chapter Four .............................................................................................. 38

Differential Phase Shifter Design ............................................................. 384-1 Designing the differential reflective phase shifter .................................................... 38

4-1-1 Theoretical approach ............................................................................................. 384-1-2 Differential reflective phase shifter’s formulation ................................................ 41

4-2 Design Results with Elliptical-Shaped Microstrip Load ......................................... 424-3 Parametric Analysis .................................................................................................... 464-4 Manufacture and Test Results of the 45o Differential Phase Shifter ...................... 484-5 comparison of the designed phase shifter with others ............................................. 514-6 Conclusion .................................................................................................................... 54

Chapter 5 .................................................................................................... 55

Conclusions and future works .................................................................. 555.1 Conclusions .................................................................................................................. 555.2 Future works ................................................................................................................ 56

References ................................................................................................... 57

vi

Table of Figures

Figure 2 - 1. Different electronically phase shifters [14]. ............................................. 7Figure 2 - 2 PIN diode I-V characteristics [15] ............................................................ 8Figure 2 - 3. PIN diode phase shifters can shift the phase with switching signal between

two paths with two different lengths l0, l0 + l . [15] ............................................ 8Figure 2 - 4. Switched line phase shifters [14]. .......................................................... 10Figure 2 - 5. Hybrid-coupled phase shifters [14] ........................................................ 11Figure 2 - 6. High-pass-low-pass phase shifter [21] ................................................... 12Figure 2 - 7. an edge-coupled configuration [26] ....................................................... 14Figure 2 - 8. Slot coupling configuration [28] ............................................................ 15Figure 2 - 9. Error-correcting network and its two possible differential phase responses

[38]. ..................................................................................................................... 18Figure 2 - 10. Type-B network and its differential phase response. ........................... 19Figure 2 - 11. Configuration of multi-section directional coupler. a) required coupling

response of multi-section coupler. b) decomposition of coupling response ....... 20Figure 2 - 12. Phase shifter a) upper layer b) middle layer c) lower layer d) whole

structure [40] ....................................................................................................... 21Figure 2 - 13 Phase shifter as a four-port device with two open-circuit ports [40] .... 21

Figure 3 - 1. General structure of Tanaka coupler with plane coupling a) transmission

lines, gap and sub layers’ arrangement b) upper view of coupler c) odd and even

modes equivalent circuit [42] .............................................................................. 25Figure 3 - 2. Offered coupler structure’s layers with plane coupling ........................ 26Figure 3 - 3. The coupler model for semi-static analysis [42] ................................... 28Figure 3 - 4. Conversions to find unit capacitor of coupler’s even mode ................. 29Figure 3 - 5. the offered wideband structure layers in 1.2-6 GHz frequency band with

plane coupling ..................................................................................................... 33Figure 3 - 6.Simulation results of the hybrid coupler in 1.2-6 GHz frequency band a)

reflective coefficients b) transmission coefficients c) phase shift d) phase and

amplitude unbalance. .......................................................................................... 35

vii

Figure 3 - 7. the manufactured coupler measurement results. ................................... 35

Figure 4 - 1 Schematic version of offered reflective phase shifter by using the loaded

symmetric directional coupler ............................................................................. 42Figure 4 - 2 a) General structure of the reflective phase shifter with elliptical load, b)

the upper and c) lower transmission lines with elliptical shaped patch and load 43Figure 4 - 3 a) phase shift, b) phase ripple, c) return loss, d) insertion loss of the ±30!

, ±45! , ±60! and ±90! differential phase shifters with elliptical shaped

microstrip load in 1.2-6 GHz frequency band. ................................................... 46Figure 4 - 4 Simulation results of the 45o differential reflective phase shifter for

parametric analysis a) the effect of changing the load’s dimensions on phase shift

b) the effect of changing the length of the reference line on the phase shift c) the

effect of changing the load’s dimensions on the return loss d) the effect of

changing the load’s dimensions on the insertion loss ......................................... 48Figure 4 - 5 a) Measurement system, b) the manufactured 45o phase shifter (top, bottom

and middle views). .............................................................................................. 50Figure 4 - 6 Measurement results of the 45o phase shifter, Return and insertion loss 50Figure 4 - 7 Comparing measurement and simulation results for S11 ......................... 51Figure 4 - 8. Comparing measured and simulated phase shift .................................... 51

viii

Table of Tables

Table 2 - 1 comparison between slot-coupled and edge-coupled configurations [29] 15Table 2 - 2. Comparison between different topologies of phase shifters .................... 23

Table 3 - 1 Dimensions of the designed coupler in 1.2-6 GHz frequency band. ........ 33Table 3 - 2 Comparison of the designed coupler characteristics vs. existing published

works. .................................................................................................................. 36

Table 4- 1 Designed coupler dimensions for reflective phase shifter ......................... 42Table 4- 2 Dimensions of the designed elliptical shaped load to get the ±30! , ±45! ,

±60! and ±90! phase shifts ................................................................................ 44Table 4- 3 Comparison of the designed phase shifter’s specification with other phase

shifters with different structures ......................................................................... 52

ix

List of variables

% Phase constant

βef Effective phase constant

Γ Reflection coefficient

&'(( Effective permittivity

) Electrical length of a uniform line

* Wavelength

+ Phase

, Angular frequency

b Reflective waves

CTe Total unit capacitor of even mode

CTo Total unit capacitor of odd mode

IL Insertion loss

- Length

lref Reference line length

L Length of patch

LS Length of gap

RL Return loss

S Scattering parameters

W Width of patch

Ws Width of gap

./ Characteristic impedance

Z0e Characteristic impedance in even-mode

Z0o Characteristic impedance in odd-mode

x

List of Acronyms

ADS Advanced Design Systems Software

BW Bandwidth

DCS Digital Communication systems

DGS Defected Ground Structure

GPS Global Positioning System

MEMS Micro-electromechanical systems

MMIC Monolithic Microwave Integrated Circuit

PCS Personal Communication Systems

S-DMB Satellite-Digital Multimedia Broadcasting

SIW Substrates Integrated Waveguide

UMTS Universal Mobile Telecommunications Systems

WiMax Worldwide Interoperability for Microwave Access

WLAN Wireless Local Area Network

1

Chapter One

Introduction

1-1 Motivations In modern communication systems, there is a need to control the direction of signal

radiation in order to improve its properties in a particular direction. This is called beam

steering [1].

To this aim, phase shifters introduce a configured amount of time delay (or phase at a

certain frequency) in the signal passing through it; the amplitude of the radiated signal

in each lobe is being controlled by a Variable Gain Amplifier. The radiated waves

interact with each other either destructively or constructively. Therefore, by adjusting

the phases and amplitudes of the transmitted signals, it is possible to reduce the

radiation in all unwanted directions (by destructive interaction) while increasing it in a

particular direction (by constructive interaction) [1]- [3].

Phase shifters are then a critical component in many RF and microwave systems.

Applications include controlling the relative phase of each element in a phase array

antenna in a radar or a steerable communication link, and in cancelation loops used in

high linearity amplifiers.

Because of their importance in communication systems, a quite impressive number of

researches have been already published regarding their design while many others are

still ongoing to enhance the phase shifter characteristics such as increasing bandwidth,

decreasing ripple phase, reducing size, and/or decreasing losses, to name a few.

Depending on how to control the phase shift, phase shifters can be categorized into two

main groups: mechanical and electronical, the latter being much popular, especially in

2

phase array antennas [1]. Because of the ever-increased need for efficient integrated

communication systems with wider bandwidth, the objective of the present work was

to design integrated phase shifters with wide bandwidth and minimum attenuation.

In this optic, features such as flexibility, availability, and ease to use, have made

wireless communication systems unavoidable components in our professional as well

as personnel life.

We have then focused in this work on the design of a multiband 1.2-6.0 GHz phase

shifter, i.e., a frequency range that covers most commercial, satellite and personal

mobile communication bands such as Global Positioning System (GPS, 1.575 GHz),

Digital Cellular System (DCS, 1710 - 1880 GHz), Personal Communications Service

(PCS, 1850- 1990 MHz), Universal Mobile Telecommunications System (UMTS,

1900-2200 MHz), Wireless Broadband (WiBro, 2300 - 2390 MHz), Bluetooth (2.4

GHz), Wireless Local Area Network (WLAN, 5.1–5.9 GHz), and Worldwide

Interoperability for Microwave Access (WiMAX, 2.5 and 5.8 GHz).

Slot-coupled structures are a good candidate for wideband applications because of their

tight coupling, low phase deviation and reduced size [4]. In addition, among integrated

phase shifters, microstrip differential phase shifter are widely used in modern

communication systems [4]-[10]. In fact, such phase shifters have broad applications

in microwave circuits such as butler matrices, monopulse networks, beam-scanning

phased arrays, microwave instrumentation and measurement systems, modulators and

many other industrial applications. They can provide suitable phase difference between

two different paths with minimum effect on each other.

Wideband differential phase shifters are mainly based on the design proposed by

Schiffman [6], an edge-coupled stripline transmission configuration. This structure

consists of a reference line and two edge-coupled striplines integrated together at their

bottom. By choosing the proper length of these lines and the coupling, the phase

difference between them can be made constant over one octave frequency band.

However, a very narrow gap between the edge-coupled lines is needed for a broadband

performance and when this circuit is fabricated in microstrip technology, its operation

decreased. So, Abbosh [4] proposed a microstrip-slot-microstrip coupler with an

elliptical-shaped broadside coupled structure that shows good wideband properties.

3

1-2 Thesis contributions

In this thesis, the main contribution has been to propose a modified configuration of

Abbosh’s phase shifter [4] that can operate over a quite wideband range, covering the

1.2-6.0 GHz band. The proposed enhancements allowed to successfully complete the

design of a coupler with suitable characteristic such as 3 dB coupling with minimum

return loss, phase imbalance and amplitude imbalance and also the design of a phase

shifter with high performance in the targeted frequency range, i.e., 137% bandwidth,

((f2-f1)/fc) maximum of 3o phase deviation, as well as return and insertion loss better

than 10 dB and 1 dB, respectively.

1-3 Thesis Organization

This thesis is divided into four chapters. After this introductory chapter, the following

chapter presents an overview of the existing phase shifters types, designs and

requirements.

In the third chapter, the design process of a wideband directional hybrid coupler is

described including the odd and even modes analysis and related simulations.

Chapter Four provides the design, implementation and results of the multiband

differential reflective phase shifter. Simulations were performed using both a circuit

simulator (Advanced Design System - ADS, from Keysight Technologies) and a full-

wave electromagnetic solver (High-Frequency-Simulation-Software – HFSS, from

ANSYS). The results for phase ripple, phase shift are shown and compared to other

works. The manufactured test results for this phase shifter have been brought in this

chapter as well.

4

Chapter Two

Phase Shifters Review

In this chapter, we will first review the different topologies of phase shifters, depicting

the advantages and disadvantages of each structure. This step will allow us selecting

the most suitable for the targeted application.

2-1 Definitions

Phase shifters are used to change the transmission phase angle (i.e., the phase of the

S21 parameter) of a network. Ideal phase shifters provide low insertion loss and equal

amplitude (loss) in all phase states. Most phase shifters are passive reciprocal networks,

meaning that they work effectively on signals passing in either direction [11].

While the applications of microwave phase shifters are numerous, perhaps the most

important is within a phased array antenna system (as known as Electrically Steerable

Array, or ESA), in which the phase of a large number of radiating elements are

controlled to force the electro-magnetic (EM) wave to add up at a particular angle to

the array. For this very purpose, phase shifters are often embedded in TR modules [11].

The total phase variation of a phase shifter needs only be 360o to control an ESA of

moderate bandwidth. Networks that stretch phase more than 360o are often called time

delay bits or true time delays (part of a time delay unit or TDU), and are built similar

to the switched line phase shifters that will be described below [11].

5

2-2 Design Parameters

In recent decades, one of the most important applications of phase shifters is in phase

array structures. In antenna theory, a phased array is an array of antennas in which the

relative phases of the respective signals feeding the antennas are set in such a way that

the effective radiation pattern of the array is reinforced in a desired direction and

suppressed in undesired directions. The phase relationships among the antennas may

be fixed, as in a tower array, or may be adjustable, as for beam steering.

Common phase arrays can have up to thousands radiation elements. In these structures,

phase shifters are in the form of constant time delay or phase delay devices [12].

Because of this, phase array structures are in the form of phase scanning and time-delay

scan. Phase shifters with small loss, low handling power, continuous adjusting

capability and low production costs are among the most fundamental phase array

antennas part that direct EM waves with electronically controlling the signal’s phase

while antenna is not physically moving.

In designing a phase shifter, beside cost and tolerances (which we did not consider in

this work), different design parameters have to be considered:

• Insertion Loss: The insertion loss of a phase shifter is largely driven by the

number of stages needed and the operating frequency. 4-6dB is a typical range

for a design with 360o of control. The variation with frequency at a given phase

is also critical and performance of +/-1dB over an octave bandwidth is often

required [13].

• Amplitude Imbalance: The magnitude of the RF signal should not be affected by

the phase shifter in ideal conditions. In practice, variations are expected in the

amplitude of the RF signal over frequency, resulting in unequal distribution of

amplitude during the beam steering of the antenna [13].

• Return Loss: Return loss is a measure of the input impedance matching of the

phase shifter. A typical number for a broadband device would be –10dB to –15dB

[13].

6

• Transmission coefficient: In telecommunications, the transmission coefficient is

the ratio of the amplitude of the complex transmitted wave to that of the incident

wave at a given point in a transmission chain.

• Phase deviation: In phase modulation, the phase deviation is stated as the

maximum difference between the instantaneous phase angle of the modulated

wave and the phase angle of the unmodulated carrier.

• Size: Size could be an important issue depending on the application as well as

other requirements. However, covering a smaller place is always desired to make

room for other components in a system.

• Power Handling Capability: Although power-handling capability of the phase

shifter depends on the application, it can be a very important property in

designing the system. Especially in transmitter mode of phased array radars,

phase shifters must be capable of handling much power. However, generally each

phase shifter in the array is required to handle some portion of the total power

since transmitter power is distributed among all the phase shifters [12].

2-3 Different Types of Phase Shifters

In general, phase shifters can be categorized into three groups: Ferrite, Semiconductors

and Bulk Semiconductors. Figure 2.1 shows different common categories of

electronically phase shifters, in which semiconductor-based ones are the most

important [13].

In fact, although Ferrites phase shifters have less input loss, they are complex and

expensive. Also they require manual adjusting and have high consumption power [14].

Bulk semiconductor phase shifters are cheaper and smaller but they have limited

applications because of high input loss in high frequencies. Also they cannot exhibit

continuous phase shift [14].

7

Figure 2 - 1. Different electronically phase shifters [14].

Semiconductor-based phase shifters can be classified as analog or digital. Analog phase

shifters provide a continuously variable phase, usually controlled by a voltage [13],

while digital phase shifters provide a discrete set of phase states that are controlled by

two-state "phase bits." The highest order bit is 180o, the next highest is 90o, then 45o,

etc., as 360o is divided into smaller and smaller binary steps. A three-bit phase shifter

would have a 45o Least Significant Bit (LSB), while a six-bit phase shifter would have

a 5.6o least significant bit. Most phase shifters are of the digital variety, as they are

more immune to noise on their voltage control lines.

Considering a semiconductor controlling element as a switch, phase shifters are usually

based on PIN diodes or FETs. A PIN diode is a P-N junction with a large depletion

region rather than normal diode (Figure 2-2 [15]). Adding this intrinsic region will help

controlling the conductance capability in forward bias and also decreasing capacitor in

8

reverse bias. In fact, in forward bias, resistance in signal path will become negligible

while decreasing capacitor in reverse bias will lead to higher impedance in this track.

Figure 2 - 2 PIN diode I-V characteristics [15]

Therefore, PIN diode phase shifters can shift the phase with switching signal between

two paths with two different lengths -/, -/01 (Figure 2.3 [15]).

Figure 2 - 3. PIN diode phase shifters can shift the phase with switching signal

between two paths with two different lengths -/, -/01 . [15]

Phase shift is proportional to extra path delay %-, where % is the mean propagation

constant and l the difference between lengths.

FET phase shifter is used as a two-terminal switch controlled by the gate voltage and

has many advantages compared to PIN diode [15]. It has higher switching speed and

9

less consumption power. While PIN diode is known as a digital shifter, FET can operate

in both analog and digital modes. Note that since semiconductor phase shifters are

expensive and have high input loss in microwave frequencies, new technologies are

using narrowband linear dielectrics and Micro-Electro-Mechanical Systems (MEMS)

to obtain phase shifts with very low input loss (< 2dB).

2-4 Phase Shifters Design Topologies

2-4-1 switched line phase shifters

The primary kind of phase shifters is the switched-line phase shifter. With this kind of

phase shifter, it is conceivable to switch between two (or more) delay lines as shown

in Figure 2-4. The phase shift given by this circuit is the distinction between electrical

lengths between two transmission lines.

There are certain disadvantages for this type. First, an advanced CMOS technology is

expected to realize this phase shifter at high frequencies with low insertion loss.

Second, the two lines have unequal lengths and different attenuations. This implies that

the two lines that the two-time delay states will have different attenuations, resulting in

amplitude imbalance between the two states. Third, to realize large time delays, long

line lengths are needed which is impractical for integrated circuit realization for most

frequency ranges and increases loss for larger time delays [17].

10

Figure 2 - 4. Switched line phase shifters [14].

2-4-2 Hybrid-coupled phase shifter

A reflective type phase shifter is composed of a 90° Hybrid coupler with two identical

reflective loads as shown in Figure 2.5. The Hybrid coupler divides the input signal at

port 1, equally between the two output ports, port 3 and 4, with a phase difference of

90°. Signals reflected back from the termination add up at port 2 and no signals returns

to port 1 [12]. Since isolation between the input and output ports is improved, there can

be better matching at each port. However, the circuit will occupy more chip area and

loss is worse because of the coupler [19].

2-4-3 Loaded-line phase shifters

Another category of phase shifters is the loaded-line phase shifter, often used for 45o

or lower phase shift bits. The loads ZL are synthesized such that they create a

perturbation in the phase of the signal when switched into the circuit, while they have

only a small effect on the amplitude of the signal.

The loads must have a very high reflection coefficient in order to minimize the loss of

the phase shifter (they should utilize purely reactive elements). Obviously, the loads

must not be too close to a short circuit in phase angle, or the phase shifter will suffer

11

from extreme loss. By spacing the reactive loads, approximately a quarter-wavelength

apart, the amplitude perturbation can be minimized and equalized in both states [20].

Figure 2 - 5. Hybrid-coupled phase shifters [14]

The phase versus frequency response of a loaded line phase shifter is usually flatter

than the switched line phase shifter, but not usually as flat as the high-pass/low-pass

phase shifter. Usually only one control signal is required for a loaded-line phase shifter,

since the loads can be biased simultaneously.

One big issue with this topology is that it is impossible to have a matched circuit in

both states. The matching also deteriorates when a larger phase shift is needed. The

circuit will also have different attenuations depending on whether the switches are open

or closed [19].

2-4-4 High-pass-low-pass phase shifters

if a constant phase shift is desired over a wide frequency range, the switched line phase

shifter isn’t going to cut it. A high-pass/Low-pass phase shifter can provide near

constant phase shift over an octave or more. By high-pass/low-pass we refer to the fact

that one arm forms a high-pass filter while the opposite arm forms the low-pass filter.

The second advantage of the high-pass/low-pass phase shifter is that it offers a very

compact layout because lumped elements are typically used instead of delay lines. This

12

is an important consideration for low frequency designs because delay transmission

lines can be quite large. The cut-off frequencies of the two filter networks obviously

must be outside of the phase shift band for this scheme to work [21].

Figure 2 - 6. High-pass-low-pass phase shifter [21]

Various high-pass-low-pass phase shifters have been reported in different technologies,

typically designed as integrated circuits [22]-[23]. They are generally limited to no

more than 40% fractional bandwidth. Sometimes, switching transistors are also

integrated into the phase shifter, thus reducing size and the number of required

components [19].

2-4-5 Discussion From the above, all these types have certain advantages and disadvantages. However,

reflection phase shifters are still the most suitable in our case because of their high

bandwidth.

To achieve suitable operation in a wide bandwidth, phase shifters in planar technology

(e.g., microstrip lines) use coupled transmission lines working as couplers [24].

13

2-5 Planar Couplers

As mentioned, since we chose the reflection type, we need directional hybrid couplers.

A simple directional coupler is just two transmission lines put close together, such that

the secondary line can absorb energy from the field created around the primary. The

resulting structure can have many properties, some of which are desirable and others

are not. Coupled line couplers are not "DC connected", as opposed to "direct coupled"

couplers such as the Wilkinson and the branch-line. Coupled lines occur when two

transmission lines are close enough in proximity so that energy from one line passes to

the other. The Through Line, or Main Line, needs to be designed for the best possible

match and must be capable of handling whatever the maximum input power level is

specified to be. The quality of match for the Coupled Line will affect the directivity of

the coupler. Directivity is a measure of “how well the coupler isolates two opposite-

travelling (forward and reverse) signals” [25].

However, some types of couplers will show a greater sensitivity to this than others. For

example, if by using both ends of a single coupled line to provide Forward and Reverse

coupled output, then the match of this line and also the match of the external

components is absolutely critical to directivity. However, if the coupled line will have

a termination at one end, then this may be tuned to optimize directivity. It is perfectly

feasible to create a dual directional coupler using two separate coupled lines each of

which is terminated at one end.

Usually we are talking about lines that are coupled over a quarter-wave section, or

multiple sections. Bandwidth is greater than in interconnected transmission line

(uncoupled line) circuits like the branch-line coupler. Lines can be coupled in at least

two ways.

Hybrid couplers are the special case of a four-port directional coupler that is designed

for a 3-dB (equal) power split. Hybrids come in two types, 90 degree or quadrature

hybrids, and 180 degree hybrids. Hybrid couplers are often used in creating reflection

phase shifters.

14

2-5-1 Edge coupling configuration

This structure includes a reference line and two transmission lines connected at the end

(Figure 2.7 [26]). By properly choosing the line’s lengths and edge-couplings between

the two transmission lines, we can find adequate phase shift with one octave wideband

[26]. The gap between two lines is one of the important parameters to achieve wideband

shift in this edge-coupling structure and for narrower gap we have longer phase

shifter’s bandwidth.

However, producing this structure with narrow gap in microstrip technology would be

difficult, thus, application of this type is limited.

Figure 2 - 7. an edge-coupled configuration [26]

2-5-2 Broadside slot coupling configuration

The slot-coupling approach involving a double-sided substrate, which was first

proposed by Tanaka et al. [27]. It can be applied to realizing a tight coupling. The

structure is formed by two microstrip lines separated by a rectangular slot in the

common ground plane (Figure 2-8 [28]).

Table 2.1 summarizes the advantages/disadvantages of each of the above structures

highlighting that slot-coupled structures have better performance in phase deviation,

bandwidth and size reduction [28]. We will then use this type of devices.

15

Figure 2 - 8. Slot coupling configuration [28]

Table 2 - 1 comparison between slot-coupled and edge-coupled configurations [29]

Edge-coupled Slot-coupled

Operating bandwidth High Very high

Characteristics

Return loss High High

Insertion loss Low Relatively low

Phase ripple Normal Very low

Design issues very narrow gap between transmission lines

Complexity due to double layers circuit

2-6 Phase Shifter Designs

The first published wideband phase shifter based on a Slot-coupled structure was the

Schiffman phase shifter [26]. After that, several structures have been proposed [30]-

[35]. For instance, in [31], different structures based on Schiffman structure have been

16

discussed. In addition to common Schiffman’s structures, others configurations such

as SIW [36] and loaded transmission line phase shifters [37] have be proposed.

Slot coupled transmission signal’s structures consist generally on two sub layers. For

analyzing these structures, odd and even mode analysis are used. This part will be

covered in the next chapter.

2-6-1 Schifman’s phase shifter

As mentioned earlier, the Schiffman’s phase shifter has been the first published

wideband phase shifter design with 90o’ difference phase [38]. This differential phase

shifter consisted of two transmission lines, a reference transmission line and an edge

coupling due to two beside lines. By properly choosing the lines length and coupling

rate, the phase difference in the whole bandwidth can be maintained relatively constant,

i.e., 90o. Notice that Schiffman’s work was on strip line structure, which scattered odd

and even modes in the coupled line have identical phase speed. Therefore, when this

type is designed with microstrip lines, unequal speed of odd and even modes will lead

to weak phase shifter operation.

In [38], two identical parallel-coupled lines were connected at their end and the

frequency behavior of the rippled lines network has been evaluated and the image

impedance equations and constant phase defined as

(2-1)

and

(2-2)

Zr ϕ

ZI = Z0oZ0e

cos(ϕ ) =

Z0eZ0o

− tan2θ

Z0eZ0o

+ tan2θ

17

where Z0e is the characteristic impedance in even-mode (when equal in-phase currents

flow in both lines) and Z0o represents the characteristic impedance in odd-mode (when

equal out-of-phase currents flow in both lines). Also ) = %3 is the electrical length of

a uniform line of length - and phase constant %.

The most elementary form of such a network is termed Type-A network. The

Schiffman’s method for reducing the maximum phase error of type-A was to connect

another differential phase shifter network to the system, which has symmetric to

neutralize each other’s. This structure, called error-correcting network, consists of two

separate structures (Figure 2-9 [38]), a transmission line with the length of 2ml and a

coupled-line’s part with the length of 2ml, with 4 = ./'/./6.

According to the parameters 4 and , we can minimize the bandwidth’s error [38].

Therefore a type-B structure was designed to reduce the error by taking m = 3,

and , leading to a maximum 0.7o phase error in about one octave bandwidth

(Figure 2-10 [38]). In practice, the structure can be simplified by removing equal

lengths of uniform transmission line from each branch, without affecting differential-

phase response. Such a reduction is illustrated in figure for the Type-B network.

2-6-2 Mosko-Shelton’s constant phase shifter and directional coupler

To gain more bandwidth with an acceptable phase ripple by using the edge-coupling

method, Mosko and Shelton have proposed an approximate method for making

constant phase shifting, which includes different parts of a quarter of parallel coupled

wavelength [39]. The main problem with this method, which is generally for edge-

coupling phase shifters, is the need of high coupling for gaining large bandwidth.

Therefore, it may be impractical. Mosko and Shelton proposed to use series connection

to minimize this effect. Another problem of these structures is the large size of these

multi-part structures.

Δϕ

m

ρ1 = 4.1

ρ2 = 2.4

18

Figure 2 - 9. Error-correcting network and its two possible differential phase responses

[38].

19

Figure 2 - 10. Type-B network and its differential phase response.

20

Figure 2 - 11. Configuration of multi-section directional coupler. a) required coupling

response of multi-section coupler. b) decomposition of coupling response

2-6-3 Abbosh wideband phase shifter

In [40], Abbosh proposed a new approach in designing a phase shifter that exhibits

good properties in UWB (which is appropriate to our work). This method was used to

design 30 and 45o UWB phase shifters with size of 2.5cm x 2cm (Figure 2-12, [40]).

This structure is formed by two ellipsoid microstrip patch connected to input and output

microstrip lines. Upper and lower layers have been placed in front of each other. The

coupling between these patches takes place in the ellipsoid gap (ground level, middle

layer). Note that the ellipsoid form can provide a constant coupling coefficient in UWB.

Results showed a differential deviation and ripple less than 1 dB as well as a return

loss better than 10 dB. ±3!

21

Figure 2 - 12. Phase shifter a) upper layer b) middle layer c) lower layer d) whole

structure [40]

Figure 2 - 13 Phase shifter as a four-port device with two open-circuit ports [40]

The analysis starts by assuming the phase shifter as a four-port device. Like figure 2-

13 two ports are considered open circuit (with this assumption the reflection coefficient

at ports 3 and 4 will be 1). Considering that the designed phase shifter has coupling C

22

between upper and lower patches, the scattering parameters of the above structure

would be:

(2-3)

(2-4)

where βef represents the effective phase constant in the medium of the coupled

structure.

As mentioned before, edge coupling in single layers’ structures and the slot-coupled

method are two important procedures for band widening [39]. From that, enhanced

structures have been proposed [28]-[33]. For instance, in [40], an improved

Schiffman’s structure with modifying ground plane underneath coupled lines has been

proposed. This 1.5-3.1 GHz structure has phase ripple and 0.5 dB amplitude ripple.

In [31], Sorn has improved the Abbosh’s structure by adding a gap in slot-coupled

output and gained 90o phase difference. This structure works in 3-7 GHz bandwidth. A

0.75-2.4 GHz and 90o phase shifter with stepped impedance open stub and coupled line

with 1.1 dB amplitude ripple and phase deviation has been proposed in [34]. Also

in [35], a 90o phase shifter, with loaded transmission line and T form open stub, and

operating in the 2.3-5.5 GHz frequency range, has been presented. This structure has

phase deviation and less than 0.6 dB loss. In [36], 45o, 90o, and 135o phase shifters

with referenced line and coupled lines have been described. Working in the range of

2.24-3.55 GHz, they show phase ripple and loss less than 0.9 dB.

2-7 Conclusions

In this chapter, we analyzed different wideband differential phase shifters. We

reviewed the advantages and disadvantages of conventional methods for phase shifter’s

S11 =1−C 2 (1+ sin2(βef l))

1−C 2 cos(βef l)+ j sin(βef l)⎡⎣

⎤⎦2

S21 =j2C 1−C 2 sin(βef l)

1−C 2 cos(βef l)+ j sin(βef l)⎡⎣

⎤⎦2

5!

5!

6.4!

5!

23

design (Table 2-2). By comparing different structures, the operating frequency band of

slot-coupling structures is higher compared to other structures. The other benefit of this

structure is its smaller structure size and lower ripple. We, therefore, opted for a

wideband reflective phase shifter based on double layer slot-coupled coupler

configuration. From that, we first designed the required coupler as detailed in the next

chapter.

Table 2 - 2. Comparison between different topologies of phase shifters

type advantages disadvantages

Switched-line Perfect for true time delay Variable insertion loss

Reflection-type Better matching at each

port

Occupy more chip area

and more loss because of

coupler

Loaded-line The phase vs frequency is

more flatter

Impossible to have a

matched circuit in both

states

High-pass-low-pass Provide near constant

phase shift over an octave

or more

Limited to no more than

40% fractional bandwidth

24

Chapter Three

Coupler Design

In the previous chapter, we discussed about the different kinds of phase shifters and

retained the reflective phase shifter with slot-coupling configuration. To achieve our

design, we will start with the Abbosh’s structure, enhancing it to uniform the

coupling between transmissions lines on upper and lower sub layers at the minimum

and maximum frequencies of the UWB band (3.1-10.6 GHz). Then, we will vary the

sub layer dimensions to decrease the return loss and insertion loss. To do so, we have

to start by designing a wideband hybrid coupler with two loaded ports.

3-1 Design Approach

To design a coupler with tight coupling in a so wide frequency range, we retained the

plane coupling structure introduced by Tanaka (Figure 3-1, [41]). Its smaller size and

wideband operating frequency are the main reasons we selected it.

Its smaller size (e.g. compared to Schiffman’s structure) is due to the use of a plane

coupling between the two transmission lines that are placed on upper and lower sub

layers (via the gap on the ground plane). This plane coupling is substituted to the edge

coupling in the Schiffman’s structure. Also, its phase ripple in the operating band is

relatively less compared to other microstrip structures [39].

Figure 3-1-a shows the general form of this structure. Figure 3-1-b depicts the

rectangular Tanaka coupler structure. Figure 3-1-c shows the electric field lines for odd

and even modes excitation.

45!

25

Figure 3 - 1. General structure of Tanaka coupler with plane coupling a) transmission

lines, gap and sub layers’ arrangement b) upper view of coupler c) odd and even

modes equivalent circuit [42]

However, one of the issues in designing the Tanaka coupler is its ports’ location [39].

The solution is usually to change the location of adjacent ports by angulating the

connected transmission lines, as seen in figure 3-2. The operating frequency band can

be also adjusted by changing the geometric form of the gap on the ground plane (e.g.

from rectangular to ellipsoid) and then optimizing the physical dimensions. This

approach can be also considered in the Abbosh’s coupler.

26

Figure 3 - 2. Offered coupler structure’s layers with plane coupling

3-1-1 Odd and even mode analysis

We designed the hybrid coupler on Rogers RO4003C substrate with 20 mil thickness

with dielectric constant εr=3.38 mainly because of its availability.

The first step was to compute the odd and even impedances for a 3 dB coupling and

50Ω characteristic impedance Z0 using the following relations [42]

kc(dB) = 20 logZ0e − Z0oZ0e + Z0o

⎛⎝⎜

⎞⎠⎟

(3-1)

Z0 = Z0eZ0o (3-2)

27

leading to Z0e = 120.5Ω and Z0o = 20.7Ω . Then, it is necessary to analyze the coupler

operation in dual modes.

When the odd mode is excited, the gap on ground plane can be replaced by a perfect

electric conductor (PEC). The upper part of the resulting equivalent coupler converted

to a microstrip line with Z0o characteristic impedance. The related line width, wp , can

be then obtained using the standard designing equations for microstrip lines.

As for the even mode signal’s excitation, the perfect magnetic conductor (PMC) is

substituted to the gap on the ground. The magnetic plane will push the electric field

from the microstrip ports to the outside area of the parallel planes because the magnetic

conductor in the lower plane will not let the electric field to be perpendicular on its

surface. Therefore, the even mode signal moves outside the area of parallel planes as

shown in figure 3-1-c. To allow the even mode signal moving smoothly from the port

to the lines, it is recommended to modify the shape of the upper and lower transmission

lines as well as the ground gap from rectangular to ellipsoid (figure 3-2) [42].

3-1-2 Coupler analytical formulation

As mentioned before, to improve the coupler characteristics, the rectangular form of

Tanaka’s coupler with dimensions wp and ws was changed to ellipsoid with diameters

W and Ws (figure 3-2). Since Eqs. (3-1) and (3-2), for finding the dual mode

characteristic impedances, are independent of the geometric shape, they are still valid.

Then, we found the closed form of analytical equations for dual mode semi static

parameters. The results gained by this method are totally similar to Tanaka’s

expressions, which used the complex numerical Finite element method for semi-static

cases. Then, full wave analysis by using spectral domain approach was performed to

gain the dispersion characteristics of dual mode coupler’s parameters.

Note that in this semi static analysis, we assumed the isolation planes, which surround

the coupler from up and down, as infinite. Also we assumed that the transmission line

thickness and can be neglected. As shown in figure 3-3, the planes that separating the

substrate and the air (i.e., AD and A’D’) operate as perfect magnetic conductors.

28

Figure 3 - 3. The coupler model for semi-static analysis [42]

So the existence of the gap in the mutual ground plane does not affect the coupler’s

odd mode characteristics.

Next, we calculated the coupler’s dual mode impedances, Z0o and Z0o . Figure 3-4

shows the conversions for the even mode.

This was achieved by determining the dual mode capacitance value for each unit. Let

the total unit capacitor of even and odd modes, respectively CTe and CTo , be:

• C1 for the bounded area between the upper shield and the upper microstrip half

plane (filled with air), or

• C2 for the bounded area between the upper microstrip planes and the common

ground (filled with dielectric).

C1 and C2 are computed for both modes. Note that for the even capacitance, we have

CTe = C1e +C2e (3-3-a)

C1e = 2ε0K k2( )K ' (k2 )

and C2e = 2ε0ε rK ' k1( )K(k1)

(3-3-b)

29

where K(k) is the type one elliptical integral K '(k) = K(k ') as k ' = 1− k2 and

K(k)K '(k)

=

2πln 2 1+ k

1− k⎛⎝⎜

⎞⎠⎟

0.707 ≤ k ≤1

π

2 ln 1+ 1− k2

1− 1− k2⎛

⎝⎜⎜

⎞

⎠⎟⎟

0 ≤ k ≤ 0.707

⎧

⎨

⎪⎪⎪⎪

⎩

⎪⎪⎪⎪

(3-4)

Figure 3 - 4. Conversions to find unit capacitor of coupler’s even mode

30

k1 and k2 are determined from

k1 =sinh2 πws

4h⎛⎝⎜

⎞⎠⎟

sinh2 πws

4h⎛⎝⎜

⎞⎠⎟ + cosh

2 πws

4h⎛⎝⎜

⎞⎠⎟

and k2 = tanhπwp

4h⎛⎝⎜

⎞⎠⎟

(3-5)

Then, going back to the original design of Tanaka (figure 3-1), it is possible to deduce

the length lp and width wp of the middle transmission line, as well as the length ls

and width ws of the gap.

Finally, the effective dielectric constant εeffe( ) and the characteristic impedance (Zoe )

for the even mode can be obtained by

εeffe =CTe

CTe(ε r →1)= ε r

K '(k1)K(k1)

+ K(k2 )K '(k2 )

⎡

⎣⎢

⎤

⎦⎥ /

K '(k1)K k1( ) +

K(k2 )K '(k2 )

⎡

⎣⎢

⎤

⎦⎥ (3-6)

Zoe =60πεeffe

K '(k1)K(k1)

+ K(k2 )K '(k2 )

⎡

⎣⎢

⎤

⎦⎥

−1

(3-7)

Note that, similarly, the BB’ plane in figure 3-3 works as a perfect magnetic conductor,

leading to the total capacitor in the odd mode as:

CTo = C1o +C2o (3-8)

C1o = 2ε0K k4( )K ' k4( ) and C2o = 2ε0ε r

K ' k3( )K k3( ) (3-9)

with

k3 = tanhπwp

4h⎛⎝⎜

⎞⎠⎟

and k4 = tanhπwp

4ho

⎛⎝⎜

⎞⎠⎟

(3-10)

31

The effective dielectric constant εeffe( ) and the characteristic impedance (Zoe ) of the

odd mode can be then deduced as

εeffe =CTo

CTo(ε r →1)= ε r

K '(k3)K(k3)

+ K(k4 )K '(k4 )

⎡

⎣⎢

⎤

⎦⎥ /

K '(k3)K k3( ) +

K(k4 )K '(k4 )

⎡

⎣⎢

⎤

⎦⎥ (3-11)

Zoo =60πεeffo

K '(k3)K(k3)

+ K(k4 )K '(k4 )

⎡

⎣⎢

⎤

⎦⎥

−1

(3-12)

The next step was to determine the coupler’s characteristics using MATLAB. The Zoo

and Zoe are linked with the values of ws ,wp , and the substrate thickness. So, with a

coupling kc( ) of 3 dB and a line characteristic impedance Z0( ) of 50Ω , Zoo and Zoe

can be achieved by equation (3-1).

Now we have two unknown variablesws and wp and two equations. The system was

solved after 50 iterations using the newton method in MATLAB. The final results give

78 = 6.5<< and 7= = 4.2<<. Once the transverse dimensions obtained, we had to

compute the longitudinal dimensions.

Now we use an approach, which consists to start from the design method described by

Schiffman, i.e., to use the concept of two coupled quarter-wavelength transmission

lines. Note that the geometric shapes of the transmission lines and the gap on the ground

plane were not taken into account during simulations. In fact, the above equations were

general and independent of the shapes. We therefore included such information by

starting the simulations with the initial design of Tanaka shown in figure 3-1.

Then, by considering the operating frequency range, 1.2-6.0 GHz, the central frequency

will be 3.6 GHz. For the design, we selected the cost-effective Rogers’ substrate

RO4003C with a relative permittivity &A = 3.38 and 20 mil thickness (equals to 0.508

mm). The quarter-wavelength length of the transmission line will be:

-= = -8 =D

E FG= H

E( FG= 11.33<< (3-13)

32

However, the above results are for a rectangular coupler shape. Thus, after determining

the equations for the rectangular coupler, we can use the equivalence principle to find

the equations for an elliptical coupler, assuming that the length of the rectangular

coupler is equal to the elliptical coupler.

Let L and LS be the respective lengths of the patch and the gap, and W and Ws the

corresponding widths (figure 2-2). With the above assumption, converting the gained

dimensions from rectangular to ellipsoid can be achieved as: (geometric equivalence)

L = lp + lp2 +wp

2( ) / 2 LS = ls + ls

2 +ws2( ) / 2 (3-14)

J ≈ 1.273(7=×-=)/P J8 ≈ 1.273(78×-8)/P8 (3-15)

3-2 Designing the hybrid coupler

The above equations can be used to design a hybrid coupler for different frequencies.

As for the operating bandwidth, it is determined based on the application specifications.

In this thesis, the purpose is to design a hybrid coupler with suitable characteristics in

the 1.2-6 GHz operating bandwidth with minimum return loss, phase and magnitude

ripple.

Designed at a central frequency of 3.6 GHz, the optimized dimensions of the designed

coupler (Figure 3-5) are summarized in Table 3-1.

The simulated coupler’s characteristics, shown in figure 3-6, are a return loss greater

than 23 dB in the whole 1.2-6 GHz frequency bandwidth with 137% frequency

bandwidth (BW = Δf / f 0) relatively to the central frequency, as well as a magnitude

and phase unbalanced less than 0.75! and 0.4 dB, respectively. This design has been

also successfully compared to existing designs as summarized in Table 3-2.

33

Figure 3 - 5. the offered wideband structure layers in 1.2-6 GHz frequency band with

plane coupling

Table 3 - 1 Dimensions of the designed coupler in 1.2-6 GHz frequency band.

Parameters (mm)

Wm50 Ws1 W1 Ws W Ls1 L1 LS L

1.1 1.05 1.4 11 7 4.5 5.2 10.8 11

34

(a)

(b)

(c)

35

(d)

Figure 3 - 6.Simulation results of the hybrid coupler in 1.2-6 GHz frequency band

a) reflective coefficients b) transmission coefficients c) phase shift d) phase and

amplitude unbalance.

3-3 Manufactured coupler

in this section we compared the results of manufactured coupler with the simulated

ones (Figure 3-7). In the simulation the wave ports are used and the connectors are not

included. Also we didn’t simulate the holes and screws on the board. We used the

Vector Network Analyzer with matched load as our test equipment.

Figure 3 - 7. The manufactured coupler measurement results.

36

Table 3 - 2 Comparison of the designed coupler characteristics vs. existing published

works.

Couplers’ characteristics

Structure

Ref. Fractional

Bandwidth

(%)

Frequency

band

(GHz)

Transmission

coefficient

(dB)

Phase

ripple

( ! )

137 1.2-6 3± 0.4dB ±0.75!

Elliptical shaped slot-

coupled microstrip coupler

our

work

42 3.6-5.5

3± 0.5dB ±1! N-section microstrip

tandem structure coupler [43]

180 1-9

3± 0.65dB

±5!

Vertically installed planar

multi section quadrature

hybrid coupler

[44]

111 0.8-2.8

1dB

±1.5!

Rat-race couplers with

coupled-line section and

impedance transformers

[45]

109 3.1-10.6

3±1dB

6 ±1.4dB

10 ±1.5dB

±1!

Elliptical shaped slot-

coupled microstrip coupler [46]

120 2-8

3±1.5dB

±7!

CPW slot-coupled

directional coupler [47]

109 3.1-10.6

3± 0.75dB

±2!

Rectangular shaped slot-

coupled microstrip coupler [48]

161 1.2-12

10 ± 0.6dB

±0.7!

Multi section corrugated

slot-coupled directional

coupler

[49]

37

Note that, to the best of the authors’ knowledge, there is no microstrip phase shifter in

this frequency range with this large bandwidth (137%). Furthermore, the designed

phase shifter’s characteristics are better than those mentioned articles.

3-3 Conclusion

In this chapter, we discussed about the hybrid coupler design approach and analytical

formulation. The simulations are performed in HFSS and ADS. The simulated coupler

exhibits a return loss greater than 23 dB in the whole desired 1.2-6 GHz frequency

bandwidth (137%) as well as a magnitude and phase unbalanced less than 0.75! and 1

dB, respectively for the measured results.

38

Chapter Four

Differential Phase Shifter Design

As already mentioned, the purpose of this thesis is to design a wideband differential

phase shifter operating in the 1.2-6 GHz frequency band. For this purpose, we used the

concept of reflective phase shifters.

For building a reflective phase shifter, we used the wideband hybrid coupler we

designed in previous chapter, while focusing on the load. In a first step, we designed

the required reactance load as a lumped inductor or capacitor.

After obtaining its value, the load was connected between the two output ports of the

designed coupler while the two other ports were used as input and output of the phase

shifter. Thus, the desired differential phase can be obtained by comparison with a

reference line.

In a second step, we modeled the lumped load as an equivalent microstrip line in order

to obtain a planar structure.

4-1 Designing the differential reflective phase shifter

4-1-1 Theoretical approach

As stated, the purpose was to obtain a reflective differential phase shifter. To start, the

analytic expression of the load should be found. Let port 1 and port 2 of the designed

4-ports dual mode coupler be the respective input and output port of the phase shifter

39

(Figure 4-1). In order to create a certain coupling kc between the upper and lower

patches, the corresponding reflective waves b1 and b2 can be computed as:

b1 =jkc sinθa3 + 1− kc

2a41− kc

2 cosθ + j sinθ

b2 =jkc sinθa4 + 1− kc

2a31− kc

2 cosθ + j sinθ (4-1)

with θ the electrical length of the coupled structure, which can be related to the

physical length l and the effective physical constant βef by:

θ = βef l (4-2)

leading to

βef =

βe + βo

2= 360!

ε rλ

(4-3)

In the above equation, βe and βo are the dual mode constants, λ is the free space

wavelength and ε r the substrate dielectric constant.

By assuming that port 2 (output port) is matched, the reflective signal in ports 3 and 4

will be as follows:

b3 =jkc sinθa1

1− kc2 cosθ + j sinθ

b4 =1− kc

2a11− kc

2 cosθ + j sinθ (4-4)

40

If the ports 3 and 4 are open, their reflective coefficients are equal to one, so a3 = b3

and a4 = b4 . Thus, the return loss and insertion loss will be:

S11 = S22 =1− kc

2 (1+ sin2θ )

1− kc2 cosθ + j sinθ⎡

⎣⎤⎦2

S12 = S21 =j2kc 1− kc

2 (1+ sin2θ )


⎣⎤⎦2 (4-5)

Therefore, the phase shift between the output signal and input signal can be expressed

as:

∠S12 = 90! − 2 tan−1 tanθ

1− kc2

⎛

⎝⎜

⎞

⎠⎟ (4-6)

To find the differential phase shift, we must compare the structure response to that of

a reference line. Thus, a 50Ω microstrip line with physical length lref , phase constant

βref and effective dielectric constant εef has been chosen as reference line with its

phase expressed as:

∠S34 = −βref lref = −360!lref εef / λ (4-7)

The equations relevant to εef , brought from [27], lead to

ΔΦ = ∠S12 −∠S34 = 90! − 2 tan−1 tanβeff l

1− kc2

⎛

⎝⎜

⎞

⎠⎟ + βeff lref (4-8)

41

4-1-2 Differential reflective phase shifter’s formulation

Following the above equations, we loaded the coupler’s ports to determine the phase

shift coming from it. Let Γ be the reflection coefficient. The new scattering matrix

parameters S’ (S matrix with loads), will be as follows:

S11' = S22

' = Γ 1− kc2 (1+ sin2θ )


⎣⎤⎦2

S12' = S21' = Γ

j2kc 1− kc2 (1+ sin2θ )


⎣⎤⎦2 (4-10)

The phase of S12' can be computed by summing the load phase and the phase of S12 .

∠ ′S12 = ∠S12 +∠Γ = 90! − 2 tan−1 tanθ1− kc

2

⎛

⎝⎜

⎞

⎠⎟ +∠Γ (4-11)

Then we needed to compare the phase with a reference line with suitable length lref as

its phase is given (equation (3-32)) as

∠ ′S12 = ∠S12 +∠ ′S34 = −βref lref (4-12)

and the final phase shift of reflective phase shifter will be as follows:

∠ ′Φ = ∠ ′S12 −∠ ′S34 = 90! − 2 tan−1 tanβeff l

1− kc2

⎛

⎝⎜

⎞

⎠⎟ +∠Γ + βref lref (4-13)

42

Figure 4 - 1 Schematic version of offered reflective phase shifter by using the loaded

symmetric directional coupler

As can be seen, we can design different phase shifters only by changing the load.

Although loading has most effects on changing the phase, the load also has an impact

on the magnitude or phase of other scattering parameters (according to (4-10)). In this

case, we had to change the dimensions of the coupler, as it will be discussed later.

4-2 Design Results with Elliptical-Shaped Microstrip Load

Following the coupler’s design process and formulation described in parts 3-3 and 3-

2-2, respectively, its geometrical dimensions were obtained (Figure 4-2 and Table 4-

1). As mentioned, the circuit has been designed on a Rogers RO4003C two-layer board

with 20 mil thickness, dielectric permittivity ε r = 3.38 and central frequency 3.6 GHz.

Table 4- 1 Designed coupler dimensions for reflective phase shifter

Parameters (mm) Coupler’s

characteristics Wm50 W1 W1 Ws W Ls1 L1 LS L

1.1 1.05 1.4 11 7 4.5 5.2 10.8 11 Values

43

Figure 4 - 2 a) General structure of the reflective phase shifter with elliptical load, b)

the upper and c) lower transmission lines with elliptical shaped patch and load

Also, by computing the load’s reflective coefficient value we can achieve the required

phase shift. So, we first determined the Γ value for the desired phase shift in order to

get the initial dimensions of the elliptical load using ADS. Then, we optimized its

dimensions in HFSS (Table 4-2). As mentioned, the structure should be capable of

creating the −90! to +90! phase shift on the 1.2-6 GHz frequency band as well as

different other phase shifts by changing the dimensions of the elliptical shaped load in

coupler’s ports. Therefore, the designed load dimensions for ±30! , ±45! , ±60! and

±90! phase shifters have been also brought in Table 4-2.

As can be seen in figure 4-3, the reflection and crossing coefficients of the designed

structures are less than 10 dB and 1 dB, respectively. The −90! , −60! , −45! , −30! , 30!

, 45! , 60! and 90! phase shifters exhibit a respective 1.6! , 1.2! , 1! , 1.2! , 1.1! ,

2.1! , 1.25! and 2! phase deviation in the operating bandwidth. Therefore, the proposed

phase shifters have a phase ripple less than 2.1! in a 137% frequency bandwidth

(BW = Δf / f 0) relatively to the central frequency.

44

Table 4- 2 Dimensions of the designed elliptical shaped load to get the ±30! , ±45! ,

±60! and ±90! phase shifts

a (mm) b (mm) Considered phase shift

18 18.5 −90! 7 14 −60! 3 17 −45! 3 12.2 −30! 1 11 +30! 3 3 +45! 5 1 +60! 1 2 +90!

45

(a)

(b)

(c)

46

(d)

Figure 4 - 3 a) phase shift, b) phase ripple, c) return loss, d) insertion loss of the

±30! , ±45! , ±60! and ±90! differential phase shifters with elliptical shaped

microstrip load in 1.2-6 GHz frequency band.

4-3 Parametric Analysis

In this part, our purpose is to analyze the effects on coupler’s characteristics caused by

changing the phase shifter parameters. Let us take the 45o phase shifter as illustration.

Figures 4-4 show the effect of changing a) the load’s dimensions on the phase shift, b)

the effect of changing the length of the reference line on the phase shift, c) the effect

of changing the load’s dimensions on the return loss, and d) the effect of changing the

load’s dimensions on the insertion loss. For all the simulations, the wave ports are used

and connectors not included.

47

(a)

(b)

(c)

48

(d)

Figure 4 - 4 Simulation results of the 45o differential reflective phase shifter for

parametric analysis a) the effect of changing the load’s dimensions on phase shift b)

the effect of changing the length of the reference line on the phase shift c) the effect

of changing the load’s dimensions on the return loss d) the effect of changing the

load’s dimensions on the insertion loss

From these figures, we can note that changing the length and the width of the elliptical

load, i.e., the load’s reactance, have a significantly impact on the phase shift but much

less effect on the return and insertion loss. So, we can control the phase shift by

changing the load in the ports. On the other hand, the length of the transmission line

has an important role in adjusting the required differential phase shift and choosing the

suitable length for the transmission line that would give the desired phase shift for the

considered load.

4-4 Manufacture and Test Results of the 45o Differential Phase Shifter

To demonstrate our design, we manufactured the 45o phase shifter using the parameter

values got from simulations (figure 4-5). The dimensions of the designed circuit with

the length of the reference line is 24<<×63<< which is equal 0.14*'((×0.3*'((

based on effective wavelength. The test results of the fabricated phase shifter are shown

49

in figure 4-6. From this figure, we can conclude that there is a little difference between

measured and simulated results. The return loss and insertion loss are better than 10 dB

and 1.5 dB, respectively, and the maximum phase ripple is 3°. For the measurement,

we used the Keysight (Agilent/HP) 8720ES Vector Network Analyzer as in figure 4-5.

The simulation is carried out considering the holes in where screws are located. We

had done simulation by considering the screws as some rod metals. And we have used

the wave ports and connectors are not included. It also might be a mismatch between

load and connectors. The simulations are performed in HFSS and ADS. The screws

can cause resonances. It also might be a mismatch between load and connectors.

a)

50

b)

Figure 4 - 5 a) Measurement system, b) the manufactured 45o phase shifter (top,

bottom and middle views).

Figure 4 - 6 Measurement results of the 45o phase shifter, Return and insertion loss

51

Figure 4 - 7 Comparing measurement and simulation results for S11

Figure 4 - 8. Comparing measured and simulated phase shift

4-5 comparison of the designed phase shifter with others

Because of its better response in terms of bandwidth, the elliptical load was used and

its dimensions varied to improve the phase shifter characteristics. Therefore, different

1.2-6 GHz phase shifters were designed with ±30! , ±45! , ±60! and ±90!

differential phase. The simulation and computation results show a maximum of 3! phase deviation in the 1.2-6 GHz frequency operating range. The return and insertion

loss are less than 10 dB and 1 dB, respectively. Such results can be successfully

compared to existing designs (Table 4-3), thus demonstrating the contributions made

in this area.

52

Table 4- 3 Comparison of the designed phase shifter’s specification with other phase shifters with different structures

Different phase shifters’ operating characteristics

Structure Ref. Operating

bandwidth

percentage

Operating

frequency

band (GHz)

Size

(Wave length)

Transmission

coefficient

Phase

ripple Phase shift

70 1.5-3.1 0.25 × 0.6 0.5 dB ±5! 90! Improved Schiffman’s structure by changing

the ground plane under the coupled line [50]

109.5 3.1-10.6 0.2 × 0.25 1 dB ±3! 30!, 45!

Multilayer structure with slot-coupled

Abbosh [51]

80 3-7 0.15 × 0.75 2.5 dB ±4! 90! Slot-coupled multilayer structure with loaded

ports of Abbosh’s coupler [52]

109.5 3.1-10.6 0.2 × 0.45 1.2 dB ±7! ±90!,±180!

Slot-coupled multilayer structure with short-

circuited and open-circuited stub [53]

105 0.75-2.4 0.3× 0.8 1.1 dB ±5! 90! Coupled-line single layer structure with open

stub with step impedance [54]

82 2.3-5.5 0.15 × 0.2 0.6 dB ±6.4! 90! Single layer microstrip by using the modern

loaded transmission line [55]

53

Different phase shifters’ operating characteristics

Structure Ref. Operating

bandwidth

percentage

Operating

frequency

band (GHz)

Size

(Wave length)

Transmission

coefficient

Phase

ripple Phase shift

105 3.1-10.6 0.5 × 0.8 1 dB ±9! 90! Single layer microstrip structure by using the

transmission line and electromagnetic model [56]

45 2.24-3.55 0.1× 0.5 0.9 dB ±5! 45

!,90!

135!

Multi-path single layer microstrip structure

with monolithic transmission line by using

the coupled line structure

[57]

137 1.2-6

0.2 × 0.43

(with

reference line)

1 dB ±2.1! ±30

!,±45!

±60!,±90!

Slot-coupled multilayer structure with

placing the elliptical-shaped load on

coupler’s port (we haven’t done process

variation)

our

work

54

4-6 Conclusion

In this chapter we designed reflective phase shifters in 1.2-6 GHz band. The

simulations are performed in HFSS and ADS. After the theoretical approach,

formulation and simulations, we compared the results with previous works. Also we

compared the measured and simulated results of the 45° differential phase shifter. The

simulation and computation results show a maximum of 3! phase deviation in the

operating frequency range. The return and insertion loss are less than 10 dB and 1 dB,

respectively.

55

Chapter 5

Conclusions and future works

5.1 Conclusions

In this thesis, we discussed about the design process of slot-couple wideband reflective

phase shifters. First, we detailed the concepts behind couplers and phase shifters. Then,

for each of them, we presented the related formulation to theoretically approximating

their geometrical dimensions in terms of the considered design parameters such as

coupling coefficient and required phase shift.

For band widening the phase shifter, the reflective phase shifter has been retained.

Therefore, a wideband coupler has been designed first using a modified Abbosh’s

structure. Such modification includes variation in the gap created on ground. This

change uniforms the coupling between transmissions lines on upper and lower sub

layers at, respectively, the minimum and maximum frequencies of the operating band.

To design the reflective phase shifter, we used the above coupler as its two ports are

loaded. Loading in ports has been achieved by placing a required reactance load (as

either lumped element or planar microstrip line).

The microstrip load first was simulated as an open circuit stub. Then we placed an

elliptical load, which was the best form in terms of operating bandwidth. All

simulations were performed in the full-wave HFSS software beside ADS. Our purpose

was to design a coupler with suitable characteristics such as providing the 3 dB

coupling in targeted 1.2 – 6 GHz frequency band with minimum return loss, phase and

amplitude imbalance while maximizing the operating bandwidth. This frequency

interval includes important frequency bands such as such as GPS, DCS, PCS, UMTS,

WiBro, Bluetooth, S-DMB, WiMax and WLAN. These frequencies have application

56

in satellite telecommunication, marine communication and mobile communication and

etc.

Once measured, the performance of the designed device was compared to recent works,

and, to the best of the author’s knowledge, there is no microstrip phase shifter in this

frequency range with the obtained bandwidth (137%). Furthermore, the designed phase

shifter’s characteristics are better than those mentioned in existing publications.

Furthermore, the designed phase shifter with elliptical load is capable of creating a

−90! to +90! phase shift on 1.2-6 GHz frequency range. Also, by changing the elliptical

shaped load dimensions, different phase shifts can be obtained.

Therefore, design and simulation of ±30! , ±45! , ±60! and ±90! differential phase

shifters were presented. The simulation and computation results show a maximum of

3° phase deviation, as well as return and insertion loss less than 10 dB and 1 dB

respectively.

5.2 Future works

As future direction to enhance this work, we can first use the stripline technology

instead of microstrip, since it has lower loss.

Also depending on the application needed, we can change our priorities. Here our main

priority was bandwidth.

Another option is changing the geometrical shapes of patch and slot based on

simulation’s feedbacks.

57

References

[1] Vijay K. Varadan, K. J. Vinoy, K. A. Jose, RF MEMS and their applications,

(Wiley, 2002) [2] Ian Robertson and Jim Turner, “MMIC Design”, Institution of Electrical

Engineers, Micheal Faraday House, London, 1995.

[3] Sunil Kansal, “4-Bit MMIC Digital Phase Shifter”, B.Eng thesis, Department of

Electrical Engineering, National University of Singapore, 1998/1999.

[4] A. M. Abbosh, "Ultra-Wideband Phase Shifters," in IEEE Transactions on

Microwave Theory and Techniques, vol. 55, no. 9, pp. 1935-1941, Sept. 2007.

[5] B. M. Schiffman, "A New Class of Broad-Band Microwave 90-Degree Phase

Shifters," in IRE Transactions on Microwave Theory and Techniques, vol. 6, no.

2, pp. 232-237, April 1958.

[6] J. P. Shelton and J. A. Mosko, "Synthesis and Design of Wide-Band Equal-

Ripple TEM Directional Couplers and Fixed Phase Shifters," in IEEE

Transactions on Microwave Theory and Techniques, vol. 14, no. 10, pp. 462-

473, Oct 1966.

[7] J. L. R. Quirarte and J. P. Starski, "Novel Schiffman phase shifters," in IEEE

Transactions on Microwave Theory and Techniques, vol. 41, no. 1, pp. 9-14, Jan

1993.

[8] Yong-Xin Guo, Zhen-Yu Zhang and Ling Chuen Ong, "Improved wide-band

Schiffman phase shifter," in IEEE Transactions on Microwave Theory and

Techniques, vol. 54, no. 3, pp. 1196-1200, March 2006.

[9] M. Sorn, R. Lech and J. Mazur, "Simulation and Experiment of a Compact

Wideband 90 Differential Phase Shifter," in IEEE Transactions on Microwave

Theory and Techniques, vol. 60, no. 3, pp. 494-501, March 2012.

[10] L. Guo and A. Abbosh, "Phase Shifters with Wide Range of Phase and Ultra-

Wideband Performance Using Stub-Loaded Coupled Structure," in IEEE

[11] A. Lüker, "Ferroelectric Phase Shifters," 2006. [Online]. Available:

http://microwaves101.com. Accessed: Aug. 15, 2016.

58

[12] E. Erkek, "S-Band Hybrid 4 Bit Phase Shifter Using Cots Components," MSC

Thesis, METU, Ankara, Turkey, 2009.

[13] Sunil Kansal, “4-Bit MMIC Digital Phase Shifter”, B.Eng thesis, Department of

Electrical Engineering, National University of Singapore, 1998/1999.

[14] S. K. Koul and B. Bhat, Microwave and Millimeter Wave Phase Shifters: Vol. I

Dielectric and Ferrite Phase Shifters, (Boston: Artech House, 1991).

[15] Vijay K. Varadan, K. J. Vinoy, K. A. Jose, RF MEMS and their applications, (Wiley, 2002)

[16] "Phase Shifter Fundamentals," 2013. [Online]. Available: http://www.rfmw.com.

Accessed: Aug. 15, 2016.

[17] I. Sarkas, “Analysis and Design of W-band Phase Shifters,” MSC Thesis,

University of Toronto, Toronto, Canada, 2010.

[18] S.Sasikumar and B. S. Narayanan, “Analysis of reflective type phase shifter for

millimeter wave application,” Signal Processing, Communication, Computing

and Networking Technologies (ICSCCN), 2011 International Conference on,

Thuckafay, 2011, pp. 342-346.

[19] T. Neil Ross, "Gallium Nitride Phase Shifters," PHD Thesis, Carleton University,

Ottawa, ON, Canada, 2015.

[20] "Loaded-Line Phase Shifters,”. [Online]. Available: http://microwaves101.com.

Accessed: Aug. 15, 2016.

[21] "High-pass/Low-pass Phase Shifters,”. [Online]. Available:

http://microwaves101.com. Accessed: Aug. 15, 2016.

[22] R. V. Garver, “Broad-Band Diode Phase Shifters,” in IEEE Transactions on

Microwave Theory and Techniques, vol.20, no. 5, pp.314-323, May 1972.

[23] C. Moye, G. Sakamoto and M. Brand, “A compact broadband, six-bit MMIC

phasor with integrated digital drivers,” Microwave Symposium Digest, 1990,

IEEE MTT-S International, Dallas, TX, 1990, pp. 457-460 vol.1.



[25] "Directional Couplers and Bridges,". [Online]. Available:

http://www.keysight.com/. Accessed: Aug. 15, 2016.

59



2, pp. 232-237, April 1958.

[27] T. Tanaka, K. Tsunoda and M. Aikawa, "Slot-coupled directional couplers on a

both-sided substrate MIC and their applications, "in Electronics and

Communications, vol. 72, no. 3, Japan 1989.

[28] A. M. Abbosh and M. E. Bialkowski, "Design of Compact Directional Couplers

for UWB Applications," in IEEE Transactions on Microwave Theory and

Techniques, vol. 55, no. 2, pp. 189-194, Feb. 2007.






473, Oct 1966.

[31] J. L. R. Quirarte and J. P. Starski, "Novel Schiffman phase shifters," in IEEE

Transactions on Microwave Theory and Techniques, vol. 41, no. 1, pp. 9-14, Jan

1993.









Microwave and Wireless Components Letters, vol. 24, no. 3, pp. 167-169, March

2014.

[35] S. H. Yeung, T. K. Sarkar, M. Salazar-Palma and A. Garcia-Lamperez, "A

Multisection Phase Correcting Network for Broadband Quadrature Power

Splitter Design," in IEEE Microwave and Wireless Components Letters, vol. 23,

60

no. 9, pp. 468-470, Sept. 2013.

[36] Y. J. Cheng, W. Hong and K. Wu, "Broadband Self-Compensating Phase Shifter

Combining Delay Line and Equal-Length Unequal-Width Phaser," in IEEE

Transactions on Microwave Theory and Techniques, vol. 58, no. 1, pp. 203-210,

Jan. 2010.

[37] S. Y. Zheng, W. S. Chan and K. F. Man, "Broadband Phase Shifter Using Loaded

Transmission Line," in IEEE Microwave and Wireless Components Letters, vol.

20, no. 9, pp. 498-500, Sept. 2010.



2, pp. 232-237, April 1958.




473, Oct 1966.






[42] M. F. Wong, V. F. Hanna, O. Picon and H. Baudrand, "Analysis and design of

slot-coupled directional couplers between double-sided substrate microstrip

lines," in IEEE Transactions on Microwave Theory and Techniques, vol. 39, no.

12, pp. 2123-2129, Dec 1991.

[43] Jeong-Hoon Cho, Hee-Yong Hwang and Sang-Won Yun, "A design of wideband

3-dB coupler with N-section microstrip tandem structure," in IEEE Microwave

and Wireless Components Letters, vol. 15, no. 2, pp. 113-115, Feb. 2005.

[44] H. C. Chen and C. Y. Chang, “Modified Vertically Installed Planar Couplers for

Ultrabroadband Multisection Quadrature Hybrid,” in IEEE Microwave and

Wireless Components Letters, vol. 16, no. 8, pp. 446-448, Aug. 2006.

[45] S. Gruszczynski and K. Wincza. “Broadband Rat-Race Couplers With Coupled-

61

Line Section and Impedance Transformers,” in IEEE Microwave and Wireless

Components Letters, vol. 22, no. 1, pp. 22-24, jan. 2012.

[46] A. M. Abbosh and M. E. Bialkowski, "Design of Compact Directional Couplers

for UWB Applications," in IEEE Transactions on Microwave Theory and

Techniques, vol. 55, no. 2, pp. 189-194, Feb. 2007.

[47] Nedil, M. and T. A. Denidni, "Analysis and design of an ultra wideband

directional coupler," Progress in Electromagnetics Research B, Vol. 1, 291-305,

2008.

[48] A. Moscoso-Martir, J. G. Wanguemert-Perez, I. Molina-Fernandez and E.

Marquez-Segura, "Slot-Coupled Multisection Quadrature Hybrid for UWB

Applications," in IEEE Microwave and Wireless Components Letters, vol. 19,

no. 3, pp. 143-145, March 2009.

[49] A. Moscoso-Martir, I. Molina-Fernandez, and A. Ortega-Monux, "High

performance multi-section corrugated slot-coupled directional couplers,"

Progress in Electromagnetics Research, Vol. 134, 437-454, 2013.











Microwave and Wireless Components Letters, vol. 24, no. 3, pp. 167-169, March

2014.




Jan. 2010.

62



20, no. 9, pp. 498-500, Sept. 2010.

[56] Q. Liu, Y. Liu, J. Shen, S. Li, C. Yu and Y. Lu, "Wideband Single-Layer 90

Phase Shifter Using Stepped Impedance Open Stub and Coupled-Line With

Weak Coupling," in IEEE Microwave and Wireless Components Letters, vol. 24,

no. 3, pp. 176-178, March 2014.

[57] S. H. Yeung, Z. Mei, T. K. Sarkar and M. Salazar-Palma, "Design and Testing

of a Single-Layer Microstrip Ultrawideband 90° Differential Phase Shifter," in

IEEE Microwave and Wireless Components Letters, vol. 23, no. 3, pp. 122-124,

March 2013.

[58] X. Y. Pu, S. Y. Zheng, J. Liu, Y. Li and Y. Long, "Novel Multi-way Broadband

Differential Phase Shifter With Uniform Reference Line Using Coupled Line

Structure," in IEEE Microwave and Wireless Components Letters, vol. 25, no. 3,

pp. 166-168, March 2015.

[59] C. R. Boyd, "A Dual-Mode Latching Reciprocal Ferrite Phase Shifter," in IEEE


1124, Dec 1970.

[60] B. Pillans, S. Eshelman, A. Malczewski, J. Ehmke and C. Goldsmith, "Ka-band

RF MEMS phase shifters," in IEEE Microwave and Guided Wave Letters, vol.

9, no. 12, pp. 520-522, Dec 1999.




473, Oct 1966.

[62] R. V. Garver, “Broad-Band Diode Phase Shifters,” in IEEE Transactions on

Microwave Theory and Techniques, vol.20, no. 5, pp.314-323, May 1972.

[63] A. Clavin, "Reciprocal Ferrite Phase Shifters in Rectangular Waveguide

(Correspondence), " in IRE Transactions on Microwave Theory and Techniques,

vol. 6, no. 3, pp. 334-334, July 1958.

63

[64] R. R. Jones, "A Slow Wave Digital Ferrite Strip Transmission Line Phase

Shifter," in IEEE Transactions on Microwave Theory and Techniques, vol. 14,

no. 12, pp. 684-688, Dec 1966.

[65] M. Abbosh, "Broadband Fixed Phase Shifters," in IEEE Microwave and Wireless

Components Letters, vol. 21, no. 1, pp. 22-24, Jan. 2011.

[66] Y. Wang, M. E. Bialkowski and A. M. Abbosh, "Double Microstrip-Slot

Transitions for Broadband Microstrip Phase Shifters," in IEEE

Microwave and Wireless Components Letters, vol. 22, no. 2, pp. 58-60, Feb.

2012.

[67] F. V. Minnaar, J. C. Coetzee and J. Joubert, "A novel ultrawideband microwave

differential phase shifter," in IEEE Transactions on Microwave Theory and

Techniques, vol. 45, no. 8, pp. 1249-1252, Aug 1997.

[68] X. Tang and K. Mouthaan, "Design Considerations for Octave-Band Phase

Shifters Using Discrete Components," in IEEE Transactions on Microwave

Theory and Techniques, vol. 58, no. 12, pp. 3459-3466, Dec. 2010.

[69] M. A. Morton, J. P. Comeau, J. D. Cressler, M. Mitchell and J. Papapolymerou,

"Sources of Phase Error and Design Considerations for Silicon-Based Monolithic

High-Pass/Low-Pass Microwave Phase Shifters," in IEEE Transactions on

Microwave Theory and Techniques, vol. 54, no. 12, pp. 4032-4040, Dec. 2006.

[70] R. V. Garver, "Broad-Band Diode Phase Shifters," in IEEE Transactions on

Microwave Theory and Techniques, vol. 20, no. 5, pp. 314-323, May 1972.

[71] T. Itoh and R. Mittra, “Spectral-Domain Approach for Calculating the Dispersion

Characteristics of Microstrip Lines (Short Papers),” in IEEE Transactions on

Microwave Theory and Techniques, vol. 21, no. 7, pp. 496-499, July. 1973.

[72] Q. Xiao, "A compact L-band broadband 6-bit MMIC phase shifter with low phase

error," Microwave Integrated Circuits Conference (EuMIC), 2011 European,

Manchester, 2011, pp. 410-413.

[73] X. Tang and K. Mouthaan, "Phase-Shifter Design Using Phase-Slope Alignment

With Grounded Shunt Stubs," in IEEE Transactions on Microwave Theory

and Techniques, vol. 58, no. 6, pp. 1573-1583, June 2010.

64

[74] S. Sim, L. Jeon and J. G. Kim, "A Compact X-Band Bi-Directional Phased-Array

T/R Chipset in 0.13 CMOS Technology," in IEEE Transactions on

Microwave Theory and Techniques, vol. 61, no. 1, pp. 562-569, Jan. 2013.

[75] K. Gharibdoust, N. Mousavi, M. Kalantari, M. Moezzi and A. Medi, "A Fully

Integrated 0.18- CMOS Transceiver Chip for -Band Phased-Array

Systems," in IEEE Transactions on Microwave Theory and Techniques, vol. 60,

no. 7, pp. 2192-2202, July 2012.

[76] M. Hangai, M. Hieda, N. Yunoue, Y. Sasaki and M. Miyazaki, " - and -

Band Ultra-Compact Phase Shifters Based on All-Pass Networks," in IEEE


Jan. 2010.

[77] Microwave and Wireless Components Letters, vol. 24, no. 3, pp. 167-169, March

2014.




Jan. 2010.



20, no. 9, pp. 498-500, Sept. 2010.

[80] Q. Liu, Y. Liu, J. Shen, S. Li, C. Yu and Y. Lu, "Wideband Single-Layer 90

Phase Shifter Using Stepped Impedance Open Stub and Coupled-Line With

Weak Coupling," in IEEE Microwave and Wireless Components Letters, vol. 24,

no. 3, pp. 176-178, March 2014.

[81] S. H. Yeung, Z. Mei, T. K. Sarkar and M. Salazar-Palma, "Design and Testing

of a Single-Layer Microstrip Ultrawideband 90° Differential Phase Shifter," in

IEEE Microwave and Wireless Components Letters, vol. 23, no. 3, pp. 122-124,

March 2013.

[82] X. Y. Pu, S. Y. Zheng, J. Liu, Y. Li and Y. Long, "Novel Multi-way Broadband

Differential Phase Shifter With Uniform Reference Line Using Coupled Line

65

Structure," in IEEE Microwave and Wireless Components Letters, vol. 25, no. 3,

pp. 166-168, March 2015.




[84] R.C. Johnson, Antenna Engineering Handbook, third edition, McGraw-Hill,

New York, 1993.

[85] DR. John and L.Volakis, Antenna Engineering Handbook, forth edition,

McGraw-Hill, 2007.

[86] M. Vahdani, Low- profile, Ultra Wideband and Dual Polarized Antenna and

Feeding Systems, Télécom ParisTech, 2008.

[87] R. G. Corzine and J.A. Morkel, Four-Arm Spiral Antenna, Artech House,

Norwood, MA, 1990.

[88] C. Free and C. Aitchson, "Improved analysis and design of coupled—Line phase

shifters", IEEE Trans. Microw. Theory Tech., vol. 43, no. 9, pp. 2126-2131, 1995.

[89] T. Ozdemir, J. L. Volakis and M. W. Nurnberger, "Analysis of thin multioctave

cavity-backed slot spiral antennas," in IEE Proceedings - Microwaves, Antennas

and Propagation, vol. 146, no. 6, pp. 447-454, Dec 1999.

[90] V. Rumsey, "Frequency independent antennas," 1958 IRE International

Convention Record, New York, NY, USA, 1957, pp. 114-118.

[91] D. M. Pozar, Microwave Engineering, 3rd ed. New York: Wiley, 2005.

[92] M. F. Wong, V. F. Hanna, O. Picon and H. Baudrand, "Analysis and design of

slot-coupled directional couplers between double-sided substrate microstrip

lines," in IEEE Transactions on Microwave Theory and Techniques, vol. 39, no.

12, pp. 2123-2129, Dec 1991.

[93] H. C. Chen and C. Y. Chang, "Modified Vertically Installed Planar Couplers for

Ultrabroadband Multisection Quadrature Hybrid," in IEEE Microwave and

Wireless Components Letters, vol. 16, no. 8, pp. 446-448, Aug. 2006.

[94] S. Gruszczynski and K. Wincza, "Broadband Rat-Race Couplers with Coupled-

Line Section and Impedance Transformers," in IEEE Microwave and Wireless


66

[95] S. M. Sherman, Monopulse Principles and Techniques. Dedham, Artech House,

Dedham, MA, 1984.

[96] G. M. Kirkpatrick, "Development of A Monopulse Radar System," in IEEE

Transactions on Aerospace and Electronic Systems, vol. 45, no. 2, pp. 803-818,

April 2009.

[97] S. Gruszczynski and K. Wincza, "Broadband Rat-Race Couplers with Coupled-

Line Section and Impedance Transformers," in IEEE Microwave and Wireless


[98] N. S. Barker and G. M. Rebeiz, "An octave bandwidth monopulse processor,"

Microwave Symposium Digest, 1997., IEEE MTT-S International, Denver, CO,

USA, 1997, pp. 405-407 vol.2.

[99] J. G. Kim, S. Raman and G. M. Rebeiz, "A 24 GHz Amplitude Monopulse

Comparator in 0.13 m CMOS," in IEEE Microwave and Wireless Components

Letters, vol. 18, no. 9, pp. 632-634, Sept. 2008.

[100] S. M. Sherman, Monopulse Principles and Techniques. Dedham, Artech House,

Dedham, MA, 1984.

[101] R. G. Corzine and J.A. Morkel, Four-Arm Spiral Antenna, Artech House,

Norwood, MA, 1990.

[102] Y. W. Wang, G. M. Wang, Z. W. Yu, J. G. Liang and X. J. Gao, "Ultra-Wideband

-Plane Monopulse Antenna Using Vivaldi Antenna," in IEEE Transactions

on Antennas and Propagation, vol. 62, no. 10, pp. 4961-4969, Oct. 2014.

[103] A. S. Abdellatif et al., "Wide-band phase shifter for mmWave phased array

applications," Millimeter Waves (GSMM), 2015 Global Symposium On,

Montreal, QC, 2015, pp. 1-3.

[104] Y. Wu, L. Yao, W. Wang and Y. Liu, "A wide-band 180-degree phase shifter

using a pair of coupled-line stubs," 2015 IEEE International Symposium on

Antennas and Propagation & USNC/URSI National Radio Science Meeting,

Vancouver, BC, 2015, pp. 240-241.

[105] S. Navin Andrew Prince, P. Muthukumaran and N. Jagatheesh, "Design and

implementation of two 1-bit switched line phase shifter," Advanced

Communication Control and Computing Technologies (ICACCCT), 2014

67

International Conference on, Ramanathapuram, 2014, pp. 497-501.

[106] Y. M. Yang, C. W. Yuan and B. L. Qian, "A Novel Phase Shifter for Ku-Band

High-Power Microwave Applications," in IEEE Transactions on Plasma

Science, vol. 42, no. 1, pp. 51-54, Jan. 2014.

[107] E. Atasoy, F. Piri and B. S. Yarman, "Symmetric lattice 45°, 90° and 180° digital

phase shifter at 3–6 GHz for LTE, WIFI, Radar applications," Signals, Circuits

and Systems (ISSCS), 2015 International Symposium on, Iasi, 2015, pp. 1-4.

[108] A. J. Abumunshar, W. G. Yeo, N. K. Nahar, D. J. Hyman and K. Sertel, "Tightly

coupled Ka-band phased array antenna with integrated MEMS phase shifters,"

Radio Science Meeting (Joint with AP-S Symposium), 2014 USNC-URSI,

Memphis, TN, 2014, pp. 104-104.

[109] S. Y. Zheng and W. S. Chan, "Differential RF Phase Shifter with Harmonic

Suppression," in IEEE Transactions on Industrial Electronics, vol. 61, no. 6, pp.

2891-2899, June 2014.

[110] B. Muneer, Z. Qi and X. Shanjia, "A Broadband Tunable Multilayer Substrate

Integrated Waveguide Phase Shifter," in IEEE Microwave and Wireless

Components Letters, vol. 25, no. 4, pp. 220-222, April 2015.

[111] S. Wen, Q. Wang, Tao Wu and Y. Tan, "Design of a compact 3dB Ka-band

directional coupler," Microwave and Millimeter Wave Circuits and System

Technology (MMWCST), 2012 International Workshop on, Chengdu, 2012, pp.

1-4.

[112] M. Gillick, I. D. Robertson and J. S. Joshi, "A 12-36 GHz MMIC 3dB Coplanar

Waveguide Directional Coupler," Microwave Conference, 1992. 22nd European,

Helsinki, Finland, 1992, pp. 724-728.

[113] Z. Wei, E. Li, G. Guo and J. Wu, "Design of 3dB wideband multisection

quadrature directional coupler," Consumer Electronics, Communications and

Networks (CECNet), 2012 2nd International Conference on, Yichang, 2012, pp.

2518-2520.

[114] M. Moosazadeh, A. M. Abbosh and Z. Esmati, "Design of compact planar

ultrawideband antenna with dual-notched bands using slotted square patch and

pi-shaped conductor-backed plane," in IET Microwaves, Antennas &

68

Propagation, vol. 6, no. 3, pp. 290-294, February 21 2012.

[115] Rashid Ahmad Bhatti, Hyung-Sik Yoon and Seong Ook Park, "Design of a novel

multiband internal antenna for personal communication handsets," TENCON

2007 - 2007 IEEE Region 10 Conference, Taipei, 2007, pp. 1-3.

[116] B. Yu, J. Byun, and B. Lee, "T-shaped compact mobile handset antenna for

wideband applications, " Microw. Opt. Technol. Lett., 2008

[117] J. Jung, H. Lee and Y. Lim, "Modified meander line monopole antenna for

broadband operation," in Electronics Letters, vol. 43, no. 22, pp., Oct. 25 2007.

design of a multiband microstrip differential phase ... · 2-6 phase shifter designs ......

Documents