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Application Note 22Issue 1 May 1996
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High Frequency DC-DC Conversion using HighCurrent Bipolar Transistors
Neil ChaddertonDino Rosaldi
Introduction
DC-DC conversion is one of thefundamental circuit functions within theelectronics industry and addresses awide f ie ld o f market sectors ,applications and supply requirements. Ageneral trend, (to pursue cost, size andreliability advantages) has been toreduce the size of the DC-DC converter
systems, and as the inductive elementsof such a system are quite often thebulkiest components, increases inswitching frequency provide onemethod to a l low th is . Switch ingfrequencies therefore have increasedfrom tens of kHz to hundreds of kHz,which has forced a revision of theenabling switching devices.
It is an often assumed maxim thatbipolar transistors are useful for DC-DCconversion functions up to perhaps50kHz, and for service beyond thisfrequency, that MOSFETs provide theonly solution. The purpose of thisapplication note is to demonstrate thatbipolar transistors possessing superiorgeometrys - thereby providing a highercurrent capability per die area, can andare operated to reasonably h igh
switching frequencies with minimalloss. This often allows a more compact
design (due to the higher sil iconefficiency of bipolar technology) andlower cost.
Background
To obtain the most efficient performancefrom bipolar t ransis tors a t h igh
switching frequencies, an appreciationof the basic switching mechanisms andbase charge analysis is useful. AppendixA provides an introduction to bipolartransistor switching behaviour, andappendix B provides references formathematical treatments.
There are various methods forincreasing the maximum switchingspeed of bipolar transistors. Some of
these rely on preventing saturation ofthe device, thus vastly reducing thestored charge, while other methodsremove the charge at turn-off. Figures 1,2 and 5 show a number of commonspeed-up networks. Figures 1a through1d show various options for preventingsaturation, and effectively reduce/limitbase drive when the collector terminalhas fallen to a specific level. (Figure 1a is
often termed a Schottky transistor, andis often used for IC transistors, and
400kHz Operation with Optimised Geometry Devices
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Figures 1c and 1d are variants of the socalled Baker clamp circuit).
These methods (the Baker clamp beingpreferred for high power applications) ofcourse do not allow the transistorscollector-emitter voltage to saturate to avery low level, and so the resultanton-state loss may be high and thereforeprohibit the use of smaller packaged,though adequately current capabledevices. Figures 2 and 5 show twomethods that allow true saturation bypermitting sufficient forward base drive,while removing charge quickly at turn-off.Design of such networks is non-critical,and by suitable choice of componentsallow high levels of base overdrive with nopenalty to turn-off time duration.
Figure 2 shows a method of speeding upPNP devices to enable replacement oflarge P-Channel MOSFETs - this particularcircuit being designed for fast charging ofbattery packs by BenchmarqMicroelectronics. Figures 3 and 4 show therelevant waveforms, for a standardpassive turn-off method (base-emitterpull-up resistor) and the speed-up circuitof Figure 2 - the combined storage and falltimes being reduced from 1.2s to 80ns.Importantly, the major switching losscontributor - the fall time, has beensignificantly reduced to 40ns, which iscomparable to, or better than P-ChannelMOSFET performance. This allows costeffective replacement of large packageddevices.
Application Note 22Issue 1 May 1996
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Q2
Q1
Q1
Q1
R1
R2
R3C1
D1
D2
D3
Q1
a b
c d
Figure 1.
Non-saturating Bipolar Transistor Speed-up Methods.
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Application Note 22Issue 1 May 1996
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Q12N3904
1K
L110uH
2N3904
1N5820
L2
200uH
100uF
+Bat
1N4148
Q3
ZTX789AInput
Drive
Feedback
Q2
330
Figure 2.PNP Transistor Speed-up Circuit to Allow Replacement of P-Channel MOSFETs withinHigh Current Converters. (The circuit shown is suitable for output currents up to 1.5A;other variants are capable of operation to 5A output).
Figure 3.Turn-off Waveforms for PNP Step-downConverter using Passive Turn-off.Upper Trace - Collector-to-0V, 5V/div;Middle Trace - PNP base-to-0V;Lower Trace - PWM IC Output, 5v/div.Timebase at 2s/div.
Figure 4.Turn-off Waveforms for PNP Step-downConverter (of Figure 2) using ActiveTurn-off.Upper Trace - Collector-to-0V, 5V/div;Middle Trace - PNP base-to-0V;Lower Trace - PWM IC Output, 5v/div.
Timebase at 2s/div.
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Application Note 22Issue 1 May 1996
FMMT718
22pF
LL5818
220uH
30K
470uF
1000uF
150
1.8n
PWM IC
+ve
-ve
Osc
Gnd
E
CILim
Vin
Vin
0V
+5V/+3.3V
0V
0.05
220
470K
220pF 13K7K5/
(Set Vout)
Figure 9.Basic Step-Down DC-DC Converter using the LM3578 and the FMMT718.(With values shown, Fop=50kHz, IB=43mA, peak efficiency=88%).
0.01 10Output Current (A)
Efficiency v Output Current
10
100
Efficiency(%)
0.1 1
20
40
60
80
1
2
3
5 15
Efficiency(%)
100
0
Input Voltage (V)
Efficiency v Input Voltage
20
40
60
80
7 9 11 13
Figure 10.Efficiency against Load Current for theConverter of Figure 9, with IB as aparameter.Curve 1 - IB=9.4mA; Curve 2 - IB=43mA;Curve 3 - IB=170mA. Fop=50kHz; Vin=7V;Vout=5V.
Figure 11.Efficiency against Input Voltage for theConverter of Figure 9. (IB=43mA; Iout=1A).
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Application Note 22Issue 1 May 1996
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Optimisation, and some scope for higherfrequency operation is possible with thisbasic converter. Figure 12 shows theeffect of varying the inductors value,with the switching frequency set to90kHz. Turn-on time was recorded at55ns, and turn-off at 1.5s.
To investigate options for higherswitching frequencies requires a changeto the PWM controller IC. Figure 13shows a circuit based on the TI5001device. This IC is capable of operation upto a switching frequency of 400kHz, andcan sink a maximum of 20mA into the VOpin.
The initial circuit was configured tooperate at 150kHz, with the FMMT718base current set to 9mA by a 560 baseresistor (R1) - effectively a forced gain of
222 at 2A. Turn-on and turn-off timeswere measured at 200ns and 1.44s
0.01 10
Output Current (A)
Efficiency v Output Current
10
100
Efficiency(%)
0.1 1
20
40
60
80
1
2
Figure 12.Efficiency against Load Current for theConverter of Figure 9.IB=43mA; Fop=90kHz; Vin=7V; Vout=5V.
Effect of Variation of Inductor Value.Curve 1 - 25H, Curve 2 - 180H.
TL5001
Vcc
Vo
Comp
F/b
Gnd
SCP
DTC
RT
ZTX718
4.7nF
R1
560
1000uF
470uF
L1 40uH
D1
1N5818
Rf
22k
2k2
5k6
5k6
Vin
Vout
10nF
0.1uF
Figure 13.Basic Step-Down DC-DC Converter using the TI5001 and the FMMT718. (With values
shown, Fop=150kHz, IB=9mA, peak efficiency=81%).
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Application Note 22Issue 1 May 1996
0.01 10
Efficiency(%)
100
0
Output Current (A)
Efficiency v Output Current
20
40
60
80
0.1 1
1
2
Figure 14.Efficiency against Load Current for theConverters of Figure 13 and 16.
Q1
R1
Q2
R2
C1
D1
PWM IC
Figure 15.Bipolar Transistor Turn-off Circuit to allowCapacitive Turn-off Charge Neutralisationwith PWM Controller IC.
TL5001
Vcc
Vo
Comp
F/b
Gnd
SCP
DTC
RT
4.7nF
Rf
2k2
5k6
5k6
VinVout
0.1uF
10nF
560
FMMT718
390
2.2nF
1N4148
470uF
D1
1N5818
L1 40uH
FMMT3904
1000uF
22k
Figure 16.Basic Step-down DC-DC Converter using the TI5001/FMMT718 Combination with
Capacitive Turn-off Circuit.
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Application Note 22Issue 1 May 1996
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respectively. The chart shown in Figure14 (curve 1) illustrates the resultant
efficiency versus load profile. Thecircuits conversion efficiency peaks at81% at 23mA output, but falls thereafterto 71% at 2A, due mainly to switchinglosses, though in some part toincreasing VCE(sat) at higher currents. Thelatter factor could be addressed in partby considering the load requirements,and opt imis ing for the re levantcondition, but the switching losseswould remain.
By considering the base turn-off chargeand effecting a suitable turn-off circuit(Figure 15) the circuit shown in Figure 16was produced. This shows a significantimprovement in conversion efficiency atmedium to higher currents; as shown in
Figure 17.Switching Waveforms for the circuit ofFigure 16, operating at 150kHz. Uppertrace; 2V/div, Collector-to-0V. Lowertrace; 2V/div, Base-to-0V. Vin=7V,Vout=5V, IL=220mA.
0.01 10
Efficiency(%)
100
0
Output Current (A)
20
40
60
80
0.1 1
1
2 4
3
5
Figure 18.Efficiency against Load Current for the Converter of Figure 16. Curve 1 - F op=150kHz;curve 2 - 220kHz; curve 3 - 300kHz; curve 4 - 400kHz. I B=10mA. Curve 5 - 400kHz andIB=5mA.Vin=7V, Vout=5V.
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Figure 14, curve 2. This shows a peak
efficiency of 92%. Figure 17 shows theswitching waveforms, including thecollector-to-0V waveform - note therapid turn-off edge; this was measuredto be 25ns. The efficiency at lowercurrents has of course reduced, due tothe extra current taken by the turn-offcircuit. This efficiency profile can bemodified for specific applications bysacrificing high current efficiency infavour of low current performance or
vice-versa.
Further modifications were effected toassess performance at still higherswitching frequencies. Figure 18 showsthe efficiency/load current profiles forswitching frequencies from 150kHz to400kHz. Figure 19 shows the efficiencyagainst input voltage for the 220kHz
Application Note 22Issue 1 May 1996
155
Efficiency(%)
100
0
Input Voltage (V)
Efficiency v Input Voltage
20
40
60
80
7 9 11 13
Figure 19.Efficiency against Input Voltage for theConverter of Figure 16.Fop=220kHz; Vout=5V; load current=1A.
Figure 20.Switching Waveforms for the Converterof Figure 16 operating at 400kHz.Turn-on edges. Risetime= 20ns.Upper trace - Base-to-0V, 2V/div.Lower trace - Collector-to-0V, 2V/div.Timebase at 50ns/div.
Figure 21.Switching Waveforms for the Converterof Figure 16 operating at 400kHz.Turn-off edges. Falltime=30ns.Upper trace - Base-to-0V, 2V/div.
Lower trace - Collector-to-0V, 2V/div.Timebase at 50ns/div.
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Application Note 22Issue 1 May 1996
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version - the curve varying little over the
measured range. Figure 18 curve 5, isagain for a switching frequency of400kHz, but with lower base drive(RB=680) and a reduction to 1.5nF forthe turn-off capacitor. The oscillographsshown in Figures 20 and 21 show theturn-on and turn-of f swi tch ingwaveforms for the 400kHz version.These show collector rise and fall timesof 20ns and 30ns respectively.
Conclusions
This application note has demonstratedthat with due attention to the base andcollector charge phenomena in bipolartransistors, the operating switchingfrequency of those parts can beextended well beyond the currentlyaccepted notional maximum of 40kHz, toseveral hundred kHz. Various speed-up
methods have been summarised, andexamples of those particularly suitableand complementary to high currentcapable transistors examined, andcircuit examples presented.
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charge necessary for small signal typesZTX107, ZTX300 and ZTX500, andswitching transistors ZTX310 andZTX510.
Application Note 22Issue 1 May 1996
Figure 25.Trigger Waveforms and Effect of VaryingTrigger Capacitance. Upper trace - Inputwaveform; lower traces - output atcollector, (a) Variable capacitor C1 belowcritical value, (b) C1 at critical value withcollector voltage rising to 90% of final value,(c) C1 above critical value. Horizontalscale=200ns/div, Vertical scale=2V/div.
ZTX300, IB=1mA, IC=10mA.
Q1
RCRB
+5V
To Oscilloscope
0VInput
C1
Figure 24.Circuit to Determine Minimum Value of Trigger Charge for Bipolar Transistor.
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100u 1m 10mIB - Base Current (A)
Min. Trigger Charge v IB for Small Signal
and Switching Transistors
10p
10n
MinimumT
riggerCharge(C)
ZTX310100p
1n
abc
cba
ZTX107ZTX300ZTX500
ZTX510
Figure 26.Minimum Trigger Charge for Small Signaland Switching Transistors. Forced Gains(IC/IB) of : a)10; b)20; c)40.No significant difference observed fordifferent forced gains on the ZTX310 series.
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Application Note 22Issue 1 May 1996
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Appendix BReferences
(References 1 , 2 and 3 conta inmathematical treatment of chargeanalysis with respect to switchingcharacteristics of bipolar transistors -similar background can be found in mostsemiconductor physics books).
1. Switching Transistor Handbook,chapter five - Transient Characteristicsof Transis tors . Motoro la Inc . ,
Technical Editor William D. Roehr.
2 . Pu lse , Dig i ta l and Switch ingWaveforms, Chapter 20 - Transient
Switching Characteristics, Millman andTaub. McGraw-Hill Book Company.
3. Semiconductor Device Technology,section 2.2.26 BJT Modelling - PhysicalBJT Models, subsection 2) ChargeControl BJT Model. Malcolm E. Goodge.MacMillan Press.
4. Switching and Linear Power Supply,Power Converter Design. Abraham I.
Pressman. Hayden Press.
Appendix C
Characterisation Charts showing typical stored charge as a function of load currentand bias level.
0.1 1 10
IC - Collector Current (A)
Stored Charge v IC for ZTX869
1n
10n
100n
StoredCha
rge(C) IC/IB=50
IC/IB=100
IC/IB=150
ZTX869
IC/IB=150IC/IB=100
IC/IB=50
ZTX949
StoredCha
rge(C)
10n
1n
0.1
100p
1 10IC - Collector Current (A)
Stored Charge v IC for ZTX949
StoredCharge(C)
100n
100p0.1 1 10
IC - Collector Current (A)
Stored Charge v IC for ZTX968
IC/IB=150IC/IB=200
IC/IB=100IC/IB=50
ZTX968
1n
10n
100n
1n
0.1IC - Collector Current (A)
Stored Charge v IC for ZTX1048A
1 10
StoredCharge(C)
ZTX1048A
IC/IB=50
IC/IB=100
IC/IB=150
IC/IB=20010n
IC/IB=200
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Application Note 22Issue 1 May 1996
0.1 1 10
IC/IB=50
StoredCh
arge(C)
100n
100p
IC - Collector Current (A)
Stored Charge v IC for ZTX618
10n
1n
100p
10n
1n
100p
100u 1m 10m
IB - Base Current (A)
Min. Trigger Charge v IB for Small Signaland Switching Transistors
10p
10n
Minim
umT
riggerCharge(C)
ZTX310100p
1n
abc
cba
StoredCharge(C)
10n
1n
100p
10p
100u 1m 10m
IB - Base Current (A)
Stored Charge v IB for Small Signal and
Switching Transistors
ZTX310
abc
ba
c
1n
10n IC/IB=100IC/IB=150
IC/IB=200
ZTX618
StoredCh
arge(C)
ZTX717 IC/IB=50
IC/IB=100
IC/IB=150
IC/IB=200
IC - Collector Current (A)
Stored Charge v IC for ZTX717
0.1 1 10
StoredCharge(C)
0.1 1 10
IC/IB=200
IC/IB=150
IC/IB=100
IC/IB=50
ZTX718
IC - Collector Current (A)
Stored Charge v IC for ZTX718
100n
10n
1n
ZTX849
StoredCharge(C)
IC - Collector Current (A)
Stored Charge v IC for ZTX849
IC/IB=150
IC/IB=100
IC/IB=50
0.1 1 10
ZTX500ZTX300ZTX107
ZTX510
ZTX107ZTX300ZTX500
ZTX510
Forced gains (IC/IB) of a:)10, b)20, C) 40.
No significant difference observed for the different forced gains for the ZTX310 series
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Appendix C (continued)