improvement of light load efficiency for buck-...
TRANSCRIPT
978-1-5386-6054-6/18/$31.00 ©2018 IEEE
Improvement of Light Load Efficiency for Buck-
Boost DC-DC converter with ZVS using
Switched Auxiliary Inductors
Hayato Higa
Dept. of Energy Environment Science
Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japan
Akira Sagawa
Dept. of Electric Information
Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japan
Jun-ichi Itoh
Dept. of Electric Engineering
Nagaoka University of Technology
Nagaoka, Niigata, Japan
Abstract—This paper proposes a changing buck-boost
inductance for non-isolated bidirectional buck-boost DC-DC
converter with zero voltage switching (ZVS) modulation in
order to achieve high efficiency at wide load region and a wide
voltage variation. In the proposed converter, the auxiliary
inductor and the bidirectional switch are connected in parallel
to a main inductor, and each connection is switched depending
on the load condition. At the light load region, the bi-directional
switch is turned off for the reduction of the converter loss with
the large equivalent inductance. On the other hands, the
auxiliary circuit is utilized at the heavy load region in order to
extend the power translation range with the small equivalent
inductance. In addition, the sequence for the switching auxiliary
inductor is also proposed in order to prevent the surge voltage
of the bidirectional switch and the DC offset current in the
auxiliary inductor. From the experimental results, the root
mean square value of the inductor current is reduced by up to
23.8% compared with that of the conventional converter. In
addition, the validity of the proposed sequence is also confirmed,
e.g. no surge voltage at the turn off of the auxiliary inductor.
Moreover, 41% of the converter loss reduction at the light load
region.
Keywords— Bidirectional DC-DC converter; Zero voltage
switching; switching inductance
I. INTRODUCTION
Recently, energy storage systems (ESS) have been applied in DC micro-grid systems [1]-[3]. In ESS, a bidirectional buck-boost DC-DC converter is generally required to obtain the bidirectional operation regardless of DC-bus and battery voltage conditions. In a power converter for ESS, the power fluctuation due to renewable energy resources is compensated. In addition, the power oscillation can vary over a wide range of level, whereas the battery voltage in ESS fluctuates due to a charge and discharge operation. Hence, it is crucially required for ESS to be able to achieve the high efficiency in wide load range and under any condition of battery voltage variation [4]. However, the typical buck-boost converter topology with a continuous current mode is the low overall efficiency due to the switching loss of the hard switching operation [5].
In order to reduce the switching loss, the soft switching methods have been actively researched [6]-[8]. In the reference of [6], a zero voltage switching (ZVS) techniques which utilize a resonance between additional components and the junction capacitors have been proposed. In this method, the duty ratio range is limited by the resonance period. Besides, a zero voltage transient (ZVT) circuit is also proposed in order
to achieve ZVS with entire load region by [7], [8]. By this method, the inductor current flow is controlled in order to discharge the junction capacitor of the switching devices. However, the additional conduction and the switching losses occur in ZVT circuit, i.e. the lower converter efficiency.
As different approaches, there are several modulation methods to achieve the soft switching [9-11], such as a triangular-current mode (TCM) applied for the achievement of ZVS [9], [10]. However, the converter efficiency at the light load becomes lower due to the high current ripple. Meanwhile, the control method to achieve ZVS for a four-switch-buck-boost DC-DC converter without additional components has been proposed [11]. In this method, the current ripple is reduced compared to that of TCM. However, the current ripple is still large in order to achieve ZVS over entire load. Furthermore, there is tradeoff between the current ripple and the high power capability. In other words, the problem in this method is low efficiency at the light load.
This paper proposes adding an auxiliary circuit with small inductors in order to achieve the high efficiency in wide load conditions including the high power capability. In particular, the bidirectional switches and the auxiliary inductors are connected parallel to the main inductor in order to reduce the current ripple at the light load. The originality of this paper is changing the equivalent inductance depending on the voltage variation and the load condition. Note that the current rating of the bi-directional switches and all inductors become small because the auxiliary inductors are connected in parallel to the main inductor. Moreover, sequence method for the switching auxiliary inductor is proposed to prevent the occurrence of DC-offset current in the auxiliary-inductor current. This paper is organized as follows; first, the circuit configuration and the operation modes with the ZVS achievement are explained. Second, the auxiliary circuit with the small inductors is introduced. Third, the proposed switching sequence is introduced. Finally, the effectiveness of the proposed circuit and sequence are confirmed by the experimental results.
II. CIRCUIT CONFIGURATION AND OPERATION
PRINCIPLE
A. Circuit Configration
Figure 1 shows the circuit configuration of the four-switch buck-boost converter with the switched auxiliary small inductors. In order to minimize the inductors, the switching frequency is increased. This leads to the increase in the switching loss due to the higher switching frequency. In order
to avoid this problem, the soft switching method is employed. In addition, the auxiliary inductors are changed in accordance with the output power.
B. Operation princeple
Figure 2 shows the bidirectional operation waveforms of the switching period with the ZVS achievement [10]. In both power flow operation, there are four modes in the switching period. In order to achieve ZVS, the offset of the inductor current I0 is maintained during the zero-current interval of a conventional discontinuous current mode. The offset current for the ZVS achievement I0 is calculated by
0
2
sin2
dsin
dall
ds all
CI V
TL
C L
where Cds is the junction capacitor of MOSFET, Lall is the total inductance including the main inductance L and the auxiliary inductors.
In addition, the transferred power is determined by the secondary voltage v2 and the inductor current. Thus, the transferred power is calculated by
20
1 2 0 1
2 2 2
2 1 2
1( ) ( )
2
2 2
4
L
bat SW allbus
SW all bus bat bat
P v i d
V L IV
L V V V
Next, the each switching timing1-3 is calculated by (3)-(5).
2
0 31 2 2
SW bus bat
bus bus bat bat
LI V V
V V V V
2 3 1( )bat
bus
V
V
03
2
0 2 2
2 2
( )
4
SW all bus bat
bus bat
SW SW all ref
bus bus bat bat
bus bat
L I V V
V V
LI L PV V V V
V V
where sw is the switching angular frequency, Pref is the reference transferred power.
It should be noted that the total inductance Lall is decided by the number of the auxiliary inductors. In the discharge operation, the switching timing of S1 and S3 are switched. At
the condition of3=the maximum transferred power is achieved. The maximum average power Pmax can be calculated by
max 2 2
2
0 0 ( )4
bus bat
bus bus bat bat
SW all bus batbus bat
SW all
V VP
V V V V
L V VI I V V
L
(6)
As shown in (6), the maximum transferred power which can still achieve ZVS can be increased when the inductor value becomes small.
Figure 3 shows the inductor current waveforms at the light load condition. With the large inductance, the peak current can be reduced at the light load compared to that of the small inductor value. In addition, the offset current for ZVS I0 is also
Vbus
Vbat
SL1 SL2S1
S2
S3
S4iLL
L1
Lm
v1 v2
Fig. 1. Circuit configuration of bidirectional buck-boost
converter with switched auxiliary inductors. The DC bus voltage
varies from 300 V to 400 V, whereas the battery voltage changes
from 300 V to 350 V.
iL
iL
0
0
0
0 231 2
Ch
arg
e o
per
atio
nD
isch
arg
e o
per
atio
n
S1,S2
0
0
Mode 1 Mode 2 Mode 3 Mode 4
S3,S4
S1,S2
S3,S4
-I0
I0
0
Fig. 2. Operation waveforms of switching period with ZVS
achievement. In this method, the offset current I0 is generated in
order to satisfy ZVS condition. In the discharge operation, the
switching timing of S1 and S3 are switched.
reduced by increasing inductance from (1). Figure 4 shows the root mean square (RMS) current of the inductor current. RMS value of the inductor current is reduced by the large inductance. The problem with only one main inductor is that the maximum power becomes small when the inductance is large. In the conditions of the low inductance, the current ripple at the light load increases, i.e. the lower efficiency at the light load. In order to solve this tradeoff relationship between the power capability and the light load efficiency, the inductance is changed in accordance with the transferred power.
Figure 4 shows the root mean square (RMS) current of the
inductor current. RMS value of the inductor current is
reduced by the large inductance. The problem with only one
main inductor is that the maximum power becomes small
when the inductance is large. In the conditions of the low
inductance, the current ripple at the light load increases, i.e.
the lower efficiency at the light load. In order to solve this
tradeoff relationship between the power capability and the
light load efficiency, the inductance is changed in accordance
with the transferred power.
III. SWITCHING SEQUENCE OF AUXILIARY
INDUCTOR
At the switching auxiliary inductor, the DC offset current might occur at the turn-on timing of the auxiliary switches. Furthermore, the surge voltage occurs in accordance with the turn-off timing of the auxiliary inductor. In order to prevent the occurrence of the surge voltage and the DC-offset current, the following switching sequence is proposed.
Figure 5 shows the switching sequence for the switching auxiliary inductor. Fig. 5 (a) and (b) show the turn-on and the
iL :2L
iL :L
-I0
0
Peak current is reduced Imax_L
Imax_2L
Fig. 1. Inductor current waveforms at light load when the
inductance is changed. The peak current and RMS value are
reduced by increasing the inductance.
I Lrm
s [A
]Transferred power [W]
0.0
1.0
2.0
3.0
4.0
5.0
0 200 400 600 800 1000
L=357mH
L=268mH
L=536mH
Vin = 400 V
Vout = 300 V
Maximum value of transferred power
Fig. 2. RMS value of inductor current. At the light load, RMS value is
reduced in proportion of the inductance. The larger inductance leads
to the smaller transferred power. Therefore, the inductor value is
changed by switching the auxiliary inductors.
Carrier of input side
Carrier of output side
0
SL1
SL2
SL1
SL2
SL1
SL2
SL1
SL2
SL1
SL2
SL1
SL2
Carrier of input side
Carrier of output side
0
L
L
SL1
SL2
SL1
SL2
OFF
OFF
ON
ON
SL1
SL2
SL1
SL2
SL1
SL2
SL1
SL2
0
Mode 1 Mode 2 Mode 1 Mode 2 Mode 3 Mode 4Mode 1 Mode 2 Mode 1 Mode 2 Mode 3 Mode 4
I0
-I0
SL1
SL2
Dout
0
H
H
Din
0
(a) Turn on at discharge operation (b) Turn off at charge operation (c) Turn on at discharge operation (d) Turn off at discharge operation
Fig. 5. In the proposed switching sequence, the current detection is not required because the switching timing is synchronized to the
switching pattern of S1 to S4.
iL :2L
iL :L
-I0
0
Peak current is reduced
Imax_L
Imax_2L
Fig. 3. Inductor current waveforms at light load when the
inductance is changed. The peak current and RMS value are
reduced by increasing the inductance.
I Lrm
s [A
]
Transferred power [W]
0.0
1.0
2.0
3.0
4.0
5.0
0 200 400 600 800 1000
L=357mH
L=268mH
L=536mH
Vin = 400 V
Vout = 300 V
Maximum value of transferred power
Fig. 4. RMS value of inductor current. At the light load, RMS
value is reduced in proportion of the inductance. The larger
inductance leads to the smaller transferred power. Therefore, the
inductor value is changed by switching the auxiliary inductors.
turn-off sequence at the charge operation, whereas Fig. 5 (c) and (d) show that at the discharge operation. In Fig. 5 (a) and (c), there is only one mode at the turn on of the bidirectional switch because the offset current is smaller than at the switching timing of S1 and S3. In Fig. 5 (b) and (d), there are four modes in the turn off of the auxiliary switches. In the Mode 1 of the charge operation, SL1 and SL2 are on at the input-side carrier peak, whereas SL1 and SL2 are on at the output-side carrier peak. In the mode 1 of Fig. 5 (b), SL1 and SL2 are on.
IV. EXPERIMENTAL RESULTS
A. Experimental conditions
A 1.0-kW prototype is tested in order to evaluate the proposed buck-boost converter with the switched auxiliary inductor. Table I shows the experimental parameters. At the both legs, IRFP460 (VISHAY) is selected. IRFP460 (VISHAY) has the on-state resistance Ron = 270 mΩ, the voltage rating Vrate = 500 V and the current rating Irate = 20 A. Note that the minimum current for ZVS I0 is calculated with about two times margin.
B. Steady state operation
Figure 6 shows the operation waveforms at the charge operation. It should be noted that the experimental conditions are the input voltage of 300 V and the output voltage of 350 V. Fig. 6 (a) shows that with auxiliary inductor, whereas Fig. 6 (b) shows that without auxiliary inductor. By applying the modulation for the ZVS achievement, the inductor current at the switching timing is also larger than the minimum current I0 at light load, i.e. ZVS achievement. In Fig. 6 (b), the minimum current for ZVS is reduced by the large inductance. In Fig. 6, the RMS value with the auxiliary inductor is reduced by 23.8% compared to that without the auxiliary inductor at the same load.
Figure 7 shows the operation waveforms at the discharge operation. Fig. 7 (a) shows that with the auxiliary inductor, whereas Fig. 7 (b) shows that without the auxiliary inductor. It should be noted that the experimental conditions are the input voltage of 300 V and the output voltage of 350 V. In Fig. 7, the current direction is difference from Fig. 5, i.e. the changing the power flow. By applying the modulation for the ZVS achievement, the inductor current at the switching timing is also larger than the minimum current I0 at the light load. By applying the switching auxiliary inductor, the inductor current is reduced by 23.8%. In Fig. 7 (b), the minimum current for ZVS I0 is also reduced by the large inductor value. The RMS value with the auxiliary inductor is reduced by 18.2%.
C. Transient Response at Switching Axiliary Iductor
Figure 8 shows the transient waveforms of switching auxiliary inductor at the discharge operation. In Fig. 8(a), the auxiliary inductor current is flown after the turn on of the auxiliary switch. In addition, it is confirmed that the DC offset of the auxiliary-inductor current is only the offset current for ZVS I0. In Fig. 8(b), the auxiliary inductor current has been flown after the turn off of SL1. Then, SL2 is turned-off when the auxiliary-inductor current becomes zero. Thus, the surge voltage does not occur because the body diode of SL1 is turned off naturally. Therefore, the low voltage rating device can be selected by applying the proposed switching sequence.
Figure 9 shows the transient waveforms of switching the auxiliary inductor at the discharge operation. Fig. 9 (a) shows
Table II Experimental conditions
Element Symbol Value
DC-bus voltage Vbus 400 V, 300 V
Battery voltage Vbat 300 V, 350 V
Dead time at HV side Td 1 ms
Main inductor L 532 mH
MOSFET
Rated power Prated 1.0 kW
Auxiliary inductor L1 532 mH
Swiching frequency fsw 50 kHz
Minimum current with L1 I0
1.7 AMinimum current w/o L1 I0
1.5 A
IRFP460 (VISHAY)
Vrate=500 V, Irate=20 A, Ron=0.27 W
0
0
5 ms/div
4.3 A
3.0 A
iL 2 A/div
iL 2 A/div
iLRMS=2.1 A iLRMS=1.6 A
v1
100 V/div
v2
100 V/divv1
100 V/div
v2
100 V/div
Reduced by 23.8%
(a) With Laux (b) Without Laux
Fig. 6. Operation waveforms at charge operation. Without the
auxiliary inductor, RMS value of the inductor current is reduced
by 28.6% compared with the auxiliary inductor.
0
0
5 ms/div-4.6 A
-3.3 A
v2
100 V/divv1
100 V/div
iL 2 A/div
v2
100 V/divv1
100 V/div
iL 2 A/div
iLRMS=2.2 A iLRMS=1.8 AReduced by 18.2%
(a) With Laux (b) Without Laux
Fig. 7. Operation waveforms at discharge charge operation. In
Fig. 7, the discharge operation is achieved because the direction
of the inductor current is changed. Without the auxiliary
inductor, RMS value of the inductor current is reduced by 28.6%
compared with the auxiliary inductor.
that at the turn on of the bidirectional switch, whereas Fig. 9(b) shows that at the turn off of the bidirectional switch. In Fig. 9(a) and (b), it is also confirmed that the DC offset of the auxiliary-inductor current is only the offset current for ZVS I0. In addition, the auxiliary inductor current has been flown after the turn off of SL1. Then, SL2 is turned-off when the bidirectional inductor current becomes zero. Thus, the surge voltage does not occur because the body diode of SL1 is turned off naturally.
D. ZVS operation
Figure 10 shows the operation waveforms of the gate signal and the drain-source voltage. Fig. 10 (a) shows the operation waveforms of S1, whereas Fig. 10 (b) shows that of S4. In Fig. 8, ZVS is achieved because the main switch is turned on at zero voltage. In addition, the surge voltage does not occur due to no recovery current at the both legs. In addition, the turn off loss can be reduced by connecting a snubber capacitor in parallel to the main switches.
E. Efficiency characteristics
Figure 11 shows the efficiency characteristics with/without the auxiliary inductor at the bidirectional operation. Fig. 11 (a) shows the efficiency characteristics at the input voltage of 400 V and the output voltage of 300 V, whereas Fig. 11 (b) shows the efficiency characteristics at the input voltage of 300 V and the output voltage of 350 V. In Fig. 11, the converter efficiency at the light-load range is improved by the large inductance. In other words, the converter loss is
reduced by up to 41%. In addition, the maximum efficiency of 98.7% is achieved as shown in Fig. 11 (a) of the light load. By changing the auxiliary inductor depending on the transferred power, the high power capability is obtained. At the rated power, the converter efficiency is 98.3% at the rated power. Moreover, the high efficiency over the wide load range is achieved.
F. Load step response
Figure 12 shows the transient waveforms at the step-up load and the step-down load. Fig. 12 (a) shows the step-up load from 300 W to 500 W, whereas Fig. 12 (b) shows the step-down load from 500 W to 300 W. In Fig. 12, the stable current response is confirmed. In addition, the minimum current for ZVS is still achieved the both step-up and the step-down load, i.e. achievement of ZVS at the transient response of the step-up and the step-down load.
V. CONCLUSION
This paper proposed the four-switch-buck-boost converter with the switched auxiliary inductor in order to improve the light load efficiency and achieve the high power capacity. In the modulation method of the buck-boost converter, the inductor current including the offset current was applied in order to achieve ZVS. In the proposed circuit, the auxiliary inductors were switched in accordance with the transferred power and the voltage conditions. By switching the auxiliary inductor, the converter losses at light load was reduced. In addition, the switching sequence for auxiliary inductor was proposed. In the experiment, the validity of the proposed
0
0
2 [ms/div]
iL 2 A/div
Dorain source voltage
100 V/div
Gate source voltage
10 V/div
ZVS
(a) S1
0
0
2 [ms/div]iL 2 A/div
Dorain source voltage
100 V/div
Gate source voltage
10 V/div
ZVS
(b) S4
Fig. 10. Operation waveforms of gate signal and drain-source
voltage. In Fig. 10 (a) and (b), ZVS is achieved by the offset
current for ZVS. In addition, the turn-off loss can be reduced by
connecting a snubber capacitor in parallel to the main switches.
Auxiliary inductor
current 2 A/divAuxiliary inductor
current 1 A/div
5 ms 5 ms
Gate signal of SL1
Gate signal of SL2
0
0
0
(a) Turn on of Laux (b) Turn off of Laux
Fig. 8. Transient waveforms of switching auxiliary inductor
Laux at charge operation. In Fig. 8 (a), the DC offset current is
only the offset current for ZVS I0.
Auxiliary inductor
current 1 A/div
0
0
0
Auxiliary inductor
current 2 A/div
5 ms
Gate signal of SL1
Gate signal of SL2
(a) Turn off of Laux (b) Turn off of Laux
Fig. 9. Transient waveforms of switching auxiliary inductor
Laux at discharge operation. In Fig. 9 (a), the DC offset current
is only the offset current for ZVS I0.
method was confirmed by a 1.0-kW prototype. As results, RMS value of the inductor current was reduced by up to 23.8%. In other words, the converter loss at the light load was reduced by up to 41 % compared to that with the auxiliary inductor. Therefore, the high efficiency over wide load was achieved by switching the auxiliary inductor. In addition, the auxiliary inductor is smoothly changed by the proposed switching sequence. In future work, the design method for the auxiliary inductor will be considered.
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Eff
icie
ncy
[%
]
96.0
96.5
97.0
97.5
98.0
98.5
99.0
Input power [p.u.]
-1.0 -0.5 0.0 0.5 1.0
Loss
-31%
With auxiliary
inductor
Without auxiliary
inductor
1p.u. 1 kW
With auxiliary
inductor
Without auxiliary
inductor
Input power [p.u.]
-1.0 -0.5 0.0 0.5 1.0
Eff
icie
ncy
[%
]
96.0
96.5
97.0
97.5
98.0
98.5
99.0
Loss
-41%
1p.u. 1 kW
(a) Vin=400V, Vout=300V (b) Vin=300V, Vout=350V
Fig. 11. Efficiency characteristics with/without auxiliary inductor. By applying the large inductor value, the converter efficiency at the
light load is improved. In addition, the high power capability is obtained when the auxiliary inductor is active.
iL 5 A/div
5 [ms/div]
Input power 100 W/div
300 W500 W
VLin 100V/div 20 [ms/div]
iL 2A/div
VLout 100V/div
300 W500 W
iL 5 A/div
5 [ms/div]
Input power 100 W/div
300 W500 W
VLin 100V/div 20 [ms/div]
iL 2A/div
VLout 100V/div
500 W300 W
0
0
0
0
(a) Reference power change: 500 W to 300 W (b) Reference power change: 300 W to 500 W
Fig. 12. Transient waveforms at step-up load and step-down load. In Fig. 12, the inductor current is seamlessly changed at the step-up and
the step-down load. In addition, the minimum current for ZVS is kept both the step-up and the step-down load.