lecture #2 devices, mixers and modulators...is much higher than audio frequencies * selectivity is...

1

Radio Frequency EngineeringLecture #2 Devices, Mixers and Modulators

Lecture #2

Devices, Mixers and ModulatorsThe main objective of this lecture is to introduce a range of basic semiconductor devices and to give examples of how they can be

employed in various analogue signal processing building blocks. The technological limitations of simple implementations and more complex

solutions will be discussed

Stepan Lucyszynステファン・ルシズィンインペリアル・カレッジ・ロンドン准教授

solutions will be discussed.


OVERVIEW DevicesMixers and Modulators

Frequency Translation Variable Phase Shifters Group Delay Synthesiser Variable Attenuators

Direct-carrier Signal ProcessingV t M d l t


Vector Modulators

Radio Frequency EngineeringLecture #2 Devices, Mixers and Modulators Devices



Silicon versus Gallium Arsenide

Silicon is much cheaper!

Silicon afers ha e increased in dia eter fro 100 (4”) Silicon wafers have increased in diameter from 100 mm (4”) to 150 mm (6”) to 200 mm (8”) to 300 mm (12”).

Because of the compound nature of GaAs semiconductors it is not possible to make such large wafers.

GaAs wafer diameter increases from 50 mm (2”) to 75 mm (3”) to 150 mm (6”)


More devices per wafer run and higher yield per run

For example, if the wafer diameter doubles, the area is increased by a factor of 4 and, therefore, cost is reduced by ~ factor of 4

2


More compatible with Si3N4 passivation layer

Lower risk of charge-carrier traps g pand improved contamination protection and more scratch resistant!

Compatible with digital VLSI CMOS gate array logic

Improved thermal conductivity (factor of ~ 3) and, therefore, better thermal management

Lower band-gap energy and, therefore, lower DC voltage supply required


LDMOS™ FETs are now commercially available with unprecedented output power levels


Advantages of GaAs over Silicon (consider a MESFET)

higher bulk resistivities in undoped substrates

Gallium Arsenide versus Silicon

higher bulk resistivities in undoped substrates Passive components can have lower losses and less parasitic capacitances Lower power dissipation and lower noise figure performance

higher low-field electron mobility in the channel, μe (by a factor of ~6) Higher conduction current densities (since Jc µe)

higher carrier (electron) saturated drift velocity, vSAT, with GaAs (InGaAs is better)


Tg

SATT and

Lgv

1

higher cut-off frequency and faster switching speeds:

GaAs FETs have a higher vSAT, compared to silicon FETs, therefore, higher fT and fmax.


Cross-Section of a Recessed-Gate Depletion-ModeGaAs Metal-Semiconductor Field Effect Transistor (MESFET)( )

SOURCE DRAIN

GATE

n+ contact layer

n channel

n- buffer layer

+ + + + ++

+ +++ +

d GaAsActiveLayer< 0.5 m

AuGe/Ni/AuAlloyedOhmicContact

Ti/Pt/AuSchottkyContact

Lg


GaAs Semi-Insulating Substrate (SIS)


Dual-gate FET cross-section and representation as a cascodeSource DrainGate 1 Gate 2

n+n

Gate 2Gate 1

Drain

n

Semi-insulating substrate


Applications include mixers, modulators and variable gain control circuits

3


High Electron Mobility Transistors (HEMTs)

The HEMT has a similar design to the MESFET, however, the intrinsic physics and electrical characteristics are dramatically different

A heterojunction is a junction between 2 semiconductors with different band gap energies



50

gm [mS]

Wg

PLAN-VIEWof a 4-Finger FET

4000 Wg [μm]

Cgs [pF]

1.2 2.0

1.5

Gate Length, Lg [μm]

of a 4-Finger FET

GATE

Lg

DRAIN

MESFET Peripheryis 4 Fingers x Wg


0 400

1.0

0.5

Wg [μm]

SOURCE

Lg


Gate Drain

Source

2X100

Gate Drain

Source

4X75


A scaleable model is one which has input parameters (e.g. number of fingers, gate width) which are used to scale the model values


TransistOr PArameter Scalable (TOPAS) Nonlinear Model


4


Gate

Lg

RgCdg

Rd Drain

Cds

Cgs

Linear FET Model

• Only valid for small-signal design

• Bias dependant and scaleable

Cds

gm RdsRI

Rs

Ls


SourceLg 0.02 nH gm 40ms Cgs 0.3 pF Rds 350 Rg 2 Cds 0.06 pF RI 8 Rs 2

Cdg 0.04 pF Ls 0.05 nH Rd 2 Vds 3 V Vgs -0.5 V Ids 10 mA


Resistive or Cold-FET:

(a) equivalent circuit model (b) variation of Gds with Vgs

Cgs Gds(Vg)+Vg-

CgdG D

Rg

Rs

Rd


(b)(a)

S

Rs


Interdigitated Planar Schottky Varactor Diode (IPSVD) from GaAs MESFETs



Simple Schottky diode equivalent circuit model

I

Cj(V) Rj(V)+V-

I


RSF

5


C3

Equivalent Circuit Model for Interdigitated Planar Schottky Varactor Diodes

Cj C2

RjLgRs

C1

FrequencyDispersionModel

DelayModel



Characterization of the 6 x 150 m IPSVD



Interdigitated Planar Schottky Diodes

nkTRIVq

S

SF

eIICurrentDiode

chargeelectronqpotentialbiasforwardV

1, 6 x 150 m diode

2 x 60 m diodes are

nkT

RIVq

SS

SF

eqInkT

IIqnkT

VIV

/)(R ,Resistance LeakageJunction

diodeofetemperaturabsoluteTconstantBoltzmannsk

resistanceseriesbiasforwardRgq

1

j

SF

6 x 150 m diode

1 x 150 m diode

commonplace


6 x 150 m diode


Frequency translation by ideal multiplication of two sinusoids

Mixers and ModulatorsFrequency Translation

0.5 cos(RFt - LOt) + 0.5 cos(RFt + LOt)

cos(RF t)

cos(LOt)

LO

Input Output

Mixer


6


A simple single-ended mixer

FET)-Cold a of econductanc channel (e.g.function nsfer linear tra][FET)-Hot a of nductanceor transco diodeSchottky a of resistance leakagejunction (e.g.function transfer nonlinear ...][

)]([)()(2

bVaVgcVbVaVg

tvgtvti LORF

signal VRF cosRFt

LO VLO cosLOt

RS non-linear device

RLI


bias Vb


pump harmonics

er

Spectrum of frequencies of the form i RF j LO

IF RF

LO

Imag

e

SumSp

ectr

al p

owe


LO

-R

F

RF

LO

LO

-R

F

LO+ R

F

LO

LO

-R

F

LO

+ R

F

LO

LO

-R

F



The 1 dB compression point is an indication of a mixer’s INPUT power dynamic range and the maximum power capabilities


Homodyne (or Direct Conversion or Zero IF) Receiver

* Image rejection filter not required in theory


* Selectivity is controlled by a simple low-pass filter* Gain is split between the RF and baseband amplifiers* Since fIF = 0, there is no image frequency to deal with* High stability of the LO can be a problem

7


RF

Homodyne Approach

Thi i k f DSB SC d BPSK i l b th h USB LSB

Baseband

LO (=RF)

LO DC


This is okay for DSB-SC and BPSK signals, because they have USB = LSB. However, if the USB and LSB contain different information

(e.g. QPSK and QAM) then quadrature conversion is required


Single-Conversion Superheterodyne Receiver

* Image rejection filter should also be included !!! (otherwise the +RF that is translated down into the +IF will be combined with any Image that is translated up to the +IF)


down into the +IF will be combined with any –Image that is translated up to the +IF)* Super-sonic Heterodyne (or Superhet for short) refers to the fact that the IF frequency is much higher than audio frequencies* Selectivity is improved by sharper IF band-pass filters* Increased complexity and, therefore, cost


Single-ended diode mixer Vb

LO IFInput

MatchingLO

FilterOutput

Matching

RFFilter


RF


I

Approximation of pulse duty ratio

V

dcV

tV

kVpWhere:S =

cos-1 Vt VdckVp


Where:S = Pulse duty ratio Vt Turn on voltage Vdc DC operating point

-cos-1 dVdcdkVp

8


10

11

12

RF = 94 GHz P 93 GH

Conversion loss against pump power for different bias voltages

3

4

5

6

7

8

9C

onve

rsio

n Lo

ss (

dB)

0

0.20.30.4

0.5

0.6

DC Bias Voltage

Pump = 93 GHz


0

1

2

LO Power (dBm)

-8 -7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 7 8 9 10


9

10

11

-3.9

R F = 94 G Hz

Pum p = 93 GH z

Conversion loss against DC junction voltage for different pump power levels

3

4

5

6

7

8

Con

vers

ion

Loss

(dB

)

98dBm

(0.4V)

= M in im a

2.04dBm

(0.8V)

5.56dBm (1.2V)

-0.46dBm

(0.6V) 3.98dBm (1.0V)


0

1

2

DC Junction Voltage V dc

-0 .3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 0.5 0.6


0 4

0.5

0.6

0.7

0.8

rsio

n Lo

ss

Vdc

xS opt

1

cos 1 1. 05

1. 2

0 .1609

Optimum DC junction voltage against pump voltage

-0.5

-0.4

-0.3

-0.2

-0.1

0

0.1

0.2

0.3

0.4

C J

unct

ion

Vol

tage

For

Min

imum

Con

ve

x

x

x

x

x

V tk

0 . 65


-0.8

-0.7

-0.6

0.5

0

0.2

0.4

0.6

0.8 1

1.2

1.4

1.6

1.8 2

Pump Voltage ( V p )

Opt

imum

DC

x

x


8

9

10Conversion loss against pulse duty ratio

2

3

4

5

6

7

Con

vers

ion

Loss

(dB

)

Pulse Duty Ratio (S)

0.1609

0.35

4.0

5.1

At P = 2.04 dBmp

RF = 94 GHz Pump = 93 GHz

-0.46dBm2.04dBm3.98dBm5.56dBm

Conversion Loss (dB)


0

1

2

0 0.1

0.2

0.3

0.4

0.5

0.6

0.7

Pulse Duty Ratio (S)

S=0.

35

S

= 0.

1609

opt

9


Microstrip implementation of a single-ended mixerVb O/C Shorts LO/RF’s

2nd Harmonic

IFO/PLO/RF

I/P

g/4

g/8

g/4Shorts the IF

and gives DC Shorts the


O/CS/C

g/4g/4and gives DC return

Shorts the LO/RF


Single-balanced mixers (SBMs) using 180° hybrids (1)180 Hybrid V 1

00

0180

RF

LO

IF

+ -

I IF= I 1- I 2I 2

I 1

+ -

RF1= 0 LO1 = 0

RF2= 0 LO2 =180

V 2


SBMs can remove ½ of the unwanted spectral components generated by single-ended mixers


Single-balanced mixers using using 180° Hybrids (2)

I IF= I 1- I 2

00

0180

LO

RF

IF

180 Hybrid+ -

I 2

I1

+ -

V1

V 2

RF1= 0LO1= 0

RF2= 180LO2= 0


V 2


Single-balanced mixers (SBMs) using 90° hybrids

900

090

RF

LO

IF

90 Hybrid+ -

I 2

I 1

+ -

I IF = I 1- I 2

V 1

V 2

RF1 = 90 LO1 = 0

RF2 = 0 LO2 = 90


V 2

10


cuit

Typical mm-wave diode mixer

RF

LO

IF

g/4

ope

n st

ub/8

ope

n st

ub

RF/

LO s

hort

circ

shor

t circ

uit

Branch-line coupler

g /4


LO

g/

g /4 shorted stubFor biasing and IF short circuit

2 X

RF/

LO

Matching

cap.

GND

bias pad


SEM of a 60 GHz GaAs MMIC pHEMT-Diode mixer



Single-balanced mixers using baluns



Double-balanced mixers (DBMs) as a combination of two single-balanced mixers (SBMs)

SBM balunbalun

0

180LO IF

LO 0

LO 180

RF0

180

SBM

RF 0

RF 180


SBM

DBMs can remove ¾ of the unwanted spectral components generated by single-ended mixers

11


Double-balanced ring mixer derived from the SBM pair

IFbalun

0

180LO

LO 0

LO 180

+IF

-IF

-IF

+IF

RF 0 RF 180 °

balun

balun


0 180

RF

balun


Phase relationships in double-balanced diode mixersV 1

+ -I 1 RF1 = 0

O1 = 0

IF

I IF1 = I 1- I 2+I 3-I 4

I 2

+ -

V 3+ -

I 3

V 1

LO1 = 0

RF2 = 0 LO2 =180

RF3 = 180 LO3 = 180


I 4

+ -V 4

RF4 = 180 LO4 = 0


Single-ended Resistive FET mixerIF

LO

IF

RF

Vg



IF

Single-balanced resistive FET mixers (1)

LOBalun

IFBalun

RFLO


12


Single-balanced resistive FET mixers (2)IF

RFLO

RFBalun

IFBalun



Single-balanced resistive FET mixers (3)IF

LO

90Hybrid

90Hybrid

RFLO

5050



Single-ended Resistive FET mixer for direct conversion to baseband

RF

Low PassFilt

RFMatch

LOMatch LO

VgLf cancels Cdg at the LO frequency


Baseband

Filter


Microphotograph of the single-ended resistive pHEMT mixer for direct conversion to baseband (1 x 1 mm2)

RF

LO


Baseband

13


Microphotograph of single-balanced resistive pHEMT mixer for direct conversion to baseband (1.3 x 1.6 mm2)

Baseband


LORF


Single-ended mixer

Single-balanced mixer

Measured performance of the direct-conversion resistive mixers

10 1 dB Conversionloss

8.5 0.5 dB

30 dB LO-RF isolation 32 dB

-6 dB LOreturn loss

-10 dB

-5 dB RFreturn loss

-15 dB


return loss

25 dBm IP2 28 dBm

28 dBm IP3 30 dBm


Double-balanced Cold-FET mixer

LO Balun

LODouble-balanced Hot-FET

Gilbert Cell Mixer

RFSS

D

D

RF B

alun


IF

IF Balun


A millimetre-wave double-balanced resistive pHEMT ring mixer (3 x 3.2 mm2)


14


Hartley Image-rejection mixer (IRM) block diagram DBM

RF

USB

90Hybrid

LSB

90Splitter 0

Splitter

0

-90

0

-90


LODBM


MDB207 HBT-Diode IRM with star mixers



Single-sideband modulator block diagram

0

LO

RFin0

Combiner90

Splitter 90Splitter RFout

(LSB)-90

0

-90


LO(Modulating signal)


Sub-harmonically-pumped resistive FET mixerIF

RF

IFFilter

RFFilter

LOBalun LO


15


Microphotograph of a millimetre-wave sub-harmonically-pumped resistive pHEMT mixer (1.1 x 2.1mm2)



A phase shifter is a control device found in many microwave communication, radar,sensor and measurement systems

Variable Phase Shifters

MMIC were developed so that phase shifters could be miniaturised for phased antenna array applications (MiMiC Programme in the US)

In principle, sub-1 dB worst-case losses would relax both transmitter power amplifier and receiver low noise amplifier specifications

For phased-array applications, low DC control power and repeatable batch processing is important


important

In order to understand the subtle differences between the two main generic types of phase shifters, the true phase shifter and true delay line must be defined


Phase Array Antennas

Adaptive phased-array antenna is probably the single most important MMIC application

fo

odcSin

ooSteeringTime

fodcSinoSteeringPhase

1:

1:

Since a phased-array antenna could have thousands of transmitter-receiver modules,MMICs are essential because they give excellent reproducibility with small size and mass

TWT Antenna Feeds

Phase Shifter

Attenuator

Phase Shifter

Attenuator

Phase Shift

Attenuator

SSPA

SSPA

SSPAAntenna Feeds


Fixed Beam-Forming Network (usually waveguide "plumbing")

ShifterAttenuator

Phase Shifter

Attenuator

SSPA

Adaptive Beam-Forming Network


Differential-Phase

)(21S

Group Delay


16


True Phase Shifters

Defined as a control device that has a flat group delay frequency response within its defined bandwidth of operation; the level of which does not change as the insertion phase is varied. Its characteristic features are:

(1) a flat relative phase shift frequency response, at all levels of relative phase shift

(2) constant group delay, resulting in no change in the timing of an input RF pulse envelope

true phase shifters can be employed in multiple space diversity receiver-combiners for aligning RF signals within a pulse envelope without changing the timing of the pulse edges


they should not be employed in wideband beam-forming networks for large aperture phased-array antennas, in order to avoid the effects of ‘phase squinting’ and ‘pulse stretching’


Frequency characteristics of a true phase shifter: (a) Insertion phase; (b) Relative phase shift; (c) Group delay



True Delay Lines

Defined as a control device that has a flat group delay frequency response, with a level that changes as its insertion phase varies, within a defined bandwidth of operation:

(1) linear relative phase shift frequency response, with a gradient that changes as the valueof relative phase shift varies

(2) flat group delay frequency response, with a level that varies, resulting in a change in thetiming of an input RF pulse envelope

true delay lines find many applications in general wideband microwave signal processing


true delay lines find many applications in general wideband microwave signal processingapplications, including beam-forming networks for large aperture phased-array antennas

When component non-idealities are included, an ideal true phase shifter circuit mayexhibit frequency characteristics of a true delay line, and vice versa


A special case of a time shifter is a true delay line, which is definedby the following expression:

S21()

frequency.angular at shift phase relative)(21

delay, group inshift relative

S


q yg

17


Frequency characteristics of a true delay line: (a) Insertion phase; (b) Relative phase shift; (c) Group delay



Advantages of Analogue Modulators

High quality switches are not required

Free from clock related harmonics

No control power with reflection-topologies (having Schottky junction devices)

Can have compact circuit


Completely multifunctional and adaptive


Single-stage Reflection-type Phase Shifters



Microphotograph of the Optimum Design Analogue Phase Shifter


18


Frequency response of the optimum design MMIC phase shifter:(a) relative phase shift and (b) group delay



Microphotograph of the Analogue Delay Line


GHzofunitshavingfrequency,centrewhere10

and,pF0.3,nH0.4

of

ZoTR

ofmaxC

ofL


Frequency responses of the MMIC delay line: (a) relative phase shift and (b) group delay



Single-stage Reflection-type Group Delay Synthesiser

Diff ti l Ph G D l

Group Delay Synthesiser

Differential-Phase Group Delay:


19




Single-stage Bi-phase Reflection-type Attenuator



Balanced analogue attenuator employing PIN diodes



Small-Shift Frequency Translator (SSFT)

Mixers cannot perform small-shift frequency translations near carrier frequency

Direct-carrier Signal Processing

Mixers cannot perform small shift frequency translations near carrier frequency A serrodyne frequency translator employs a 360o phase shifter and a sawtooth generator

Linear 360o

Phase Shifterfreq. freq.

cosotInput spectrum Output spectrum

fo (fo+ 1/T)fo


Time

Control voltage

T0

20


SSFT Applications

* Velocity deception doppler radar jammers* Homodyne vector network analyzer* Frequency scanning antenna* Chirp signal processing* Frequency-hopping CDMA* Homodyne FDMA



Velocity Deception Electronic Counter MeasuresThe velocity gate stealer (VGS) is employed against enemy Doppler radars that use a speed gate and velocity tracker. The VGS generates a slowly-changing false Doppler return (Jamming Output Power = +23 dBm at the 1dB compression point), pulling the velocity tracker off the target and then dropping it (known as velocity gate walk-off, VGWO). The radar may then lock onto clutter or be forced into a reacquisition sequence.

Madni and Wan

(Systron Donner Corp.),

Microwave System News,

Oct. 1983



Frequency Hopping CDMA

LNAFixed

frequencytransmitter

Frequencytranslator

Frequencytranslator

Fixedfrequencyreceiver

Sync.detector

Pseudo-randomsequencegenerator

Pseudo-randomsequencegenerator

TRANSMITTER RECEIVER

PADATA DATALNA


generator generator


Homodyne Frequency Division Multiplexing (FDM)

MIXER/FILTER

FREQUENCYTRANSLATOR1 FILTER

MIXER/FILTER

MIXER/FILTER

TRANSLATOR1

FREQUENCYTRANSLATOR

i

FREQUENCYTRANSLATORN

i

1

POWERCOMBINER

OUTPUTSIGNAL

INPU

T SI

GN

AL

S


FILTER TRANSLATORN

N

MASTERCARRIER

OSCILLATOR

21


Multi-Functional Signal Processing

Simultaneous multifunctional operation:

Vector Modulators

multi-level digital modulation spectral filtering, using pulse shaping small-shift frequency translation power amplifier pre-distortion linearisation

The first three functions have already been demonstrated simultaneously


Microwave vector modulators have the potential of providing high performance & low-cost solutions for next generation transmitters


Integrated Homodyne Transmitter

Phase Shifteror

Vector ModulatorOUTPUT

Vector Modulator

DAC (analogue)or

Decoder (digital)

PROMLook-up

Table

CounterClock

CARRIERGENERATOR

Clk


CounterGenerator

DigitalDecoderSPCDATA

C


Pre-DistortedConstellation

CarrierOscillator

DSP/DAC

RFOutput

PAVector

Modulator

Without Pre-Filtered

Data

Transmitted Spectrum

SSFT WithPre-Filtered Data

BasebandPredistortionLinearisation


WithPre-Filtered

Data

DataInput


RF Pre-Distortion LinearisationVector modulator can give gain expansion

GAIN GAIN

Pin Pin GAIN


Pin

22


Adaptive Baseband Pre-Distortion Linearisation“Digital Predistortion” or “Cartesian Feedback”

BASEBAND INPUT

OUTPUTHPA COUPLER

MODULATOR UPCONVERTER


DEMODULATOR

I, Q


Topology Tree for All Possible Vector Modulators Implementations



1-Channel Vector Modulators


mono-phase amplitude control


2-Channel or I-Q Vector Modulators


23








Minimum Theoretical Insertion Loss for Vector Modulators



28 GHz 4-Channel Vector Modulator with Nortel

GaInP-HBTVariable (-12 to +4 dB) MMICGain Stage for Feedforward


Gain Stage for FeedforwardPower Amplifier Lineariser

24


M/A-COM MAMDCC0002 PIN diode I-Q Vector Modulator1.805-1.88 GHz and 1.93-1.99 GHz

IIP3 +41 dBm over 10 dB control range




Employing Gilbert Cell Double-balanced Mixers




Basic bi-phase reflection-type modulator

3 dB quadrature coupler1 4

6 k 6 k

2x60 m

23-90º-90º0º 0º

V = 0.3 V

V = -0.9 V

V = -2.0 V

inherent insertion loss


V

pHEMT

S21 = j

2f x 60 m Cold-pHEMT reflection termination:

S11 versus bias voltage at 60 GHz

25


T(V1)

OutputInput

T(V1)Vector sum Image by 180 phase shift

(two couplers)

OFF

I

I

T(V2)

T(V2)

Vector sum

( p )

ON ON

OFF


Push-pull bi-phase amplitude modulator Push-pull operation of the bi-phase amplitude modulator


Bi-phase amplitude modulator using Cold-HBT reflection terminations: microphotograph and static constellation calibrated at 38 GHz

S21REF 1.0 Units200.0 mUnits /

Calibrated bias points (V)S S'0.0, 3.50.54, 2.3

SS'

0 5 , 30.78, 1.951.05, 1.831.18, 1.641.24, 1.511.25, 1.391.27, 1.301.19, 0.991.15, 0.731.13, 0.431.24, 0.771.32, 0.881.36, 0.801.40, 0.731.50, 0.651.63, 0.571 75 0 13


C.W. 38.000000000 GHz

1.75, 0.132.14, 0.03.0, 0.05.35, 0.0


I-Q (Push-Pull) Vector Modulator Principles

6 k 6 k push-pull bi-phase analogue reflection-type attenuator

Inputquadraturecoupler

6 k 6 k50 50

I

I

50 6 k 6 k

100

Outputin-phasecombiner push-pull bi-phase

modulator


6 k 6 k50 50

Q

Q

full vector modulator


I-Q (Push-Pull) Vector Modulator


26


Simultaneous Multifunctionality:• Digital modulation


Microphotograph of 38 GHz I-Q (push-pull) vector modulator

• Digital modulation (QPSK with 20 Kbit/s PRBS at 60 GHz)

• Spectral filtering using pulse shaping(square-root raised-cosine digital filter)

• +8 KHz small-shift frequency translation


W-Band Vector Modulator for Software Radar

76.5 GHz vector modulator

microstrip on 100 m GaAs

0.25 m AlGaAs/InGaAs pHEMTs (2f x 60 m)

fabricated on the Marconi Caswell Ltd. H40 MMIC

1.6

mm


H40 MMIC process2 mm


Static S-parameter Measurements (using Agilent 8510XF )

swept I and Q inputs 64 QAM constellation

Extract data


p Q p 0 V to -2 V, step 0.016 V

64 QAM constellation

Minimum insertion loss 12 dB Flatness over 1 GHz ± 0.1 dB Input return loss 15 dB Output return loss 8 dB


Modulation Spectra

64-QAM

3.125 MHz

SSFT FMCW

Carrier

6 MBit/s

fclk = 1 MHz

Lowersideband

fclk = 50 MHz fclk = 10 MHz

Carrier

Lowersideband

625 kHz


16 Samples/cycleQ-channel

I-channel

27


76.5 GHz Single-Chip Radar

complete radar prototype 2.8 mm x 2 mm



110 GHz Vector Modulator

2001



Measured Static Constellations

90

135 45

50 mUnits per division 90

135 45

50 mUnits per division

180

135 45

225 315

0 180

135 45

225 315

0


225

270

315

C.W. 110 GHz

225

270

315

C.W. 110 GHz

Swept inputs Extracted 64-QAM


64-QAM Spectral Response: 60 MBit/s at 110 GHz


Baseband I-Q signals Spectral response

lecture #2 devices, mixers and modulators...is much higher than audio frequencies * selectivity is...

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