A Speed-Sensorless Startup Method of anInduction Motor Driven by a

Modular Multilevel Cascade Inverter (MMCI-DSCC)

Yuhei Okazaki, Student Member, IEEE, Makoto Hagiwara, Member, IEEE, and Hirofumi Akagi, Fellow, IEEEDepartment of Electrical and Electronic Engineering

Tokyo Institute of Technology, Tokyo, JapanE-mail: akagi@ee.titech.ac.jp

Abstract—The modular multilevel cascade inverter based ondouble-star chopper-cells (MMCI-DSCC) has been expected asone of the next-generation multilevel PWM inverters for medium-voltage motor drives. This paper has theoretical and experimentaldiscussions on a practical speed-sensorless startup method for aninduction motor driven by the MMCI-DSCC from the standstillto a middle speed. This motor drive is suitable, especially forlarge-capacity fan-/blower-like loads, the torque of which isproportional to a square of the motor mechanical speed. Unlikethe so-called “voltz-per-heltz” or “slip-frequency” controls, three-phase stator currents are based on “feedback” control, whereastheir amplitude and frequency are based on “feedforward”control. Although the motor drive has no speed sensor attachedto the motor shaft, this method makes a slow startup stable withthe help of a stator-current feedback loop. Experimental resultsobtained from a 400-V, 15-kW downscaled system verify stableoperating performance from the standstill to a middle speed of588 min−1 loaded with 60%.

Keywords— Modular multilevel cascade inverters, medium-voltage induction motor drives, speed-sensorless startup

I. INTRODUCTION

Attention has been paid to medium-voltage motor drivesfor energy savings without regenerative braking [1], [2].The modular multilevel cascade inverter based on double-starchopper-cells (MMCI-DSCC) has been expected as one of thenext-generation medium-voltage multilevel PWM inverters forsuch motor drives [3]–[11]. For the sake of simplicity, theMMCI-DSCC is referred to as the DSCC in this paper. Eachleg of the DSCC consists of two arms and a center-tappedinductor sitting between the two arms. Each arm consists ofmultiple bidirectional dc/dc choppers called just as chopper-cells. The low-voltage sides of the chopper-cells are connectedin cascade, while the high-voltage side of each chopper-cell isequipped with a dc capacitor and a voltage sensor. A synergyeffect of lower voltage steps and phase-shifted PWM leadsto lower harmonic voltage and current as well as lower EMIemissions as the number of cascaded chopper-cells increases.The design of the power conversion circuit is so flexible thatany number of cascaded chopper-cells is theoretically possible[4].

Although the DSCC has such advantages, the motor drivewould suffer from ac-voltage fluctuation of the dc capacitorin each chopper-cell, because the ac-voltage fluctuation gets

more serious as the motor mechanical speed gets lower [5].Several papers have addressed a startup method for the motordrive equipped with or without a speed sensor [7]–[11].

The authors of [7] have proposed a simple startup methodwithout a speed sensor, in which the DSCC was operated at anappropriate constant frequency, e.g., 30 Hz, and an appropriateconstant voltage for the purpose of reducing the ac-voltagefluctuation and getting a startup torque. A slip frequency,which is much higher than the rated one, would bring anovercurrent to both DSCC and motor in a low-speed range.Moreover, it is accompanied by producing a reduced motortorque.

Other startup method from the standstill has been discussedfor the motor drive with a speed sensor [8]–[11]. A seriousac-voltage fluctuation in a low-speed range can be mitigatedby injecting a common-mode voltage and a circulating currentinto each leg of the DSCC [10], [11]. However, it is desirable toeliminate a speed sensor from the motor drive, especially whenthe motor drive is introduced to a hostile environment [12],when a new DSCC is applied to an already-existing constant-speed motor without a speed sensor, or when a long lead cableis required to connect a new DSCC and a new motor.

The aim of this paper is to verify the effectiveness andpracticability of the DSCC-based induction motor drive with-out a speed sensor, in which the motor starts rotating fromthe standstill to a middle speed under a ramp change. Thespeed control method proposed in this paper is somewhatsimilar in basic idea to the conventional “volt-per-hertz”, orshortly “V/f” and “slip-frequency” controls, but different interms of combining these controls together. The proposedmotor drive is based on the “feedback” control of the statorcurrent as in the slip-frequency control, whereas the commandsfor the stator-current amplitude and frequency are based onthe “feedforward” control in consideration of the load-torquechange, as in the V/f control. Therefore, nether the motorparameters nor the speed sensor are required. Furthermore,the proposed startup method can be applied to any inverterequipped with current sensors in ac terminals. This paperdescribes a design consideration for the DSCC-based motordrive when the proposed speed control method is applied. Ex-perimental waveforms obtained from a downscaled inductionmotor drive rated at 400 V and 15 kW verify the stable speed-sensorless startup from the standstill to a middle speed of588 min−1 loaded with 60%.

1473978-1-4799-0336-8/13/$31.00 ©2013 IEEE

iPu

iNuiZui1u

i1v

i1w

vC1u

vC5u

vdc

vCjuC

(j : 1− 8)

u-phase v-phase w-phase

cell 1u

cell 4u

cell 5u

cell 8u

cell 1v

cell 4v

cell 5v

cell 8v

cell 1w

cell 4w

cell 5w

cell 8w

v1uvv1u

v1v

v1w

a

b

clab

(a) Power circuit.

(b) Chopper-cell. (c) Center-tapped inductor.

Fig. 1. Circuit configuration for a modular multilevel cascade inverter basedon double-star chopper-cells (MMCI-DSCC).

II. CIRCUIT AND CONTROL OF THE DSCC

A. Circuit Configuration

Fig. 1(a) shows the main circuit configuration for the DSCCdiscussed in this paper. Each leg consists of eight cascadedbidirectional chopper-cells shown in Fig. 1(b), and a center-tapped inductor per phase, as shown in Fig. 1(c). The center-tapof each inductor, named “c” is connected directly to each ofthe stator terminals of an induction motor, where i1u is theu-phase stator current. iPu and iNu are the u-phase positiveand negative arm currents, respectively. iZu is the u-phasecirculating current, defined as follows [5]:

iZu ≜ 1

2(iPu + iNu). (1)

iZu has dc- and ac-components to be used for dc-capacitorvoltage control. The dc component flows from the commondc link to each leg, while the ac component flows amongthe three legs. The individual ac components included in thethree circulating currents cancel each other out, so that no accomponent appears in either the motor current or the dc-linkcurrent.

iPu and iNu can be expressed by using i1u and iZu asfollows [5]:

iPu =i1u2

+ iZu (2)

iNu = − i1u2

+ iZu. (3)

The dc-capacitor voltage in each chopper-cell consists ofthe dc component and the ac component causing ac-voltagefluctuation. The peak-to-peak ac-voltage fluctuation, ∆vCju,is approximated as follows [7]:

∆vCju ≃√2I1

4πfC, (4)

where I1 is the rms value of the stator current, f is thefrequency of the stator current, and C is the capacitance valueof each dc capacitor. According to (4), ∆vCju is inversely pro-portional to f , and proportional to I1. Hence, ∆vCju increasesas the stator-current frequency decreases. The increase in theac-voltage fluctuation is not desirable due to the followingreasons [11]:

• It affects the voltage rating of IGBTs.

• It causes overmodulation of each chopper-cell.

• It makes the system unstable because the ac-voltagefluctuation can be considered as a disturbance to thecontrol system.

Therefore, the ac-voltage fluctuation should be mitigated to anacceptable level.

B. DC-Capacitor Voltage Control

This paper employs two kinds of existing capacitor voltagecontrols to mitigate the ac-voltage fluctuation and to regulatethe dc mean voltage of each dc capacitor. The dc mean voltageis regulated by using the dc component of the circulatingcurrent [5], while the ac-voltage fluctuation is mitigated byusing the common-mode voltage superimposed on three center-tap terminals of the DSCC and the ac components of thecirculating currents flowing inside the three legs [8]–[11]. Asa result, the ac-voltage fluctuation is independent of the time-varying frequencies of the stator current, but dependent on afixed frequency of the common-mode voltage.

Finally, this paper switches over the control method ac-cording to the stator-current frequency.

• At the low-speed range of f ≤ 12 Hz, the common-mode voltage and the ac components of the circulatingcurrents are controlled actively to mitigate the ac-voltage fluctuation of each dc-capacitor voltage.

• When f ≥ 20 Hz, the common-mode voltage and theac components of the circulating currents are set tozero.

During a transient speed range of 12 ≤ f ≤ 20 Hz, thecommon-mode voltage and the ac components of the circu-lating currents decrease linearly in their amplitudes. Note thatthe dc component of the circulating current is used to regulatethe dc mean voltage of each dc capacitor through all frequencyranges.

III. STATOR-CURRENT CONTROL

This paper proposes a speed control based on a feedbackcontrol of the stator current for achieving stable startup of aninduction motor. For the sake of simplicity, it is simply referredas the speed control in this paper. Firstly, the control principles

1474

dq dq−1

PI

PI

i1uvw v∗1uvw

θ∗

i∗d(=√

3/2I∗1 )

i∗q (=√

3/2I∗1 )

id

iq

Fig. 2. Block diagrams for the speed control based on a feedback control ofthe stator current.

R1 L1 −M2/L2

M2/L2 (M/L2)2R2/s

I0

I1

V1

(L2/M)I2

Fig. 3. Per-phase equivalent circuit based on the total linkage-flux of thesecondary windings [13].

of the speed control are discussed. Secondly, comparisonsamong the conventional speed controls, i.e., “voltz-per-heltz”and “slip-frequency” controls, are discussed.

A. Control Principles

The speed control forms a feedback control of the three-phase stator currents to realize a stable speed-sensorless startupfrom the standstill. This means that the current sensors arerequired to realize the feedback control. The DSCC has sixcurrent sensors to detect each of the arm currents, so that thestator currents can be calculated from (2) and (3) by using thedetected arm currents. This means that no additional currentsensor is required.

Fig. 2 shows the block diagrams for the stator-currentcontrol. The three-phase stator currents are transformed into dcquantities by using the d-q transformation to enhance currentcontrollability. In Fig. 2, θ∗ is the phase information usedfor the d-q transformation, while i∗d and i∗q are the currentcommands given by

i∗d = i∗q =

√3

2I∗1 , (5)

where I∗1 is the command of the stator rms current. Note thatI∗1 and f∗ are given not by feedback control but by feedforwardcontrol.

Fig. 3 shows a per-phase equivalent circuit of an inductionmotor based on the total linkage flux of the secondary windings[13]. Although this circuit is valid only under steady-stateconditions, it is applicable to a fan-/blower-like load in whichmotor speed changes slowly enough to be considered as thecontinuity of steady-state conditions. Here, I1 is the statorphasor current, I0 is the magnetizing phasor current, and I2 is

I1i

I1j

I1k

0 I2iI2jI2k

I0i

I0j

I0k

I1i<I1j<I1k

fsi>fsj>fsk

R

I

Fig. 4. Phasor diagrams for the stator currents with different amplitudes andthe same torque.

the torque phasor current. I0 and I2 are orthogonal each otherin steady-state conditions. The rms value of I1, I1 is given asfollows:

I1 =

√I20 + (

L2

MI2)2. (6)

The motor torque TM is expressed by using I0 and I2,which are the rms values of I0 and I2, respectively, as follows[13]:

TM = 3PMI0I2, (7)

where P is the pole-pair number.

Fig. 4 shows phasor diagrams for the three stator currentsunder the same torque, in which the relation of I1i < I1j < I1kholds. The imaginary flame I corresponds to the magnetizingcurrent I0, and the real flame R corresponds to the torquecurrent I2. It is obvious from (7) and Fig. 4 that the motortorque TM is proportional to the area of triangle surroundedby I1, I2, and I0.

When the stator phasor current increases from I1i to I1junder a constant torque condition, the torque current decreasesfrom I2i to I2j , while the magnetizing current increases fromI0i to I0j , respectively, to keep the area of the triangle constant.In other words, the magnetizing current and torque currentchange their amplitudes automatically, when I1 increases inamplitudes.

The slip frequency fs is described by using I2 and I0 asfollows [13]:

fs =R2

2πM

I2I0

. (8)

A relation of fsi > fsj > fsk exists in Fig. 4, which are the slipfrequencies at different operating points. The slip frequency isdetermined unambiguously when TM and I1 are given.

B. Comparison of the Three Speed Controls

A similarity exists between the speed control proposed inthis paper and the conventional “voltz-par-heltz”, or shortly“V/f” and “slip-frequency” controls.

Table I summarizes the comparison among the three controlmethods. The V/f control has two independent variables, i.e.,V1 and f , in which V1 is the stator voltage and f is the stator

1475

TABLE I. COMPARISON AMONG THE CONVENTIONAL VOLT-PER-HELTZ AND SLIP-FREQUENCY CONTROLS, AND THE SPEED CONTROL PROPOSED INTHIS PAPER.

Volt-per-heltz control Slip-frequency control Speed control proposed in this paperIndependent variables V1 and f I1 and fs I1 and fDependent variables I1 and fs V1 and f V1 and fs

Voltage control Feedforward - -Current control - Feedback FeedbackSpeed sensor No Yes No

frequency. On the other hand, the dependent variables are thestator current I1 and the slip frequency fs. The V/f controlis a straightforward speed control method requiring no speedsensor, which is based on feedforward control of V1 and f .However, the system may suffer from overcurrent during amotor startup or when a rapid change in torque occurs becauseit has no capability to achieve fast torque control.

The slip-frequency control has two independent variables,i.e., I1 and fs, and V1 and f are the dependent variables.Here, the commands for I1 and fs are determined by feedbackcontrol of the motor speed, thus requiring a speed sensorattached to the motor shaft. The slip-frequency control canachieve faster torque-response speed than the V/f controlowing to the feedback control of the motor speed.

The speed control proposed in this paper has two indepen-dent variables, i.e., I1 and f , and V1 and fs are the dependentvariables. Unlike the slip-frequency control, the speed controlrequires no speed sensor because the commands for I1 andf , i.e., I∗1 and f∗, are given not by feedback control, butby feedforward control as in the V/f control. This impliesthat the speed control proposed in this paper is inferior tothe slip-frequency control in terms of torque controllability.However, it is applicable to a fan-/blower-like load where thechange in torque is relatively slow. Moreover, no overcurrentoccurs during a motor startup owing to feedback control ofthe stator current. For these characteristics, the speed controlis considered as the combination of the V/f and slip-frequencycontrols.

IV. CURRENT COMMANDS

This section describes how to determine I∗1 and f∗, whichare commands for the stator rms current I1 and the statorfrequency f , respectively. The following two methods can beused to determine I∗1 and f∗:

• the method determining the commands by usingFig. 3,

• and the method experimentally determining the com-mands.

A. Design Considerations

The following conditions should be considered in designingI∗1 :

• I∗1 should be a minimum value to produce a desiredmotor torque TM ;

• the maximum value of the arm currents is less thanthe amplitude of the rated stator current.

The first condition should be met because the larger I∗1 is,the larger ∆vCju is, as predicted from (4). In other words,it is possible to minimize ∆vCju by minimizing I∗1 . Thesecond condition should be met because the increase in thearm currents causes an additional loss to the converter, andmaking the size and weight of the inductor larger and heavier.Note that the increase in the arm currents occurs especiallyat low-speed operation where a large amount of ac circulatingcurrent is superimposed on the arm current. The ac circulatingcurrent is superimposed to suppress the ac-voltage fluctuationin the dc-capacitor voltage of each chopper-cell. Hence, I∗1should be minimized because the maximum value of the armcurrent is proportional to I∗1 [11].

B. The Method Determining the Commands by Using Fig. 3

When the load-torque characteristics is already known, theequivalent circuit shown in Fig. 3 can be used to determine I∗1and f∗, which is characterized by using the motor parameterssuch as the moment of load inertia and the mutual inductancebetween stator and rotor windings. The motor torque shouldsatisfy the following equation for the startup:

TM − TL > (JM + JL)dωrm

dt, (9)

where TL is the load torque, JM is the moment of inertia ofthe motor, JL is that of the load, and ωrm is the mechanical-angular velocity. The right hand term on (9) corresponds to anacceleration torque for the startup.

For making analysis simple and easy, the following rea-sonable approximations are made:

• The stator-current frequency f agrees well with itsreference f∗ (i.e., f = f∗).

• The slip frequency fs is much smaller than f (i.e.,fs ≪ f ).

• The moment of inertia of the load is much larger thanthat of the motor (i.e., JM ≪ JL).

The first assumption is valid under the condition that themotor is applied to a fan-/blower-like load in which the motorfrequency, or the motor mechanical speed, is controlled slowly,e.g., spending a few or several minutes, to complete its startupprocedure. The third assumption is valid when the motor isapplied to the fan-/blower-like load where JL is typically fiftyor hundred times larger than JM .

Finally, (9) is simplified as follows:

TM − TL > JL2π

P

df∗

dt, (10)

1476

V ∗C f∗ I∗1

DSP(TMS320C6713) FPGA(Altera Cyclone II)

MUX

vC24

vdc

6

648gatesignals

MUX: Multiplexer

Regenerative load

400V/200V200V

200V/400V

IM

IG

v1uvτLNrm

Fig. 1

iPiN

Fig. 5. The 400-V 15-kW downscaled system used in experiments.

TABLE II. CIRCUIT PARAMETERS USED IN EXPERIMENTS.

Rated active power 15 kWRated line-to-line rms voltage VS 400 V

Rated dc-link voltage Vdc 560 VCenter-tapped inductor lab 4.0 mH(12%)

DC capacitor of chopper-cell C 3.3 mFDC-capacitor voltage VC 140 V

Unit capacitance constant H 52 ms [14]Cell count per leg N 8

Triangular-wave-carrier frequency fC 2 kHzEquivalent carrier frequency NfC 16 kHz

*The value in () is on a 400-V, 15-kW, and 50-Hz base.

TABLE III. MOTOR PARAMETERS USED IN EXPERIMENTS.

Rated output power 15 kWRated frequency 50 Hz

Rated line-to-line rms voltage V 380 VRated mechanical speed Nrm 1460 min−1

Rated stator rms current I1 32 APole-pair number P 2

Moment of motor inertia JM 0.1 kg・m2

Moment of load inertia JL 0.1 kg・m2

where ωrm = 2πf∗/P . Equation (10) means that the acceler-ation torque is proportional to the slope of change in f∗. Thissuggests the minimum torque required for the motor startupis TM = TL when the term on the right hand side in (10) issmall enough to be negligible. In other words, the slope of f∗

should be set as small as possible to reduce the accelerationtorque.

The motor torque TM in Fig. 3 is propotinal to the areasurrounded by I1, I2, and I0. The stator rms current I1 toproduce a required motor torque TM becomes minimal whenthe following relation is met:

I0 =L2

MI2. (11)

Substituting (11) into (7) yields

I2 =

√TML2

3PM. (12)

I1 =6.4 Arms

(20% of 32-A base)

↓

16 Apeak

(36% of 45-A base)

↖t = t0

↖t1

↖t2

29 V(21% of 140-V base)↓

↑

� -20 s600 min−1

[min−1]

Nrm

750

375

0

[V]

vuv

400

0

-400

[A]

iu

30

0

-30

[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

90

0

Fig. 6. Experimental startup waveforms when I∗1 = 6.4 A (20%)and τL = 0%.

Finally, I1 is obtained by substituting (12) into (6) as follows:

I1 =

√2L2TM

3PM2. (13)

C. The Method Experimentally Determining the Commands

The current command I∗1 should be determined experimen-tally when the load torque is unknown; the initial value of I∗1 isset to zero. Then, it is increased gradually to find a minimumvalue where the motor can start up. This method is similar tothe V/f control in that the motor parameters are unnecessarilyand that the slope of V/f is adjusted experimentally to producea desired motor torque.

It is difficult to apply the speed control method proposedin this paper to applications where the load-torque changeis steep and unpredictable, because this method is based onfeedforward control with no capability of fast torque control.However, this method is applicable to a fan-/blower-like loadwhere the torque-change speed is relatively slow, and the loadtorque of which is proportional to a square of the motormechanical speed. In this case, I1 should be given so that it isproportional to the motor mechanical speed, as predicted from(13). I1 is also proportional to the stator-current frequency f ,because the slip frequency fs is typically negligible comparedto the stator-current frequency (fs ≪ f ).

Finally, the adjustment of the slope of I1/f (= I∗1/f∗) by

experiment is required to achieve stable startup of the motor.The initial value of I1 when f = 0 should be increased whena startup torque is required. This method is similar to the so-called “torque boost” function in the V/f control.

1477

I1 =10 Arms (31% of 32-A base)

↖t = t0

34 V (24% of 140-V base)↓

↑

� -20 s591 min−1

↓23 Apeak (51% of 45-A base)

[min−1]

Nrm

750

375

0[V]

vuv

400

0

-400[A]

iu

30

0

-30

[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

90

0

Fig. 7. Experimental startup waveforms when I∗1 = 10 A (31%)and τL = 20%.

V. EXPERIMENT

A. Experimental System Configuration

Fig. 5 shows the system configuration of the 400-V 15-kWdownscaled system. Table II summarizes the circuit parametersused in experiments. Table III summarizes the specificationsof a 380-V 15-kW induction motor tested. Here, a 12-pulsediode rectifier, consisting of a three-winding transformer witha ∆−∆−Y connection and two six-pulse diode rectifiers, isused as the front end. When the supply voltage matches themotor voltage, a transformerless medium voltage motor drivecan be achieved by replacing the 12-pulse diode rectifier witha six-pulse diode rectifier. Neither an electrolytic capacitor nora film capacitor is connected to the common dc link.

The ac output terminals of the inverter are directly con-nected to an induction motor rated at 380 V and 15 kW.The regenerative load in Fig. 5 consists of an inductiongenerator rated at 190 V and 15 kW and two identical PWMconverters connected back to back. The field-oriented control isapplied to the induction generator, which enables an arbitraryinstantaneous torque τL to be loaded on the induction motor.

The control system shown in Fig. 5 detects each dc-capacitor voltage vC , both positive- and negative-arm currentsiP and iN , and a dc-link voltage. These are input signals to theA/D converters. Here, the multiplexer unit is used to reducethe number of the analog signals from 24 to six. A digitalsignal processor unit using a Texas Instrument TMS320C6713takes in the digital signals from the A/D converters andproduces the voltage command of each chopper-cell. Notethat the motor mechanical speed, Nrm is obtained from atachogenerator attached to the motor shaft, not for control but

I1 =14 Arms (44% of 32-A base)

↖t = t0

38 V (27% of 140-V base)↓

↑

� -20 s

↓34 Apeak (76% of 45-A base)

590 min−1[min−1]

Nrm

750

375

0[V]

vuv

400

0

-400[A]

iu

30

0

-30

[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

90

0

Fig. 8. Experimental startup waveforms when I∗1 = 14 A (44%),and τL = 40%.

for measurement.

The dc-capacitor voltage command was set to V ∗C = 140 V.

A square-wave common-mode voltage and square-wave circu-lating currents are used to mitigate the ac-voltage fluctuationof each dc capacitor [11], in which the rms value of thecommon-mode voltage and its frequence, Vcom and fcom, wereset to Vcom = 180 V and fcom = 50 Hz, respectively. Thecommand for the stator rms current, I∗1 , was determined bythe experiment-based method mentioned above.

B. Startup Performance

Figs. 6–9 shows the experimental startup performance withdifferent load torque. The harmonic voltages included in theline-to-line voltage of the DSCC, vuv, were cut off by usinga low-pass filter with a cut-off frequency of 400 Hz. Thecommand for the stator-current frequency, f∗, was changedfrom zero to 20 Hz under a ramp change rate of 1 Hz/s. Theacceleration torque is obtained from Table III and (10) as 0.7%of the rated torque, which is small enough to be negligible.Hence, the relation of TM ≈ TL holds.

Fig. 6 shows the experimental startup performance with noload torque. Here, I∗1 was set to 6.4 A, which is 20% of therated stator rms current of 32 A. The motor mechanical speedincreased from zero to the synchronous speed of 600 min−1

without overshoot or undershoot. The rms value of the statorcurrent i1u is regulated at 6.4 A without any steady-state errorby applying the feedback control shown in Fig. 2. The relationof I1 = I0 exists in Fig. 4 because the load torque including theacceleration torque is zero from a practical point of view. Theamplitude of vuv increased linearly as Nrm increased because

1478

I1 =17 Arms (53% of 32-A base)

↖t = t0

51 V (36% of 140-V base)↓

↑

� -20 s588 min−1

←46 Apeak

(102% of 45-A base)

[min−1]

Nrm

750

375

0[V]

vuv

400

0

-400[A]

iu

30

0

-30

[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

90

0

Fig. 9. Experimental startup waveforms when I∗1 = 17 A (53%),and τL = 60%.

I0 (= I1) is regulated at a constant value, which is predictedfrom Fig. 3.

A square-wave common-mode voltage with Vcom = 180 Vand fcom = 50 Hz and square-wave circulating currents aresuperimposed during t0 ≤ t ≤ t1 to mitigate the ac-voltagefluctuation of each dc-capacitor voltage. During t1 ≤ t ≤ t2,the amplitudes of the common-mode voltage and the accirculating currents were decreased linearly, and they were setto zero when t2 ≤ t. As a result, the amplitudes of iPu andiNu during t0 ≤ t ≤ t1 are larger than those during t2 ≤ t.However, the peak value of the arm currents are smaller thanthe amplitude of the rated stator current of 45 A (=

√2×32 A).

The maximum amplitude of the arm currents is 16 A, whichis 36% of 45 A. The dc-mean voltages of vC1u and vC5u areregulated at the command value of 140 V. The maximum ac-voltage fluctuation of vC1u and vC5u is 29 V, which is 21%of 140 V.

Fig. 7 shows the experimental startup performance withτL = 20%. I∗1 was set to 10 A (31%), which is the minimalvalue to produce a motor torque of 20%. The motor mechanicalspeed increased up to 591 min−1; hence, the slip frequency isfs = 0.30 Hz. The maximum amplitude of the arm currentsis 23 A, which is 51% of 45 A. The maximum ac-voltagefluctuation of vC1u and vC5u is 34 V, which is 24% of 140 V.

Fig. 8 shows the experimental startup performance withτL = 40%. I∗1 should be increased to

√2 times compared to

when τL = 20%, because I∗1 is proportional to a square rootof torque, according to (13). Hence, I∗1 was changed to 14 A(=

√2× 10 A, 44%). The motor mechanical speed increased

up to 590 min−1; hence, the slip frequency is fs = 0.33 Hz.The maximum amplitude of the arm currents is 34 A, whichis 76% of 45 A. The maximum ac-voltage fluctuation of vC1u

↓19 min−1

34 V (24% of 140-V base)↓

↑

� -1 s

←38 Apeak (84% of 45-A base)

[min−1]

Nrm

750

375

0[V]vuvvvwvwu

400

0

-400[A]iuiviw

30

0

-30[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

140

100

Fig. 10. Experimental steady-state waveforms when I∗1 = 17 A (53%),f∗ = 1 Hz, and τL = 60%.

and vC5u is 38 V, which is 27% of 140 V.

Fig. 9 shows the experimental startup performance withτL = 60%. I∗1 was set to 17 A (=

√3 × 10 A, 53%). The

motor mechanical speed increased up to 588 min−1; hence,the slip frequency is fs = 0.4 Hz. The maximum amplitudeof the arm currents is 46 A, which is almost 100% of 45 A.The maximum ac-voltage fluctuation of vC1u and vC5u is 51 V,which is 36% of 140 V.

C. Steady-State Performance

Figs. 10–12 show the experimental steady-state perfor-mance for different frequencies of operation. Here, I∗1 and τLwere set to I∗1 = 17 A (53%) and τL = 60%, respectively.

Fig. 10 shows the experimental steady-state performanceunder an ultralow-speed operation at f∗ = 1 Hz. Here, thecommon-mode voltage with Vcom = 180 V and fcom = 50 Hzand the square-wave circulating currents are superimposed.The motor mechanical speed and the slip frequency are ob-tained as Nrm = 19 min−1 and fs = 0.38 Hz, respectively.The maximum amplitude of the arm currents is 38 A, whichis 84% of 45 A. The ac-voltage fluctuation of vC1u and vC5u

is 34 V, which is 24% of 140 V.

Fig. 11 shows the experimental steady-state performanceat f∗ = 15 Hz. Here, Vcom was reduced to 113 V, and theamplitude of the square-wave circulating currents were reducedsimultaneously. The motor mechanical speed and the slipfrequency are obtained as Nrm = 438 min−1 and fs = 0.4 Hz,respectively. The maximum amplitude of the arm currents is31 A, which is 69% of 45 A. The ac-voltage fluctuation ofvC1u and vC5u is 43 V, which is 31% of 140 V.

1479

↓438 min−1

43 V (31% of 140-V base)↓

↑

� -67 ms

←31 Apeak (69% of 45-A base)

[min−1]

Nrm

750

375

0[V]vuvvvwvwu

400

0

-400[A]iuiviw

30

0

-30[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

140

100

Fig. 11. Experimental steady-state waveforms when I∗1 = 17 A (53%),f∗ = 15 Hz, and τL = 60%.

Fig. 12 shows the experimental steady-state performanceunder a middle-speed operation at f∗ = 20 Hz. Here, Vcom

and the amplitude of the square-wave circulating currents werereduced to zero, because the ac-voltage fluctuation of eachdc-capacitor voltage is not serious in this frequency range.Although the peak value of the arm currents can be reducedto 19 A, which is 42% of 45 A, they contain the 40-Hz second-order frequency component resulting from the control system[7]. The ac-voltage fluctuation of vC1u and vC5u is 50 V, whichis 36% of 140 V.

VI. CONCLUTION

This paper has proposed a practical startup method foran induction motor driven by the MMCI-DSCC from thestandstill to a middle speed which requires no speed sensor.This method is characterized by controlling the stator currentby feedback control; but its amplitude and frequency aregiven by feedforward control. A 400-V 15-kW downscaledsystem has shown that the motor loaded with 60% can achievea stable startup from the standstill to a middle speed ofNrm = 588 min−1 without overvoltage and overcurrent. Thismethod is especially suitable for a fan-/blower-like load toachieve energy savings.

REFERENCES

[1] P. W. Hammond, “A new approach to enhance power quality for mediumvoltage ac drives,” IEEE Trans. Ind. Appl., vol. 33, no. 1, pp. 202–208,Jan./Feb. 1997.

[2] S. Malik and D. Kluge, “ACS 1000 world’s first standard ac drive formedium-voltage applications,” ABB Review, no. 2, pp. 4–11, 1998.

[3] H. Akagi, “Classification, terminology, and application of the modularmultilevel cascade converter (MMCC),” IEEE Trans. Power Electron.,vol. 26, no. 11, pp. 3119–3130, Nov. 2011.

↓588 min−1

50 V (36% of 140-V base)↓

↑

� -50 ms

↓19 Apeak (42% of 45-A base)

[min−1]

Nrm

750

375

0[V]vuvvvwvwu

400

0

-400[A]iuiviw

30

0

-30[A]

iPu

iNu

50

0

-50[V]

vC1u

vC5u

180

140

100

Fig. 12. Experimental steady-state waveforms with I∗1 = 17 A (53%),f∗ = 20 Hz, and τL = 60%.

[4] A. Lesnicar and R .Marquardt, “An innovative modular multilevelconverter topology suitable for a wide power range,” IEEE Conf. Rec.,Bologna PowerTech 2003, CD-ROM.

[5] M. Hagiwara and H. Akagi, “Control and experiment of pulse-width-modulated modular multilevel converters,” IEEE Trans. Power Elec-tron., vol. 24, no. 7, pp. 1737–1746, Jul. 2009.

[6] M. Hiller, D. Krug, R. Sommer, and S. Rohner, “A new highly modularmedium voltage converter topology for industrial drive applications,” inConf. Rec. EPE 2009, pp. 1–10.

[7] M. Hagiwara, K. Nishimura, and H. Akagi, “A medium-voltage motordrive with a modular multilevel PWM inverter,” IEEE Trans. PowerElectron., vol. 25, no. 7, pp. 1786–1799, Jul. 2010.

[8] A. Antonopoulos, L. Angquist, S. Norrga, K. Llves, and H. P. Nee,“Modular multilevel converter ac motor drives with constant torquefrom zero to nominal speed,” Conf. Rec. IEEE-ECCE 2012, pp. 739–746.

[9] J. Kolb, F. Kammerer, and M. Braun “Dimensioning and design of amodular multilevel converter for drive applications,” in Conf. Rec. EPE2012, CD-ROM.

[10] A. J. Korn, M. Winkelnkemper, and P. Steimer, “Low output frequencyoperation of the modular multilevel converter,” in Conf. Rec. IEEE-ECCE 2010, pp. 3993–3997.

[11] M. Hagiwara, I. Hasegawa, and H. Akagi, “Startup and low-speedoperation of an adjustable-speed motor driven by a modular multilevelcascade inverter (MMCI),” Conf. Rec. IEEE-ECCE 2012, pp. 718–725,to be published in IEEE Trans. Ind. Appl.

[12] J. Holtz, “Sensorless control of induction motor drives,” Proceedingsof the IEEE, vol. 90, no. 8, pp. 1359–1394, Aug. 2002.

[13] N. Hirotami, H. Akagi, I. Takahashi, A. Nabae, “A new equivalentcircuit of induction motor based on the total linkage flux of thesecondary windings,” Electrical Engineering in Japan, vol. 103, no. 2,pp. 68–73, 1983.

[14] H. Fujita, S. Tominaga, and H. Akagi, “Analysis and design of a dcvoltage-controlled static var compensator using quad-series voltage-source inverters,” IEEE Trans. Ind. Appl., vol. 32, no. 4, pp. 970–977,Jul./Aug. 1996.

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