on electric machinery for integrated motor drives in...

On Electric Machinery for Integrated Motor Drives inAutomotive Applications

HUI ZHANG

Doctoral ThesisStockholm, Sweden 2017

TRITA-EE 2017:036ISSN 1653-5146ISBN 978-91-7729-381-1

KTH School of Electrical EngineeringSE-100 44 Stockholm

SWEDEN

Akademisk avhandling som med tillstånd av Kungl. Tekniska högskolan framläggestill offentlig granskning för avläggande av teknologie doktorsexamen onsdagen den14:e juni 2017, klockan 10.15 i sal V1, Kungl. Tekniska högskolan, Teknikringen 76,Stockholm.

© Hui Zhang, June 2017

Tryck: Universitetsservice US AB

Learning without thought is labor lost; thought without learning is perilous.

– Confucius

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Abstract

Compact, electric drives for automotive traction applications represent animportant enabler towards realizing tomorrow’s fossil free transport solutions.One attractive solution is to integrate the power electronic converter withits associated electric machinery into a single unit. This thesis, along withits appended papers, considers design and analysis of electric machinery forintegrated electric drives intended for automotive applications. Particular focusis put on permanent-magnet synchronous machines (PMSMs) with interior-mounted permanent magnets combined with modular converter topologies.

In the first part of the thesis, different converter concepts and windingarrangements suitable for an integrated drive are reviewed. Compared to theconventional solution utilizing a three-phase two-level converter, a compactintegration can be implemented by physically splitting the converter and itsassociated dc-link capacitor into a number of converter submodules. Moreover,a modular concept also enables a certain level of fault tolerance.

In the second part of the thesis, fractional-slot concentrated windings(FSCWs) are analyzed. First, a review for how to determine suitable slot, pole,and phase combinations is identified considering mainly the winding factor forthe main harmonic and the associated rotor losses. Then, integrated modularconverter concepts and associated winding configurations are considered andslot, pole and phase combinations that also comply with the consideredmodular converters are proposed. Further, two possible winding arrangementssuitable for the stacked polyphase bridges (SPB) and the parallel polyphasebridges (PPB) type converter are compared with respect to torque duringpost-fault operation in the event of failure of a single converter submodule.

In the third part, an iterative process adopting both finite element analysisand analytical techniques is proposed for the design of PMSMs with interior-mounted permanent magnets and FSCWs. The resulting machine designsillustrate tradeoffs in terms of fault tolerance, power factor, torque density,and potential for field-weakening operation. From a given set of specifications,an experimental prototype is also designed and built.

Finally, since a FSCW generally results in a large harmonic content ofthe resulting flux-density waveform, models for predicting eddy-current lossesin the permanent magnets are analyzed and compared. Particularly, modelsadopting different formulations to the Helmholtz equation to solve for the eddycurrents are compared to a simpler model relying on an assumed eddy-currentdistribution. Boundaries in terms of magnet dimensions and angular frequencyare also identified in order to aid the machine designer whether the mostsimple loss model is applicable or not. With a prediction of the eddy-currentlosses in the permanent magnets together with a corresponding thermal model,predicted volumetric loss densities exemplified for combinations of slot andpole numbers common in automotive applications are presented along withthe associated thermal impact.

Keywords: Automotive applications, concentrated windings, eddy-currentlosses, fault tolerance, finite element analysis, fractional slot windings, heattransfer, integrated motor drive, interior permanent magnets, modular con-verter.

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Sammanfattning

Kompakta, elektriska drivsystem för fordonsdrifter representerar en viktigmöjliggörare för att kunna realisera morgondagens fossilfria transportlösning-ar. En attraktiv lösning är att integrera den kraftelektroniska omvandlarentillsammans med tillhörande elmaskin i en enhet. Denna avhandling, till-sammans med tillhörande publicerade rapporter, fokuserar kring design ochanalys av elektriska maskiner för integrerade elektriska drivsystem avseddaför fordonsdrifter. Särskilt fokus är lagt vid permanentmagnetiserade synkron-maskiner (PMSM) med interiört monterade permanentmagneter kombineratmed modulära omriktartopologier.

I avhandlingens första del beskrivs olika omriktarkoncept och lindningsar-rangemang lämpliga för ett integrerat drivsystem. Jämfört med en konventio-nell lösning nyttjande en trefasig tvånivåomvandlare så kan en kompakt lösningimplementeras genom att dela upp omriktaren och tillhörande mellanledskon-densator i ett antal submoduler. En medföljande fördel med modularisering äratt ett modulärt koncept även möjliggör en viss nivå av feltolerans.

I avhandlingens andra del analyseras koncentrerade delspårslindningar.Först presenteras en genomgång av hur lämpliga spår, pol och faskombinationerkan identifieras med hänsyn till lindningsfaktorn för den våg som är synkronmed rotorns varvtal samt de resulterande rotorförlusterna. Sedan inkluderasintegrerade, modulära omriktarkoncept och tillhörande lindningsarrangemangoch spår, pol och faskombinationer som även är lämpliga för modulära omrik-tarkoncept identifieras. Vidare analyseras två möjliga lindningsarrangemanglämpliga för stacked polyphase bridges (SPB) och parallel polyphase bridges(PPB) omriktare i det fall att en omriktarmodul har fallerat.

I avhandlingens tredje del presenteras en iterativ process som nyttjar bådefinita elementmetoden samt analytiska yttryck för att designa PMSMs medinteriört monterade permanentmagneter och koncentrerad delspårslinding. Deresulterande maskindesignerna illustrerar tradeoffs i termer av feltolerans,effektfaktor, momenttäthet samt potential för drift i fältförsvagningsområdet.En experimentell prototyp från givna specifikationer har även designats ochkonstruerats.

Slutligen, då koncentrerade delspårslindningar allmänt resulterar i ettstort övertonsinnehåll så har modeller för att beräkna virvelströmsförslusteri permanentmagneter jämförts och analyserats. Specifikt jämförs modellersom nyttjar två olika formuleringar av Helmholts ekvation för att beräknavirvelströmmarna mot en enkel modell där virvelströmsfördelningen ansätts.Gränser i termer av magnetdimensioner och vinkelfrekvenser har också iden-tifierats för att enkelt kunna avgöra om den enklare förlustmodellen kanappliceras eller inte. Med en noggrann prediktion av virvelströmsförlusterna irotorns permanentmagneter kombinerat med en tillhörande termisk modellhar volymetriska förlustätheter i permanentmagneterna samt resulterandemagnettemperaturer identifierats för spår- och poltalskombinationer som ärvanliga i fordonsdriftsapplikationer.

Nyckelord: delspårslindning, feltoleran, finita elementmetoden, fordons-drifter, integrerade elektriska drivsystem, interört monterade permanentmag-neter, modulära omriktare, virvelströmsförluster, värmeöverföring.

Acknowledgement

This work presented in this thesis has been carried out at the Department ofElectric Power and Energy Systems, School of Electrical Engineering (EES), KTHRoyal Institute of Technology. The project is started from September 2012. Asone of the integrated motor projects, the objective is to investigate the electricmachinery for the integrated motor drive system. The parallel projects, includingthe converter topologies investigation, control aspects of the overall system, andintegration schemes, are carried by Lebing Jin and Mojgan Nikouei.

Special thanks to Joachim Lindström at Volvo Car for providing very usefulsuggestions on the machine design. Also thanks to Åke Nyström and LennartEdström at Bevi for their practical experiences on prototype manufacturing. Manythanks to Jesper for installing the prototype machine in the laboratory.

At the beginning, I would like to express my gratitude to my supervisor Dr.Oskar Wallmark, for his support, guidance, help and enthusiasm. I also thank toMats Leksell for his technical knowledge and useful suggestion in the project, aswell as Dr. Staffan Norrga and Prof. Hans-Peter Nee throughout the project.

Special thanks to my co-worker, previous office mate and also friend, LebingJin, for her collaboration and company during the Ph.D. research and study. Manythanks to my previous and current office mates, Shafigh Nategh, Andreas Krings,Shuang Zhao, Mojgan Nikouei, Rudi Cavalerio Soares, Jonas Millinger, and AndersHagnestål for the support and help.

I am very grateful to all my former and current colleagues for the nice momentswe had at the department. Thanks to Matthijs for organizing wonderful Roebels’activities. I would like to thank Naveed, Luca, Panagiotis, Kalle, Yanmei, Erik, andArman for the enjoyable time during the conferences abroad and also at KTH. Manythanks to Alija, Simon and Nicholas for the helps in the laboratory. Thanks to Eva,Eleni, Brigitt and Viktor for the administrative works and laughs we shared. Further,special thanks to Peter for yearly updating the JMAG license and nice advices forvacations. Many thanks to Sjoerd Bosga for giving the valuable comments andadvices on my papers and works. Thanks to Arash, Christian, Tim, Diane, Stefanie,Keijo, Hongyang, Juan, Georg, Tomas and more for the lovely time we had in thisdepartment.

Last, but not least, I would like to thank my parents and brothers for theirendless support. Special thanks to my girlfriend Xizi for her love, understanding

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and encouragement.

Hui ZhangStockholm, June 2017

Contents

1 Introduction 11.1 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.2 Main objectives . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21.3 Main contributions of the thesis . . . . . . . . . . . . . . . . . . . . . 21.4 Outline of the thesis . . . . . . . . . . . . . . . . . . . . . . . . . . . 31.5 List of appended publications . . . . . . . . . . . . . . . . . . . . . . 41.6 Related publications . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2 Integrated motor drives 72.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72.2 Converter topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . 92.3 Motor topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122.4 Summary of chapter . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

3 Fractional-slot concentrated windings 173.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 173.2 Winding layout and resulting harmonic content . . . . . . . . . . . . 173.3 Rotor losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 183.4 Recommended winding layouts . . . . . . . . . . . . . . . . . . . . . 203.5 Summary of chapter . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

4 Electromagnetic design of FSCW-IPM machines 234.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 234.2 Design procedure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 244.3 Prototype design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254.4 Summary of chapter . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

5 Losses and thermal modelling 335.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 335.2 Loss calculation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 345.3 Thermal modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . 375.4 Summary of chapter . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

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6 Concluding remarks 416.1 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 416.2 Recommendations for future work . . . . . . . . . . . . . . . . . . . 42

A FSCW-IPM machine prototype 45

List of symbols 49

List of acronyms 51

List of Figures 53

List of Tables 55

Bibliography 57

Publication I 65

Publication II 73

Publication III 81

Publication IV 91

Publication V 113

Publication VI 121

Chapter 1

Introduction

This chapter provides a background and motivation of the research work presented inthis thesis and its appended papers. The main objectives and scientific contributionsare highlighted and a brief outline of the thesis content is presented. Finally, thepublications originating from this work are listed.

1.1 Background

As is well known, low-carbon, high efficient transportation solutions represent animportant mean to meet society’s demand on reduced emissions and increased energysecurity [1]. In such a scenario, pure-electric or hybrid-electric vehicles (EVs andHEVs), using electric drives to replace or assist the internal combustion engine (ICE),belong and could be considered as environmentally friendly. The development paceof EVs and HEVs has since 2010 accelerated dramatically and at the end of 2015,the entire world stock of EVs and plug-in HEVs had risen to 1.26 million. In [2],the target of the overall EV stock by 2020 is set to 20 million with almost half ofthese vehicles sales enabled by price reduction supported by different governmentalpolicies.

An electric drive can be divided two main parts, the electric machine andits associated power electronics (PEs). To achieve a compact electric drive withhigh efficiency and low cost, integration of the PE and the electric machine is anappealing and competitive solution [3, 4]. Thereby, cabling between the PE and theelectric machine can be substantially reduced, as well as improvements in termsof weight reductions and reduced electromagnetic interference (EMI). Further, ashared housing and cooling system is possible which further challenges the overallsystem design [4].

Permanent-magnet synchronous machines (PMSMs) are widely used in auto-motive applications due to their high efficiency and torque density. High-speeddesigns is an interesting trend aimed to reduce the machine volume provided thatthe associated mechanical and thermal issues can be handled [5, 6].

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2 Chapter 1. Introduction

The volume of the PE can potentially be further reduced by adopting a numberof low power-rating converters connected in series or in parallel. The conventionaldc-link capacitor, usually taking up approximately 30% of the PE volume in anautomotive traction application [7], is then divided into smaller units mountedin each converter unit. In this framework, suitable machine candidates shouldbe carefully selected to realize modularity to adapt to the introduced converterstructures and potentially also enabling a certain amount of post-fault capabilityfollowing, e.g., a short circuit in one of the converter units.

1.2 Main objectives

The main objectives of this thesis are as follows.

• To analyze suitable motor topologies adapted to the considered modularconverter concepts in order to realize a compact integrated motor drive intendedfor an automotive application.

• To determine winding arrangements layouts suitable for the considered mod-ular converter concepts with particular focus on the winding factor for themain harmonic, the associated rotor losses, and ease of machine-converterintegration.

• To study the potential for post-fault operation following a short circuit of asubmodule of the stacked polyphase bridges converter.

• To develop a suitable machine design process incorporating the above describeddemands for fault tolerance.

• To analyze suitable models for predicting eddy-current losses (and the associatethermal impact) induced in interior-mounted permanent magnets due to theharmonics arising from fractional-slot concentrated winding arrangements,particularly, to identify boundaries in terms of permanent magnet dimensionsand angular frequency where a very simple loss model, relying on an assumededdy-current distribution, is applicable or not.

• To provide a comparative evaluation of the studied modular machine andconverters topologies using the conventional two-level three-phase drive systemas reference.

1.3 Main contributions of the thesis

The main scientific contributions of the work presented in this thesis are:

1. In [Paper I], by exploiting the fundamentals of fractional-slot concentratedwindings (FSCWs), suitable slot, pole and phase combinations are identifiedto adapt to considered modular converter concepts.

1.4. Outline of the thesis 3

2. In [Paper II], an evaluation of post-fault operation with two different windingarrangement alternatives is presented for both concentrated and distributedwindings.

3. A previously reported iterative design process is extended in [Paper III] toinclude both finite element analysis and analytical methods for the design ofPMSMs with FSCWs. The resulting output illustrates the tradeoffs of theresulting designs in terms of fault tolerance, power factor, torque density, andpotential for field-weakening operation.

4. Three analytical models to estimate the losses in PMs of FSCW-IPM machinesare compared and analyzed in [Paper IV] and boundaries when the simplestmodel is applicable are identified.

5. Using part of the FreedomCAR 2020 specifications, a comparative evaluationof the studied modular machine and converter topologies is presented in[Paper VI] in terms of machine design aspects, power losses, capacitor energystorage requirements, costs, and cell redundancy using the conventional two-level three-phase drive system as reference.

1.4 Outline of the thesis

Organized in the form of compilation thesis, this Ph.D. thesis presents the keyconcepts and important results in each chapter. The scientific contributions aredetailed in the appended publications.

The thesis is outlined as:

Chapter 2 discusses recent integrated motor drives (IMDs) concepts intended forautomotive applications.

Chapter 3 reviews and discusses the theory of FSCWs, in terms of winding lay-outs, harmonic contents, and rotor losses. Suitable slot, pole, and phasecombinations are recommended for different modular converter concepts.

Chapter 4 outlines an iterative design process for fault-tolerant FSCW-IPM ma-chines.

Chapter 5 presents a brief introduction to thermal management and loss mecha-nism in electric machinery used in automotive applications. Particular focusis put on models for predicting the induced eddy-current losses in the PMsegments using a static, three-dimensional, thermal finite-element model topredict the resulting temperature distribution.

Chapter 6 concludes the results from the appended papers and proposes directionsfor future research.


1.5 List of appended publications

[Paper I] H. Zhang, O. Wallmark, M. Leksell, S. Norrga, M. Nikouei. Harnefors,and L. Jin, “Machine design considerations for an MHF/SPB-converter basedelectric drive,” in Proc. IECON 2014 - 40th Annual Conference of the IEEEIndustrial Electronics Society, pp. 3849–3854, Oct. 2014.The first publication reviews the modular high frequency (MHF) and stackedpolyphase bridges (SPB) converter concepts as potential candidates for inte-grated drives in automotive applications. The theory of FSCWs is discussedand suitable slot, pole, and phase combinations for the two modular converterconcepts are identified.

[Paper II] H. Zhang and O. Wallmark, “Evaluation of winding arrangementsin electric machinery for modular electric drives,” in Proc. 2016 IEEE 8thInternational Power Electronics and Motion Control Conference (IPEMC-ECCE Asia), pp. 2820–2825, May 2016.[Paper II] considers different winding connection configurations for a modularelectric drive based on series/parallel connected polyphase bridges converter.Suitable stator winding connection configurations are identified to adapt tothe converter topologies considered. Particular focus is put on the post-faultperformance.

[Paper III] H. Zhang, O. Wallmark, and M. Leksell, “An iterative FEA-basedapproach for the design of fault-tolerant IPM-FSCW machines,” in Proc. 201517th European Conference on Power Electronics and Applications (EPE’15ECCE-Europe), Sep. 2015.[Paper III] presents an iterative finite-element analysis method for the designof FSCW-IPM machines suitable for an integrated modular motor drive. Par-ticular focus is put on fault tolerance to handle a shorted converter submoduleand the level of fault tolerance versus compactness (torque density) and powerfactor is quantified in the form of a case study considering a 30 kW, 2000 rpmmachine.

[Paper IV] H. Zhang and O. Wallmark, “Limitations and constraints of eddy-current loss models for IPM motors with fractional-slot concentrated windings,”in Energies, vol. 10, no. 3, 2017.This journal paper compares different analytical models for predicting theaverage eddy-current losses in interior permanent-magnet (IPM) machineswith FSCWs considering only the armature reaction. Particularly, loss modelsadopting different formulations and solutions to the Helmholtz equation tosolve for the eddy currents are compared to a simpler model relying on anassumed eddy-current distribution. Boundaries in terms of magnet dimensionsand angular frequencies are identified to aid the machine designer whetherthe more simple loss model is applicable or not. The assumption of a uniform

1.6. Related publications 5

flux-density variation (used in the loss models) is also investigated for thecase of V-shaped and straight interior permanent magnets. Finally, predictedvolumetric loss densities are exemplified for combinations of slot and polenumbers common in automotive applications.

[Paper V] L. Jin, S. Norrga, H. Zhang, and O. Wallmark, “Evaluation of amultiphase drive System in EV and HEV Applications,” in Proc. 2015 IEEEInternational Electric Machines and Drives Conference (IEMDC), pp. 941–945,May 2015.This paper presents a performance evaluation for a 50 kW multiphase EV/HEVdrive system comprising a power converter using 1.2 kV SiC MOSFETs and apolyphase electric machine. Power loss calculation and cost comparison withrespects to increased number of phases, are evaluated to illustrate the benefitsof the proposed concept. The paper represents a first investigation in whateventually resulted in [Paper VI].

[Paper VI] H. Zhang, L. Jin and O. Wallmark, “Evaluation of Modular IntegratedElectric Drive Concepts for Automotive Traction Applications,” manuscriptsubmitted to IEEE Transactions on Transportation Electrification, 2017.In this paper, converter topology candidates suitable for an integrated modularmotor drive are considered with particular focus on the stacked polyphasebridges converter, the parallel-connected polyphase bridges converter, and themodular high frequency converter. A comparative evaluation of the studiedtopologies is presented and discussed in terms of machine design aspects, powerlosses, capacitor energy storage requirements, costs, and cell redundancy usingthe conventional two-level three-phase drive system as reference.

1.6 Related publications

The following publications include work related to a variety of aspects within thearea of electric machinery, which has been carried out alongside this project and inother projects where the author of this thesis has contributed to.

• H. Zhang, O. Wallmark and M. Leksell, “On fault tolerance for IPM-FSCWmachines adopting a modular converter,” in Proc. 2014 17th InternationalConference on Electrical Machines and Systems (ICEMS), pp. 1633–1638, Oct.2014.This paper investigates fault-tolerant capability of a number of 30 kW FSCW-IPM machines intended for the MHF- and SPB-type converters. Due to thestrong PM rotor magnetization when using NdFeB-type magnets, the resultingshort-circuit currents during a converter submodule short-circuit was found tobe larger than the rated current which, potentially, may cause the machineto overheat. Reducing the PM rotor magnetization will result in a reductionof the short-circuit current. The results further highlight the importance


of a tailored design specification when designing FSCW-IPM machines inautomotive applications.

• Y. Hu, A. Cosic, S. Östlund and H. Zhang, “Design and optimizationprocedure of a single-sided linear induction motor applied to an articulatedfuniculator,” in Proc. 2016 IEEE 8th International Power Electronics andMotion Control Conference (IPEMC-ECCE Asia), pp. 3096–3102, May 2016.This paper proposes a single-sided linear induction motor (SLIM) for anarticulated funiculator application. An analytical approach is provided toobtain a preliminary design based on approximate equivalent circuits underconstrained conditions. A two-dimensional finite element based analysis isapplied to verify the designs and identify suitable candidates.

Chapter 2

Integrated motor drives

This chapter, serving as a background for [Paper V] and [Paper VI], provides a briefdescription of different integrated motor drive (IMD) concepts intended for or usedin different applications.

2.1 Introduction

Nowadays, integrated motor drives (IMDs) are considered in a range of applicationswith a key benefit including the torque density that can be reached provided that theintegration of the electric machine and the corresponding power electronics can berealized compactly. Other benefits include the reduction of cabling with associatedreductions of electromagnetic interference and the possibility of a shared coolingsystem [8–10]. By simply mounting the three-phase converter onto the outer surfaceor axial end (non-driven side) of the electric machine housing (see the integrationschemes illustrated in Figure 2.1), a number of IMDs intended for both industrialand automotive applications are available today [11–13]. Other prototype-leveldesigns intended for a large variety of applications can be found in [14–17].

As exemplified in [10,16,18–21], the integration can be realized by adopting amodularized structure where the conventional converter is divided into a number ofsubmodules. These integrated modular motor drive (IMMD) concepts also enable acertain level of fault tolerance which is a potential benefit in automotive tractionapplications (as highlighted in, e.g., [22]). In an IMMD, the conventional three-or multi-phase converter is replaced by a number of converter units rated at lowerpower which are connected in series or parallel. Thereby, the relatively large dc-link capacitor (see below) can be divided into a number of smaller capacitor units(realized, e.g., with ceramic-type capacitors) mounted on each converter submodulewhich further promotes the system integration. However, an IMMD representsan increase in terms of system complexity and may also increase cost due to theincrease of gate drivers, current and voltage sensors and the distributed controlsystem required for each converter submodule. Furthermore, control algorithms on

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8 Chapter 2. Integrated motor drives

shaft rotor

stator

power

converter

shaft rotor

stator

po

wer

con

verter

shaft rotor

stator

po

wer

con

verter

(a) (b) (c)

Figure 2.1: Alternatives for power converter integration: (a) surface mount; (b)end-plate mount; (c) within axial-end interior mounting.

balancing different cells and eliminating high-frequency components of the batterycurrent need to be addressed [23].

As illustrated in Figure 2.2 (a), one feasible IMMD configuration is realizedby structurally separating a conventional converter into a number of single phase-leg inverters [10, 24] which is functionally identical to a conventional multi-phaseconverter. Other IMMD concepts are realized by series- or parallel-connecting anumber of three- or multi-phase converter submodules [18, 20, 21, 25–29] and areexemplified in Figure 2.2 (b).

(a) (b)

Figure 2.2: IMMD converters concepts: (a) phase-leg inverter; (b) modular-conceptconverter.

Regarding the placement and integration of the converter modules in the machine,the common configurations are categorized as [8]:

• Surface mount: In this configuration, illustrated in Figure 2.1 (a), theconverter (or converter modules) is mounted onto the surface of the machinehousing. This offers ease and low cost of manufacturing and the cooling system

2.2. Converter topologies 9

can be shared between electric machine and the converter. However, the totalsize of the IMD expands in the radial direction which may put limitations onits use in some applications.

• End plate mount: This integration is realized through mounting the con-verter (or converter modules) onto the external surface of the machine housingend plate as depicted in Figure 2.1 (b). A shared cooling system is possiblealso here. A potential drawback is that the end winding is located close tothe converter which must be considered when designing the cooling system.In the prototype manufactured within this project adopts this configuration(see Figure 2.3).

• Stator iron mount: The PE is mounted between the end plate and statorback iron. Both the stator iron and converter have to be modularized in asingular housing. The lack of thermal barrier between the PEs and the motormakes the thermal management more challenging;

rw*(coil span)/Qs*2*pi

L

Lc=

Figure 2.3: Prototype end plate (non drive end) for connection of the stator coils tothe respective converter submodules.

2.2 Converter topologies

Two-level three-phase converterA motor drive system in an automotive traction application commonly comprises ofa battery, a dc-link capacitor, a two-level three-phase converter and a three-phaseelectric machine. The conventional two-level three-phase converter is illustratedin Figure 2.4 and consists of three phase legs where each leg comprises of two


semiconductor switches with anti-parallel diodes. Silicon IGBTs with voltage ratingsin the range 600–1200 V are commonly selected since the battery voltage typically isin the range 300-600 V. In the dc side of the converter, typically a dc-link capacitorof electrolytic type is mounted which, being a relatively large and bulky devicehampers a compact integration in the conventional motor drive [7].

C

S1

S2

S3

S4

S5

S6

D1 D3 D5

D4D2 D6

+

-

ABC

idc

iaib

ic

vdc

Figure 2.4: Conventional two-level three-phase converter.

Phase-leg inverter conceptsTo enable a more compact converter and machine integration, the phase legs ofa three- or multi-phase motor drive (see Figure 2.5) can be physically separatedincluding a distribution of the dc-link capacitor. A high number of phase legs (eachrepresenting a converter submodule following the notation adopted in this thesis)are favorable in order to reduce the power rating of each converter submodule which,at the same time, increases the post-fault operation output power. Examples five,six, and nine phases are described in [10, 30–32], [24, 26, 28, 33], and [16, 34–36],respectively. Provided that a faulted phase leg is safely disconnected, post-faultoperation can be commenced, preferably with the current angles of remaining,non-faulted phase legs shifted to re-construct a balanced system [23,37–39].

Modular converter conceptsTo enable a modular machine-converter integration and, at the same time, obtaina high level of fault tolerance in case of faults arising in the electric machine, themachine windings can be arranged into a number of independent units, either asentirely separated single-phase windings or as number of separated three-phase ormulti-phase winding sets.

The modular high frequency (MHF) converter concept is a modular converterconcept presented relatively recently [19]. As illustrated in Figure 2.6, each sub-module of the MHF converter connects to a single phase winding of the electric

2.2. Converter topologies 11

C

SM1 SMN

S1

S2

A2

AN

A1E

idc

+

vdc

-

Figure 2.5: A multi-phase drive comprising of a multi-phase machine and corre-sponding converter phase legs.

machine. The submodule, in turn, consists of a step-up converter and an H-bridgeinverter with a capacitor in-between. This series-connected structure allows theuse of low-voltage power semiconductors, i.e., silicon MOSFETs or gallium FETswhich both enable high switching frequencies which, in turn, results in very smallcapacitors [40].

SMN

SM3

SM2

SM1

S2 S1

S4 S3

S6 S5

C

X1

X1

X2

X2

A1

A2

A1A2

idc

E

- vsm +

- vcap +

Submodule

Figure 2.6: The MHF converter concept [19].

In the converter topologies illustrated in Figure 2.7, each converter submoduleconsists of a two-level three-phase or multi-phase converter along with its dc capacitor.The ac terminal of each submodule is then connected to a set of star- or delta-


connected three- or multi-phase winding sets [41,42]. In the stacked polyphase bridges(SPB) converter concept, each converter submodule is connected in series [18,20],allowing the use of low-voltage power semiconductors, whereas in the parallelpolyphase bridges (PPB) converter, the submodules are connected in parallel [28,41].For both the SPB- and PPB-type converter concepts, interleaving of the PWM canbe exploited to cancel certain harmonics which, in turn, reduces the submodulecapacitor requirements. Post-fault operation can be potentially be realized by shortcircuiting the faulted submodule in the case of the MHF- and SPB-type converters orby disconnecting the faulted submodule the case of the PPB-type. The current flowduring post-fault operation for the MHF-, SPB-, and PPB-type converter conceptsare illustrated in Figure 2.8.

SMN

SM3

SM2

SM

SMN

SM3

SM2

SM

CX1X2

dci

(a) (b)

X1

X2

A1

A2A3

S1 S2

S3 S4

S5 S6

A1A2A3

E E

X2

X1A1

A2

A3

dci

Figure 2.7: (a) SPB-type converter; (b) PPB-type converter.

2.3 Motor topologies

Electric machinery with a relatively large variety in terms of specifications can befound in modern EVs [43]. During the last decade, a significant focus has beenput on the design of rotor topologies that would enable a high torque density withlittle or no use or rare-earth based permanent magnets. However, since the statorwinding connects to converter, also the stator structure is of special concern whenaddressing IMDs. Below, key stator and rotor concepts are briefly summarized.

2.3. Motor topologies 13

SMN

SM3

SM2

SM

X1

X2

A1

A2

A3X2

X1A1

A2

A3

SMN

SM3

SM2

SM

SMN

SM3

SM2

SM

X1

X2

A1

A2

sca,i

E

(a) (b) (c)

E E

sca,i

scb,i

dci dcidci

Figure 2.8: Post-fault operation: (a) MHF-type converter; (b) SPB-type converter;(c) PPB-type converter.

Stator

As evident from above, a stator winding with a modular structure is an interestingcandidate in order to enable a compact integrated solution. Preferably, each convertersubmodule should be placed closely to its corresponding set of stator windings [42].

The presence of relatively independent coil units in FSCWs provides a betteropportunity to realize a more compact machine converter integration than withinteger-slot distributed windings (ISDWs). The benefits of FSCWs include shortend-turns and a higher slot-fill factor (particularly with a segmented stator core [44])which further can contribute to reduce the total size of an IMDs targeted forautomotive applications [44,45]. Moreover, FSCWs display a higher fault tolerancedue to the lower mutual phase coupling compared to ISDWs. However, the, ingeneral, higher content of space harmonics in FSCWs may result in excessive eddy-current losses in the permanent magnets of the rotor which must be addressed bythe machine designer. Further, FSCWs may result in a lower effective saliency whichrequires a higher stator flux for the production of an identical torque compared toan ISDW counterpart. This may lead to lower efficiencies in the high-speed region,despite of overall reduction of copper losses for FSCWs [45,46].

Rotor

In Table 2.1, a summary of electric machine types adopted in recent EVs are listed(for other commercial EVs up to 2012, see [3]). As shown, induction machines (IMs)and permanent-magnet synchronous machines (PMSMs) are the most commonly


used machine types in commercial vehicles. Switched reluctance machines (SRMs)are also being considered for automotive traction applications, see, e.g., [47].

Table 2.1: Electric machine types in all-electric vehicle models.

Model EM type Max. power[kW] YearNissan Leaf PMSM 80 2016Fiat 500e PMSM 83 2015Chevy Spark EV PMSM 105 2015Ford Focus Electric PMSM 107 2015Kia Soul EV PMSM 81 2015Mercedes-Benz B250e IM 132 2015Tesla ModelS 60 IM 225 2015Toyota RAV 4 EV IM 115 2014BMW i3 PMSM 125 2014Honda Fit EV PMSM 92 2014Mitsubishi i-MiEV PMSM 49 2014Volkswagen e-up! PMSM 85 2013

Reviews of the merits of different rotor topologies in terms of cost, controlrobustness, speed range, and efficiency are presented in, e.g., [47–50]. From thesereviews, it is clear that synchronous machines with interior mounted permanentmagnets (IPMs) demonstrate a high torque density and a sufficiently wide speedrange making them a well suited rotor topology for automotive traction applications[50]. In this thesis, the combination of a FSCW with an IPM rotor is consideredbased on the arguments listed above. In Chapter 3 the theory of FSCWs is detailedand in Chapter 4, the design methodology for IPM-FSCW machines is presented.

2.4 Summary of chapter

A common electric drive for automotive traction applications commonly comprises ofan IPM-ISDW machine and its associated three-phase two-level converter. However,integrating these two components into a single unit may be troublesome due tothe relatively large capacitor that is required. An IPM-FSCW machine combinedwith a modular-type converter (illustrated in Figure 2.9) represents a solution thatpotentially is both very compact and has an inherent fault-tolerance.

2.4. Summary of chapter 15

§

Cap.

Cap.

MHF

SPB/PPB

Phase-leg

Figure 2.9: An IPM-FSCW machine with integrated converter submodules.

Chapter 3

Fractional-slot concentratedwindings

This chapter, serving as an introduction to [Paper I] and [Paper II] reviews anddiscusses theory for double-layer FSCWs with focus on finding suitable slot, pole,and phase combinations for IMD concepts.

3.1 Introduction

In fractional-slot concentrated windings (FSCWs), each coil is wound around a singlestator tooth resulting in a slot pitch (when measured in number of stator slots) yq toequal to 1. If the FSCW is of single-layer type, each slot is filled by a single coil side.The single-layer type winding displays a lower mutual phase coupling and a higherinductance compared its double-layer counterpart where each slot is filled by twocoils (exemplified in Figure 3.1). This is potentially beneficial both in terms of faulttolerance and enabling a wide constant-power speed range. However, the harmoniccontent is generally increased which limits the number of feasible slot, pole, andphase combinations [51]. For this reason, only double-layer FSCWs are consideredin this thesis. The number of (stator) slots per pole per phase qs is introduced as

qs = Qsmphp

. (3.1)

where Qs, p, and mph represent the number of stator slots, poles and phases,respectively. In integer-slot type distributed windings, qs is usually integer valuewhereas in a fractional-slot type winding, qs is a fractional value.

3.2 Winding layout and resulting harmonic content

As a method for determining winding arrangements that maximize magnitude ofthe main harmonic and, thus, developed torque, the star-of-slot method was applied

17

18 Chapter 3. Fractional-slot concentrated windings

Figure 3.1: Double-layer FSCW (photograph of experimental prototype beforewinding impregnation).

to FSCWs in [52]. Analogous to distributed windings (see also [52]), a closed-formexpression for the winding distribution factor (for harmonic order ν) kd,ν is given byequation (9) in [Paper I]. The pitch factor kp,ν for harmonic order ν can, from [53],be expressed as

kp,ν = sin(πν

Qs

). (3.2)

Thereby, the winding factor for harmonic order ν is obtained as

kν = kd,νkp,ν . (3.3)

With a closed-form expression for the winding factor for each harmonic orderν available, the resulting harmonic content for different combinations of Qs, p,and mph can be readily investigated. However, as demonstrated in [Paper I], theclosed-form expression for kd,ν (and, in turn, kν) is not valid for all combinationsof Qs, p, and mph. For example, consider the combination Qs = 24, p= 10, andmph = 3 for which the star-of-slot and resulting winding layout are illustrated inFigure 3.2. Directly adopting the closed-form expression for the winding factorsyields erroneous results as illustrated in Figure 3.3 where the magnitudes obtainedusing the closed-form expression is compared to computing the winding factorsnumerically. In [Paper I], a condition for when the closed-form expression of kν isvalid is provided.

Remark: The recent reference [54] also presents generalized winding factorexpressions for FSCWs.

3.3 Rotor losses

Except for the main harmonic (which is in synchronism with the rotation of therotor), each harmonic order ν risk to induce eddy-current losses in the permanent

3.3. Rotor losses 19

−1 −0.5 0 0.5 1−1

−0.5

0

0.5

1(a) Star of the slot

1

2

3

45

6

7

8

910

11

12

13

14

15

1617

18

19

20

2122

23

24

a b c

−2 −1 0 1 2−2

−1

0

1

2(b) Winding layout

12

34

56789

1011

12131415

161718192021

222324

a b c

Figure 3.2: Coil arrangement for Qs = 24, p = 10, and mph = 3: (a) Star of slot;(b) Resulting winding layout.

1 5 7 11 13 17 19 23 25 29 31 350

0.15

0.3

0.45

0.6← main harmonic

k ν

Harmonic orderν

FormulaNumerical

Figure 5: Machine designs forJ = 10 Arms/mm2 (red curves),J = 15 Arms/mm2 (blue curves) andJ =20 Arms/mm2 (black curves): (a) Outer diameter (Do); (b) Torque density; (c) Copper weight; (d) Total weight.

Machine 1

J=10 Arms/mm²

Ns=40

Fault tolerant

Machine 2

J=10 Arms/mm²

Ns=18

Non-fault tolerant

Machine 3

J=20 Arms/mm²

Ns=40

Fault tolerant

Machine 4

J=20 Arms/mm²

Ns=20

Non-fault tolerant

Figure 6: Comparison of fault tolerant and non-fault tolerantmachine designs.

References

[1] J. Tangudu, T. Jahns, and T. Bohn, “Design, analysis and loss minimization of a fractional-slotconcentrated winding IPM machine for traction applications,” inEnergy Conversion Congress andExposition (ECCE), 2011 IEEE, Sept 2011, pp. 2236–2243.

[2] S. Gjerde and T. Undeland, “Power conversion system for transformer-less offshore wind turbine,”in Power Electronics and Applications (EPE 2011), Proceedings of the 2011-14th European Con-ference on, Aug 2011, pp. 1–10.

[3] S. Norrga, L. Jin, O. Wallmark, A. Mayer, and K. Ilves, “A novel inverter topology for compactEV and HEV drive systems,” inProc. 39th Annual Conference of the IEEE Industrial ElectronicsSociety, IECON 2013, Nov 2013, pp. 6590–6595.

[4] Y. Han, “Design, modeling, and control of multilevel converter motor drive with modular designand split winding machine,” inControl and Modeling for Power Electronics (COMPEL), 2014IEEE 15th Workshop on, June 2014, pp. 1–10.

[5] B. Welchko, T. Jahns, W. Soong, and J. Nagashima, “IPM synchronous machine drive responseto symmetrical and asymmetrical short circuit faults,”Energy Conversion, IEEE Transactions on,vol. 18, no. 2, pp. 291–298, June 2003.

[6] H. Zhang, O. Wallmark, and Leksell, “On fault tolerance for IPM-FSCW machines adopting a mod-ular converter,” inElectrical Machines and Systems (ICEMS), 2014 17th International Conferenceon, Oct 2014, pp. 1633–1638.

Figure 3.3: Winding factors for the different harmonic orders for Qs=24, p = 10,and mph = 3 using the closed-form expression (light blue bars) and numericallycomputed (blue bars).

magnets of the rotor. For each harmonic order, the resulting eddy-current lossesPe,ν could be considered as proportional to the product B2

νω2ν where Bν is the

magnitude of the resulting air-gap flux density harmonic and ων the angular speedof the harmonic of order ν relative to the rotor (see, e.g., [55, 56]). Thereby, the

A+C-

B+

A-C+

B-

24s. 10p. 3m.

20 Chapter 3. Fractional-slot concentrated windings

total rotor eddy-current losses Pe can now be expressed as

Pe =∑ν

Pe,ν

∝∑ν

( p2ν

)2(

kνkν=p/2

)2 ( p2ν sgn(ν)− 1

)2

︸︷︷︸ploss

(3.4)

where ∝ denotes ‘proportional to’. The rotor loss indicator ploss in (3.4) is introducedin [57] as a criteria used for identifying combinations of Qs, p, and mph that shouldnot lead to excessive rotor losses. This rotor loss indicator is also adopted in[Paper I]. In [Paper IV] (and introduced in Chapter 5) more advanced eddy-currentloss models are adopted in order to compute the resulting eddy-current losses in thepermanent magnets more accurately.

Remark: The flux density harmonic in the permanent magnet is same as in theair gap for the machine with surface-mounted permanent magnets. However, forinterior-mounted permanent magnets, the flux density harmonic is proportional toBν given by equation (A11) in [Paper IV].

3.4 Recommended winding layouts

Apart from a low harmonic content and a high magnitude of the main harmonic,to enable a tight integration, the submodules of the converter should be placedclosely to the corresponding stator coils. Thereby, the connecting power cables canbe kept short which also is beneficial from an electromagnetic interference (EMI)point of view. In [Paper I], these considerations are summarized and recommendedcombinations of Qs, p, andmph are presented for the MHF- and SPB-type converters.

For the SPB- and PPB-type converters, each converter submodule is connectedto a set of three- (or multi-phase) windings (see Figure 2.7). These groups ofcoils (connected to the same converter submodule) can be termed sub windings.If the coils connected to the same converter submodule belong to consecutiveslots in the stator, the sub winding is termed modular. However, if the coilsconnected to the same converter submodule belong to different rotor pole pairswith a 2π/mph mechanical-angle shifting, the sub winding is termed symmetrical.Examples of modular and symmetrical sub windings are illustrated in Figure 3.4 andFigure 3.5, respectively. While the characteristics of the two winding types should beidentical in normal operation, the different placement of the coils results in somewhatdifferent characteristics in case of a faulted converter submodule. In [Paper II], thecharacteristics of modular and symmetrical sub windings during post-fault operation(failure of one or two converter submodules) are investigated and it is found thatalthough the considered symmetrical sub-winding connection configuration resultedin a somewhat reduced torque ripple during post-fault operation, the average(post-fault) torque was also reduced.


1

23

4

5 6

1

2

3

4

(a) (b)

A phaseB phaseC phase

Figure 3.4: Winding layout examples with modular sub windings: (a) Qs=12, p=8and mph =3; (b) Qs=18, p=14, and mph =3.

(a) (b)

A phaseB phaseC phase

Figure 3.5: Winding layout examples with symmetrical sub windings: (a) Qs=12,p=8, and mph =3; (b) Qs=18, p=14 and mph =3.


The chapter, serving as an introduction to [Paper I] and [Paper II], has reviewedkey concepts related to FSCWs and highlighted aspects to consider when combiningthe winding concepts with modular converter topologies.

Chapter 4

Electromagnetic design ofFSCW-IPM machines

This chapter outlines an iterative design method for fault-tolerant FSCW-IPMmachines intended for IMMDs. The method combines both finite-element analysis(FEA) and analytical methods and is further detailed in [Paper III].

4.1 Introduction

By combining the merits of an IPM rotor with a modular FSCW, a compactintegrated motor drive can potentially be realized fulfilling demanding applicationsincluding automotive traction [58,59]. A potential fault tolerance could be obtainedby short circuiting or disconnecting the faulted submodule as illustrated in Figure 2.8.For the MHF- and SPB-type converters where the submodules are connected inseries, the resulting steady-state short-circuited rms current Isc should not exceedthe rated rms current Ir in order to avoid overheating [60]. Further, the characteristicrms current Ich, defined as the ratio of the permanent-magnet flux linkage and thed-axis inductance, should be greater or equal to the rated current in order to obtaina wide speed range [45]. Above the low-speed region (where the resistive voltagedrop is not negligible compared to its inductive counterpart), Isc and Ich are relatedas

Isc ≈ Ich = ψm√2Ld

(4.1)

where ψm is the permanent-magnet flux linkage and Ld the d-axis inductance.Therefore, in order to design a fault tolerant FSCW-IPM machine with a wide speedrange, the following condition should hold:

Is 'ψm√2Ld

. (4.2)

23

24 Chapter 4. Electromagnetic design of FSCW-IPM machines

4.2 Design procedure

Enabled by the developments of computing power, electromagnetic analysis of electricmachinery using two-dimensional FEA combined with iterative- or optimization-based design methods are exemplified in, e.g., [61–63]. Within the scope of thisproject, an iterative design procedure to quantify torque density and fault tolerancefor FSCW-IPM machines suitable for a modular converter topology has beendeveloped. The method is detailed in [Paper III] and schematically illustrated inFigure 4.1. As seen in Figure 4.1 (b), the method is implemented using the softwaresMatlab1, JMAG2, Excel3, and Visual Basic.

The iterative process in Figure 4.1 (b), can be outlined as: Matlab is employed togenerate and store machine parameters in ’Matlab 1’. The parameters obtained aresaved as an Excel file. A Visual Basic (VB) script reads the machine parameters fromthe Excel file and generates the machine FE model accordingly. A FEM software’JMAG’ is utilized to analyze selected performance characteristics in ’JMAG 1’, andthese results are exported as txt files. Loading and evaluation of the FEA resultsare then realized in ’Matlab 2’.

The input (fixed) parameters include the winding arrangement, number ofpoles, the fundamental flux density in the air gap (i.e., with PM excitation only)and the rotor diameter. The current angle that achieves maximum torque-per-ampere operation θopt and the d- and q-axis inductance saturation ratios kd andkq are initially assumed and are (in ’JMAG 1’) updated in an iterative manneruntil convergence is reached. Other machine characteristics, including short-circuitcurrent in case of a short-circuited converter submodule and speed and torquecharacteristics, are computed in ’JMAG 2’.

Specification setsIn Table 4.1, three sets of machine specifications which have been addressed withindifferent work packages of this Ph.D. project are listed. Here, Tr is the ratedcontinuous torque at rated speed, Tm the maximum intermittent torque, Ωr therated speed, Ωm the maximum speed, Pr the rated mechanical output power, Pmthe maximum mechanical output power, Trip the allowed peak-to-peak torque ripple,Jc the continuous conductor current density, Jm the maximum conductor currentdensity, L the active length, Do the outer stator diameter, δ the air-gap thickness,and Udc the dc-link voltage.

In the third column, the specifications used in [Paper III] are listed. The fourthcolumn represents the ’FreedomCAR 2020’ specifications, when fulfilled are aimed atenabling the next generation of high-performance electric drives for plug-in hybridelectric vehicles (PHEVs). These specifications are compiled from [45,46,50,64, 65].

1Matlab is a registered trademark of The Mathworks Inc., Natick, MA, U.S.A.2JMAG is a registered trademark of the JSOL Corporation, Tokyo, Japan.3Excel and Visual Basic are registered trademarks of Microsoft Corporation, Redmond, WA,

U.S.A.

4.3. Prototype design 25

Fixed

parameters

Variables

Objective

function

Machine

parameters

FEA 1

VB scripts

Assumed

parameters

Minimum criteria ?

YesFEA Result 1

No

FEA 2

FEA Result 2

Matlab 1

JMAG 1

Excel

VB scripts

Txt 1

Matlab 2

JMAG 2

Txt 2

Txt 3

(a) (b)

Figure 4.1: Design process: (a) Schematic flowchart; (b) Corresponding softwareimplementation.

In the fifth column, specifications for the prototype that has been designed andbuilt within the framework of this Ph.D. project are listed.

Remark: Thermal aspects are here addressed only indirectly through the spec-ification of conductor current density levels within ranges that are common inwater-cooled electric machinery. A more detailed thermal analysis would be re-quired in order to, e.g., accurately predict thermal boundaries during intermittentoperation.

4.3 Prototype design

The iterative design process (outlined above and detailed in [Paper III]) has beenused to design a FSCW-IPM machine fulfilling the prototype specifications listed inTable 4.1. From the results in [Paper III], it was found that the machine designs thatare fault tolerant have short stack lengths, a low permanent-magnet flux linkage,


Table 4.1: Preliminary specification

Parameter Unit [Paper III] FreedomCAR 2020 Prototype spec.Tr Nm 150 102 85Tm Nm - 200 240Ωr rpm 2000 2800 4000Ωm rpm - 14000 12000Pr kW 31.4 30 35Pm kW - 55 100Trip Nm - 10 12Jc Arms/mm2 10∼20 10 10Jm Arms/mm2 10∼20 20 30L mm - <200 <200Do mm - <250 <220δ mm 0.5 0.75 0.75∼0.95Udc V 400 200∼450 400

and large slot areas with a high number of turns. This, in turn, results in relativelarge inductances. A drawback with these designs is that they display a lower powerfactor may result in higher associated converter costs.

To adapt to an SPB- or PPB-type converter, 12 slots and 8 poles were selected(see [Paper I]) which results in a winding factor for the main harmonic of kω,1 =kν=p/2 = 0.866. The no-load fundamental flux density in the air-gap (i.e., whenexcited only by the permanent magnets) was selected to Bδ = 0.65 T in order toavoid over-saturation during overload operation.

When aiming to fulfill the prototype specifications, the stator laminations becomehighly saturated (particularly in the teeth) at the highest conductor current densityJm = 30 Arms/mm2. This makes it difficult to reach the maximum intermittenttorque Tm. Therefore, in order to obtain the maximum torque, the machine isdesigned to produce approximately 50 kW at the rated speed for Jc=10 Arms/mm2.This ensures that the intermittent torque and maximum power is reached. Foroperation at rated power (35 kW) and rated speed, this results in a conductor currentdensity of around 7.2 Arms/mm2. However, the cooling system is still assumedto handle 10 Arms/mm2 for the short-circuit operation under a single submodulefailure with the SPB converter. Since the resulting steady-state short-circuit currentdensity is below 10 Arms/mm2, the machine design is considered as fault tolerant.

Prototype parametersA prototype of the resulting machine design has been manufactured and key param-eters are reported in Table 4.2. For the permanent magnets, VACODYM 854 APpermanent magnets coated with VACCOAT 20021 and manufactured by Vacu-umschmelze were used (see [66] for the associated material parameters). Further,

4.3. Prototype design 27

M250-35A type electric steel was used for both the stator and rotor laminations.The stator winding was vacuum impregnated using EpoxyLite (Elan-tron 4260).Due to a misunderstanding, also the rotor was vacuum impregnated using the sameimpregnation material. A slot filling factor of around 0.42 was obtained.

Table 4.2: Prototype FSCW-IPM parameters and dimension details.

Parameter Unit ValueNumber of slots - 12Number of poles - 8

Stator outer diameter mm 211.5Stator inner diameter mm 140

Stack length mm 130Air-gap thickness mm 0.75Shaft diameter mm 54Magnet width mm 14.93

Magnet thickness mm 7.16Slot Area mm2 517.8

Wire diameter mm 0.95Number of turns per slot - 16Number of parallel strands - 19

Rated current Arms 96.9Phase resistance @ 20 C Ω 0.016

d-axis inductance mH 0.41q-axis inductance mH 0.79

Permanent-magnet flux linkage Vs 0.081

The costs and weights for the active parts of the machine (including the endwinding but excluding the shaft and housing) are depicted in Figure 4.2 where thecosts 7.1 USD/kg, 2.1 USD/kg, and 82.3 USD/kg have been assumed for the copper,the electric steel, and the PMs (the cost figures have been obtained from [67]). Asexpected, the PM cost dominates the material costs. Compared to this fault tolerantdesign, a corresponding non fault tolerant design would display a lower inductance(less amount of copper) and require more PM material (yielding a larger value ofm) to produce the same torque. This would result in a somewhat higher power

factor but at the expense of an even higher PM cost.

Predicted prototype characteristics using FEA

A cross-sectional view of the FEA model including a sample computational meshand the flux-density distribution due to the PMs only are shown in Figure 4.3.In Figure 4.4, the flux density in the middle of the air gap due to the PMs only(no-load) is shown. As can be seen, the desired fundamental flux density in theair gap Bδ,1≈ 0.65 T is obtained and a distinct slot effect can be observed. The


Stator Rotor PM Copper0

5

10

15

Wei

ght [

kg]

(a)

Stator Rotor PM Copper0

50

100

150

Cos

t [$]

(b)

Figure 4.2: Weight and material cost for each part of the prototype: (a) Weight;(b) Cost.

predicted phase back-EMF waveform and its associated harmonic spectrum areshown in Figure 4.5.

Figure 4.3: Cross-sectional view of the prototype design with the flux-densitydistribution due to PMs only.

Figure 4.6 shows the predicted maximum shaft power (neglecting iron losses)at rated speed for different conductor current densities. As can be seen, the shaftpower increases approximately linearly up to 15 Arms/mm2. As described above,


0 90 180 270 360−0.75

−0.5

−0.25

0

0.25

0.5

0.65

Slot effectF

lux

dens

ity [T

]

Elec. angle [o]

BδBδ,1

Figure 4.4: Predicted air-gap flux density due to PMs only.

the prototype is designed according to point ’B’ ("design point", approximately10 Arms/mm2 and 50 kW), rated according to point ’A’ ("rated point", approx-imately 7.2 Arms/mm2 and 35 kW), and aimed to operate at point ’C’ ("peakpoint", approximately 30 Arms/mm2 and 100 kW) to obtain maximum power. Thepredicted torque-speed and power-speed curves at 7.2 Arms/mm2 are depicted inFigure 4.7. As seen, the prototype machine displays a large constant power-speedrange (CPRS) which fulfills the prototype specifications. The post-fault performanceof the prototype machine design is further evaluated in [Paper II].


The chapter has provided a brief description of the design process presented indetail in [Paper III]. The resulting design for the prototype based on prototypespecifications listed in Table 4.1 is also analyzed using FEA.


0 90 180 270 360−150

−100

−50

0

50

100

150

Bac

k−E

MF

[V]

Elec. angle [o]

1 3 5 7 9 11 13 15 17 19 210

25

50

75

100

125

150

Am

plitu

de [V

]

Harmonic order

Figure 4.5: Predicted back-EMF waveform (phase to neutral pount) at rated speedand its corresponding harmonic spectrum.

0 5 10 15 20 25 30 350

30

60

90

120

←Design point

Peak point→

Pow

er [

kW]

Current density [Arms/mm2]

Figure 4.6: Predicted maximum shaft power against the conductor current density(rms) at rated speed.

←Rated point


0 2 4 6 8 10 120

20

40

60

80

100

Tor

que

[Nm

]

0 2 4 6 8 10 120

10

20

30

40

50

Pow

er [

kW]

Speed [krpm]

Figure 4.7: Predicted torque-speed and power-speed curves.

Chapter 5

Losses and thermal modelling

This chapter provides a brief introduction to cooling arrangements, the electro-magnetic loss mechanisms, and the associated temperature distribution in electricmachinery considering in particular the FSCW-IPM prototype machine described inChapter 4. Particular focus is put on models for predicting the induced eddy-currentlosses in the PM segments using a static, three-dimensional, thermal finite-elementmodel to predict the resulting temperature distribution. The chapter serves as anintroduction to [Paper IV].

5.1 Introduction

Due to the demanding requirements in terms of torque- and power density, aliquid cooling system (e.g., a water jacket) is commonly employed in automotiveapplications in order to efficiently dissipate the generated losses. Such a waterjacket is usually built into the stator frame/housing. Considering the fluid flowpaths, the cooling channels can be directed in the circumferential direction (seeFigure 5.1 (a)), following a spiral path (see, e.g., [68–70]), or in the axial directionfollowing an s-type path as shown in Figure 5.1 (b) (also exemplified in, e.g., [71]).For the experimental prototype detailed in Chapter 4, an axial direction of thecooling channels was selected due to its simple manufacturing process.

To avoid excessive hot-spot temperatures (commonly occurring in the endwindings) or overheating the PMs in the rotor, a thermal model able to predictthe temperature distribution in different parts of the machine, is required whenthe requirements on the cooling system are to be set. The lumped parameter (LP)approach shown in Figure 5.2 is based on a simplified representation of the geometryand can provide fast results with adequate accuracy provided that the simplifiedgeometrical representation is done with care [71]. However, for geometrically complexparts, realizing a proper simplified geometrical representation can be challenging(see, e.g., [72, 73]). Numerical methods include thermal FEA (see, e.g., [70, 72]) andcomputational fluid dynamics (CFD) (see, e.g., [68,71]) where the CFD approach is

33

34 Chapter 5. Losses and thermal modelling

Rotor

Stator

Shaft

Cooling channel

Rotor

Stator

Shaft

Cooling channel

(a)

(b)

Figure 5.1: Sample liquid-cooling arrangements for electric machinery:(a) Circumferentially-directed cooling channels; (b) Axially-directed channels.

increasing in use enabled by the developments in computing power.

5.2 Loss calculation

The losses in the different parts of the electric machine represent heat sources in athermal model. An accurate calculation of the losses arising in the different parts ofthe machine is therefore necessary in order to predict the associated temperaturedistribution correctly. Particularly, the critical parts including the winding hotspots and the PMs in the rotor need to be treated carefully. In this section, theloss generating mechanisms are briefly reviewed. Only electromagnetic losses areconsidered, i.e., friction and windage losses area not included.

Copper lossesProvided that the frequency is low enough so that skin and proximity effects can beignored, The copper losses in the stator winding are simply obtained as

Pcu = mphRsI2s (5.1)

5.2. Loss calculation 35

Τ0

coolant

frame

yoke

coil sides

endwindings

magnets

bearings

teeth

Rth

Cth

ï ò

Figure 5.2: A simplified lumped parameter network for prediction of the temperaturedistribution in an electric machine.

where Rs is the dc phase resistance. To account for the temperature dependentelectrical conductivity of copper, the iterative approach shown in Figure 5.3 (whereTcu0 represents the pre-assumed copper temperature) has been adopted in thethermal model discussed in Section 5.3.

cu0TCopper

losses

Thermal

FEA 1cu,cu, ii TT

Yes

cuT

1cu, iT

No

Figure 5.3: Iterative process to determine the copper losses.


Iron lossesFor the M250-35A, SiFe laminated electrical steel used in the experimental prototype,the associated iron losses are not affected significantly by the temperature variationsthat can be expected in electric machinery [74]. In this work, the iron losses forthe experimental prototype have been computed from a 2D FEA using JMAG’sin-built iron loss-model (computed during the post-process of the FEA and assumingperfectly sinusoidal currents). Sample results (including predicted copper losses atroom temperature) are reported in Table 5.1.

Table 5.1: Predicted copper, iron, and total PM losses for the FSCW-IPM prototypeat rated load (4000 rpm and 85 Nm) and two high-speed operation points (9000 rpmand 50 Nm, 12000 rpm and 35 Nm).

Part 4000 rpm 9000 rpm 12000 rpmCopper @ 20 C 450 W 450 W 450 WRotor lamination 90 W 279 W 445 WStator lamination 509 W 714 W 1003 W

PMs (13× 16 segments) 14.9 W 77 W 135 W

PM lossesAt high temperatures, the PMs in the rotor may be in risk of becoming permanentlydemagnetized. The losses in the PMs are caused by induced eddy currents, mainlydue to the asynchronous stator MMF harmonics. Due to the so called "flux con-centrating effect" (see eqs. (A10) and (A11) in [Paper IV]), the PM losses can besignificantly reduced in PMSMs with an IPM rotor structure compared to PMSMswith surface-mounted PMs. A 3D-FEA can accurately predict the resulting PMlosses but the approach is generally very time-consuming [75]. In order to enablerapid thermal simulations, a simple model for predicting the eddy-current losses inthe PM segments of a FSCW-IPM is therefore attractive.

Consider the PM segment in Figure 5.4 which is assumed to be subjected toa uniform sinusoidal variation of the flux-density with a magnitude and angularspeed denoted with Bνm and ωνm , respectively. Assuming the eddy-current pathsillustrated in Figure 5.4, the volumetric loss density pm [W/m3] for the PM segmentcan then be expressed as (see [76] or Section 2.1 of [Paper IV])

pm =σmlmwmω

2νmB2νm

32(l2m + w2m) (5.2)

where σm is the conductivity of the PM material, and lm and wm the axial lengthand width of the PM segment.

However, whether (5.2) can be applied in order to predict PM losses in a FSCW-IPM needs to be investigated. Particularly, the following concerns need to beaddressed:

5.3. Thermal modeling 37

Figure 5.4: Eddy-current paths assumed in Model A [76].

• The stator MMF harmonics that induce eddy currents in the PMs are rotatingwith respect to the rotor. To what extent can a uniform flux-density variationof the PM segments be assumed?

• The expression given by (5.2) is not in agreement to the classical expressionfor eddy-currents in thin laminations. To what extent will this restrain thevalidity of the model in the case of short PM segments (i.e., if lmwm or ifwm lm)?

• At high frequencies, the eddy-current reaction fields will change the resultingeddy-current distribution and, in turn, change the associated eddy-currentlosses. When should these effects be considered?

In [Paper IV], the above concerns are analyzed in detail. Figure 5.5 representsa key result and it illustrates limits of (5.2) for different angular speeds ωνm

andPM dimensions. Sample total PM losses with 13 segments in the axial direction atdifferent rotor speeds are also reported in Table 5.1.

For the FSCW-IPM prototype machine, the width of the PM segment is wm≈15 mm. Figure 5.6 shows the predicted volumetric loss density pm (assumingoperation at rated (sinusoidal) current) in the PM segments at different rotor speedsfor different number of axial segments. The results are obtained using (5.2) whichis found to be valid for these frequencies and axial segmentation lengths.

5.3 Thermal modeling

A 3D-FE thermal model of the prototype machine has been implemented in JMAGwith the intention to determine approximately how different values of volumetricloss densities in the PM segments pm correspond to resulting PM temperatures.


Version February 26, 2017 submitted to Energies 7 of 20

3.2. Limits for Model A Due to Eddy-Current Reaction Fields113

As seen, the expressions for the volumetric loss density pm given by Model A, Model B, and114

Model C, are of increasing complexity. An interesting issue is therefore to determine the boundaries115

when effect of116

eddy- encies.117

Howe her lm ≪118

wm or in [27],119

respe120

To relative

errors

(22)

(23)

In Figu tted for121

ωνm/ 94 kS/m122

(typic tched as123

black Hence,124

Mode Figure 3125

(or its validity126

of the

0 20 40 60 80 1000

20

40

60

80

100

300

3000

lm (mm)

wm

(mm

)

ωνm /(2π) =

30 e direction of

i

127

3.2.1.128

Fi ries in terms of129

magn lest loss model130

(Mod 27], a number of131

appro roximation using132

typica s, it is found that133

none ciently accurate.134

Figure 5.5: Limits (more than 20% of error) of (5.2) for ωνm/(2π) = 300 Hz to

ωνm/(2π)=3000 Hz in steps of 300 Hz assuming µr=1.04 and σm=694 kS/m. Thearrow denotes the direction of increasing ωνm . The black patched regions representPMs with dimension either lm wm or wm lm, where (5.2) will underestimatethe losses (more than 20% of error).

The implemented thermal model is similar to the thermal model described in [72]and key assumptions are detailed below.

The outer surface of the stator lamination is (due to the water cooling arrange-ment) assumed to be kept at 60 C. The effective heat-transfer coefficient in the airgap heq,δ is approximated using the expressions in [77] and it is shown as functionof rotor speed in Figure 5.7. The thermal conductivity of the winding impregna-tion (EpoxyLite) set to 0.68 W/(m·K) (obtained from [78]) and for the laminatedsteel and the PMs, 28 W/(m·K) [79] and 8 W/(m·K) [80] is assumed. Figure 5.8shows a sample temperature distribution assuming a PM volumetric loss density ofpm=0.1 W/cm3. In Figure 5.9, the resulting PM temperature is plotted for differentvalues of the PM volumetric loss densities pm. The results from the thermal modelare used in [Paper IV] to determine whether different slot and pole combinationswould result in excessive PM temperatures.


This chapter has briefly reviewed key aspects of thermal modeling of electric machin-ery and the associated electromagnetic loss mechanisms. Particular focus was puton models for predicting the eddy-current losses in PM segments and the resultingtemperature distribution. These models are analyzed in more detail in [Paper IV].


13

57

911

13 03000

60009000

12000

0

0.5

1

1.5

2

Speed [rpm]No. of segments

p m [

W/c

m3 ]

Figure 5.6: Volumetric loss density against the rotor speed under different numbersof PM segments.

0 3000 6000 9000 1200040

60

80

100

120

140

X: 3000Y: 52.19

h eq,δ [

W/(

m2 K

)]

Speed [rpm]

Figure 5.7: Equivalent heat transfer coefficient in the air gap as function of therotor speed.


0 0.5 1 1.5 260

90

120

150

180

pm (W/cm3)

Tm

ag

(oC

)

(a)(b)

Figure 5.8: Resulting temperature distribution with pm = 0.1 W/cm3 at ratedcurrent 9000 rpm (winding impregnation not shown).

Energies 2017, 10, 379 13 of 19

impregnated in epoxy with an assumed effective thermal conductivity of 0.68 W/(m·K). The outersurface of the stator lamination is fixed to 60 C representing a water cooling jacket with a coolanttemperature typical of what is found in automotive applications. A sample result from the 3D-FEMthermal simulation is depicted in Figure 12, and the resulting PM temperatures are reported inFigure 13. The results in Figure 13 will be used below in order to provide approximate intervals of pm,which potentially can result in excessive PM temperatures in an automotive application.

Figure 12. Implemented 3D-FEM thermal model with pm =0.1 W/cm3 at rated current and 9000 rpm(winding impregnation not shown).

Version February 26, 2017 submitted to Energies 14 of 20

Figure 12. Implemented 3D-FEM thermal model with pm =0.1 W/cm3 at rated current and 9000 rpm

(winding impregnation not shown).

0 0.5 1 1.5 260

90

120

150

180

pm (W/cm3)

Tm

ag

(oC

)

Figure 13. Resulting PM average temperature as function of PM volumetric loss density pm at rated

current and 9000 rpm. The considered low, medium and high temperature ranges have been indicated

using green, orange and red colors, respectively.

5. Conclusions231

In this paper, three models for predicting average magnet losses in IPMs with FSCWs due to232

induced eddy currents caused by the armature reaction (assuming sinusoidal phase currents) were233

analyzed and compared. Provided that the harmonic order ν that dominates the PM losses is less234

or equal to to the number of poles p, it was found that that for V-shaped PMs, a uniform flux235

density variation in the PMs can typically be assumed. Boundaries in terms of PM dimensions236

and angular frequency were identified to aid the machine designer whether the simplest loss model237

(Model A) is applicable or not. An approximate analytical expression to these boundaries was also238

identified. It was found that Model B and Model C, while relying on different formulations and239

solutions of Helmholtz equation, provide very similar loss predictions in very good agreement with240

corresponding 3D-FEM based simulations. Further, tables were provided with resulting volumetric241

loss densities pm for combinations of Qs and p commonly used in automotive applications. Finally, by242

compiling results from previous publications, a complete description on how to analytically predict243

the magnitudes Bνm and corresponding harmonic orders νm for FSCW-IPMs was provided in the244

appendix.245

Figure 13. Resulting PM average temperature as a function of PM volumetric loss density pm at ratedcurrent and 9000 rpm. The considered low, medium and high temperature ranges have been indicatedusing green, orange and red colors, respectively.

4.2. Losses for p and Qs Common in Automotive Applications

Now, resulting PM eddy-current losses for combinations of p and Qs often considered inautomotive applications are considered. The harmonic content from double-layer FSCWs is consideredin this paper. For single-layer FSCWs, the harmonic content is described in [8,12,14]. The rotor radiusrr and air-gap length δ are selected identical to what is reported in Table A1, and the main harmonicstator MMF (the ampere-turns and winding factor kν=p/2 product) is also kept the same, meaningthat each resulting machine design results in a similar output torque (the harmonic content will bedifferent, however, for each combination of p and Qs). The pole-cap coefficient is selected as αp =3/4to yield a rotor saliency, and the PM height is selected as hm = 5 mm. Further, the PM width wm isselected so that the no-load flux density in the air gap is 0.75 T. For p=8, 10, 12 and 14, this results inwm =14.2, 11.3, 9.5 and 8.1 mm. Resulting rotor geometries for p=8 and p=14 (with magnetic bridgesand air pockets inserted) are depicted in Figure 14.

The resulting PM loss densities for lm = 10 mm and lm = 30 mm are reported in Tables 1 and 2,respectively. As seen, only a few combinations of p and Qs result in sufficiently low eddy-current

Figure 5.9: Thermal impact of PM volumetric loss density pm on the resulting PMaverage temperature at rated current and 9000 rpm. The considered low, mediumand high temperature ranges have been indicated using green, orange and red colors,respectively.

Chapter 6

Concluding remarks

This chapter summarizes conclusions and provides some suggestions for furtherresearch directions related to the different topics considered in this work.

6.1 Summary

This thesis and its appended papers represents an approach to comprehensivelyinvestigate and analyze electric machinery suitable for modular converter conceptsintended to form a compact, integrated, automotive motor drive. First, differentmodular winding arrangements suitable for a modular converter concept werediscussed since a modularized structure of both the converter and machine sideis preferable in order to realize a compact integration. Three modular convertertopologies, namely, the MHF-, SPB-, and PPB-type converters have been consideredand been compared to a conventional, two-level three-phase converter and a multi-phase converter comprising of multiple single-phase legs.

The theory of double-layer FSCWs, especially the impact of different slot, pole,and phase combinations, has been reviewed in terms of the resulting harmoniccontents of the MMF waveform (affecting the rotor losses) and the winding factors forthe main harmonic. Further, recommendations for winding layouts suitable for IMDswere identified. Modular and symmetrical sub-winding arrangements were introducedand discussed when combined with the SPB- and PPB-type converters and it wasfound that the symmetrical sub-winding arrangement displayed a somewhat lowertorque ripple during post-fault operation. However, the modular sub-windingarrangement resulted in a higher average (post-fault) torque and is, since it alsorepresents a more simplified machine-converter interface, recommended.

An iterative process utilizing both FEA and analytical methods was introducedfor the design of FSCW-IPMs. The resulting output illustrates the tradeoffs ofthe resulting designs in terms of fault tolerance, power factor, torque density, andpotential for field-weakening operation. Following a given set of specifications, an

41

42 Chapter 6. Concluding remarks

experimental prototype suitable for an SPB-type converter was also designed andbuilt.

A FSCW results in a larger harmonic content of the MMF waveform comparedto a conventional distributed winding. This, in turn, increases the induced eddy-current losses in the PMs of the rotor. Therefore, models for predicting these lossesin FSCW-IPMs were analyzed and compared. Particularly, loss models adoptingdifferent formulations and solutions to the Helmholtz equation to solve for the eddycurrents were compared to a simpler model relying on an assumed eddy-currentdistribution. Boundaries in terms of PM dimensions and angular frequency were alsoidentified in order to aid the machine designer whether the more simple loss modelis applicable or not. By combining an accurate prediction of the eddy-current lossesin the PMs with a corresponding thermal model, predicted volumetric loss densitiesexemplified for combinations of slot and pole numbers common in automotiveapplications were presented along with the associated thermal impact.

Finally, the work packages above were combined and a comparative evalua-tion of the studied modular machine and converter topologies was presented (in[Paper VI]) in terms of machine design aspects, power losses, capacitor energystorage requirements, costs, and cell redundancy using the conventional two-levelthree-phase drive system as reference. From this evaluation, it was found that aconventional distributed winding represents a compact solution, particularly if theaxial build of the end winding part can be made short. It was also found that thatthe post-fault capability for all modular converter concepts considered was rela-tively similar and is mainly dependent on the number of submodules utilized. Thecost comparison indicated that both the PPB-type converter and the multi-phaseconverter comprising of several single-phase legs potentially can meet the powerconverter cost requirements since the cost of the semiconductor devices are less than50% of the FreedomCar 2020 cost requirements. This enables the largest margin forthe additional costs associated with the modularity of the two concepts.

6.2 Recommendations for future work

This thesis has analyzed suitable winding concepts for modular converter concepts,considered electromagnetic design of the corresponding electric machine, and pre-sented an analysis of PM losses and its associated thermal impact. However, thework is not completed and the following directions for future work are recommended.

First, a detailed experimental evaluation should be carried out in order to verifythat the predicted specifications of the prototype machine are fulfilled. In order toaccomplish this, the corresponding SPB-type converter prototype (being developedin a parallel project) needs to be commissioned. While larger deviations are not to beexpected in terms of the prototype machine’s electromagnetic characteristics, such anexercise will likely provide useful practical information that can be useful in the eventthat a next generation prototype, more similar to an actual product, is completed.An important event to investigate is the transient-short circuit current arising if a

6.2. Recommendations for future work 43

converter submodule is short circuited. Here, the risk of permanent demagnetizationand increase of induced eddy-current losses in the PMs are important aspects toconsider.

The liquid-cooling system for the converter submodules and the electric machineshould be analyzed and dimensioned together. It is recommended to developparameterized thermal FEM and CFD models to analyze the coolant flow and heattransfer from the stator outer surface and the power semiconductors to the coolingchannels. With such a model available, it is expected that the size of the coolingsystem can be reduced significantly; enabling a more compact integrated solution.

Last but not least, the results in [Paper VI] indicate that the PPB-type converteris an attractive solution also from a cost perspective. It would be interesting tostudy this IMD solution in more detail, including other applications, e.g., drivesused in pumps, fans, aerospace, and marine applications.

Appendix A

FSCW-IPM machine prototype

Figure A.1: The stator core of a 35 kW FSCW-IPM machine prototype .

45

46 Chapter A. FSCW-IPM machine prototype

Figure A.2: The rotor lamination of the prototype .

Figure A.3: The stator end windings.

47

Figure A.4: The impregnated stator winding.

SM1

Y1

SM2

SM3

SM4

Figure A.5: View of non-driven side.

List of symbols

Bδ,1 Magnitude of the no-load fundamental air-gap flux density

Bν Magnitude of the air-gap flux density for harmonic ν

Bνm Magnitude of the flux-density variation in PMs for harmonic νm

Do Outer stator diameter

heq,δ Effective heat-transfer coefficient in the air gap

Ich Characteristic rms current

Ir Rated rms phase current

Is Stator rms phase current

Isc Steady-state short-circuit rms current

Jc Continuous conductor rms current density

Jm Maximum conductor rms current density

kd D-axis saturation ratio

kq Q-axis saturation ratio

kd,ν Stator winding distribution factor for harmonic ν

kp,ν Stator winding pitch factor for harmonic ν

kν Stator winding factor for harmonic ν

lm Axial length of the PM segment

L Active length

Ld D-axis inductance

Lq Q-axis inductance

49


mph Number of phases

p Number of poles

Pcu Copper losses

Pe,ν Eddy-current losses for harmonic ν

Pe Total rotor eddy-current losses

Pm Maximum mechanical output power

Pr Rated mechanical output power

qs Number of stator slots per pole per phase

Qs Number of stator slots

Rs Dc phase resistance

Tcu0 Pre-assumed copper temperature

Tm Maximum intermittent torque

Tr Rated continuous torque at rated speed

Trip Peak-to-peak torque ripple

Udc Dc-link voltage

wm Width of the PM segment

δ Air-gap thickness

θopt Current angle for MTPA operation

µr Relative permeability of the PM

ν Stator MMF harmonic order

σm Conductivity of the PM

ψm Permanent-magnet flux linkage

ων Angular speed of the harmonic of order ν relative to the rotor

ωνmAngular speed of the flux-density variation in PMs for harmonic νm

Ωm Maximum speed

Ωr Rated speed

sgn(ν) Direction of the harmonic of order ν relative to the rotor

List of acronyms

CPRS Constant power-speed range

CFD Computational fluid dynamic

DW Distributed winding

EV Electric vehicle

EMI Electromagnetic interference

EMF Electromotive force

FSCW Fractional-slot concentrated winding

FEM Finite-element method

FE Finite-element

FET Field-effect transistor

GaN Gallium nitride

HEV Hybrid electric vehicle

ICE Internal combustion engine

IMD Integrated motor drive

IMMD Integrated modular motor drive

ISDW Integer-slot concentrated winding

IM Induction machine

IPM Interior permanent magnet

IGBT Insulated-gate bipolar transistor

LP Lumped parameter

51


MHF Modular high frequency

MMF Magnetomotive force

MTPA Maximum torque-per-ampere

MOSFET Metal-oxide-semiconductor field-effect transistor

PHEV Plug-in hybrid electric vehicle

PMSM Permanent-magnet synchronous machine

PM Permanent magnet

PPB Parallel polyphase bridge

PWM Pulse width modulation

PE Power electronics

SLIM Single-sided linear induction motor

SRM Switched reluctance machine

SPB Stacked polyphase bridge

SiC Silicon carbide

Si Silicon

WBG Wide-bandgap

3D Three-dimensional

List of Figures

2.1 Alternatives for power converter integration: (a) surface mount; (b)end-plate mount; (c) within axial-end interior mounting. . . . . . . . . . 8

2.2 IMMD converters concepts: (a) phase-leg inverter; (b) modular-conceptconverter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.3 Prototype end plate (non drive end) for connection of the stator coils tothe respective converter submodules. . . . . . . . . . . . . . . . . . . . . 9

2.4 Conventional two-level three-phase converter. . . . . . . . . . . . . . . . 102.5 A multi-phase drive comprising of a multi-phase machine and correspond-

ing converter phase legs. . . . . . . . . . . . . . . . . . . . . . . . . . . . 112.6 The MHF converter concept [19]. . . . . . . . . . . . . . . . . . . . . . . 112.7 (a) SPB-type converter; (b) PPB-type converter. . . . . . . . . . . . . . 122.8 Post-fault operation: (a) MHF-type converter; (b) SPB-type converter;

(c) PPB-type converter. . . . . . . . . . . . . . . . . . . . . . . . . . . . 132.9 An IPM-FSCW machine with integrated converter submodules. . . . . . 15

3.1 Double-layer FSCW (photograph of experimental prototype before wind-ing impregnation). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.2 Coil arrangement for Qs = 24, p = 10, and mph = 3: (a) Star of slot;(b) Resulting winding layout. . . . . . . . . . . . . . . . . . . . . . . . . 19

3.3 Winding factors for the different harmonic orders for Qs= 24, p = 10,and mph = 3 using the closed-form expression (light blue bars) andnumerically computed (blue bars). . . . . . . . . . . . . . . . . . . . . . 19

3.4 Winding layout examples with modular sub windings: (a) Qs=12, p=8and mph =3; (b) Qs=18, p=14, and mph =3. . . . . . . . . . . . . . . . 21

3.5 Winding layout examples with symmetrical sub windings: (a) Qs=12,p=8, and mph =3; (b) Qs=18, p=14 and mph =3. . . . . . . . . . . . . 21

4.1 Design process: (a) Schematic flowchart; (b) Corresponding softwareimplementation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

4.2 Weight and material cost for each part of the prototype: (a) Weight;(b) Cost. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

53

54 List of Figures

4.3 Cross-sectional view of the prototype design with the flux-density distri-bution due to PMs only. . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

4.4 Predicted air-gap flux density due to PMs only. . . . . . . . . . . . . . . 294.5 Predicted back-EMF waveform (phase to neutral pount) at rated speed

and its corresponding harmonic spectrum. . . . . . . . . . . . . . . . . . 304.6 Predicted maximum shaft power against the conductor current density

(rms) at rated speed. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 304.7 Predicted torque-speed and power-speed curves. . . . . . . . . . . . . . 31

5.1 Sample liquid-cooling arrangements for electric machinery: (a) Circumferentially-directed cooling channels; (b) Axially-directed channels. . . . . . . . . . 34

5.2 A simplified lumped parameter network for prediction of the temperaturedistribution in an electric machine. . . . . . . . . . . . . . . . . . . . . . 35

5.3 Iterative process to determine the copper losses. . . . . . . . . . . . . . 355.4 Eddy-current paths assumed in Model A [76]. . . . . . . . . . . . . . . . 375.5 Limits (more than 20% of error) of (5.2) for ωνm

/(2π) = 300 Hz toωνm/(2π) = 3000 Hz in steps of 300 Hz assuming µr = 1.04 and σm =694 kS/m. The arrow denotes the direction of increasing ωνm . Theblack patched regions represent PMs with dimension either lm wm orwm lm, where (5.2) will underestimate the losses (more than 20% oferror). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

5.6 Volumetric loss density against the rotor speed under different numbersof PM segments. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

5.7 Equivalent heat transfer coefficient in the air gap as function of the rotorspeed. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

5.8 Resulting temperature distribution with pm=0.1 W/cm3 at rated current9000 rpm (winding impregnation not shown). . . . . . . . . . . . . . . . 40

5.9 Thermal impact of PM volumetric loss density pm on the resulting PMaverage temperature at rated current and 9000 rpm. The considered low,medium and high temperature ranges have been indicated using green,orange and red colors, respectively. . . . . . . . . . . . . . . . . . . . . . 40

A.1 The stator core of a 35 kW FSCW-IPM machine prototype . . . . . . . 45A.2 The rotor lamination of the prototype . . . . . . . . . . . . . . . . . . . 46A.3 The stator end windings. . . . . . . . . . . . . . . . . . . . . . . . . . . 46A.4 The impregnated stator winding. . . . . . . . . . . . . . . . . . . . . . . 47A.5 View of non-driven side. . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

List of Tables

2.1 Electric machine types in all-electric vehicle models. . . . . . . . . . . . 14

4.1 Preliminary specification . . . . . . . . . . . . . . . . . . . . . . . . . . . 264.2 Prototype FSCW-IPM parameters and dimension details. . . . . . . . . 27

5.1 Predicted copper, iron, and total PM losses for the FSCW-IPM prototypeat rated load (4000 rpm and 85 Nm) and two high-speed operation points(9000 rpm and 50 Nm, 12000 rpm and 35 Nm). . . . . . . . . . . . . . . 36

55

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