power distribution network design for high-speed printed circuit...
TRANSCRIPT
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Power Distribution Network Design for High-Speed Printed Circuit
Boards
Jun Fan
NCR Corporation
April 3, 2007
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Outline
• Overview of PDN design in multi-layer PCBs
• Interconnect Inductance
• Individual Capacitor Values
• Capacitor Location
• Power/Ground Plane Pair
• Summary
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PDN in a Multi-Layer PCB
GND
VCC
DC/DC converter
(Power source)
GNDVCC
IC load
VCCGND
SMT capacitors
VCC
GND
IC driver VCC
GND
electrolyticcapacitor
VCC
GND
GND
IC driver
VDC
switch
Z0, vp
IC load
CL
VCC
RSP-MOS
N-MOS PCB trace
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Device Switching
IC driver
VCC
RS
shoot-through current GND
Z0, vp
IC load
VCC
charge
logic 0→1
IC driver
Z0, vp
IC load
0 V
discharge
VCC
RS
shoot-through current GND logic 1→0
Current drawn from power
J. L Knighten, B. Archambeault, J. Fan, et. al., “PDN Design Strategies: IV. Sources of PDN Noise,” IEEE EMC Society Newsletter, Winter 2007, Issue No. 212, pp. 66-76.
Shoot-through
current only
Shoot-through + load charging
current
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Current Supply and Power Bus Noise
VN
VCC
d
IC driver
GND
EMI
EMI
I/O line or cableVnoiseVCC
IC 2
GND
I am Mr. Charge!
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PDN Design Objectives
1. Ensure charge supply for logic transitions• Enough capacitance to store
charge • Enough charge readily
available for short transitions2. Minimize noise voltage
distribution on the VCC/GND plane pair• Low power bus impedance
over frequency• Noise decoupling• Noise isolation
IC driver
VCC
RS
shoot-through current GND
Z0, vp
IC load
VCC
charge
logic 0→1
VN
VCCd
IC driver
GND
EMI
EMI
I/O line or cableVnoiseVCC
IC 2GND
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PDN Design Issues in PCB• Capacitor interconnects; • Individual capacitor values and packaging forms;• Number of capacitors needed; • Capacitor placement;• PCB stack-up;• Power/ground plane pair geometry;• Segmentation and isolation
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PDN Charging/Discharging Hierarchy
The planes can be regarded as a C only @ low frequencies
VDC
Lbulk
Cbulk
Lvia
CSMT
Cplanes
Connectors and wiring Capacitor leads
Electrolytic capacitors
Interconnect
SMT capacitors
VCC/GND planeUltimate Charge Source 100’s uF 0.01-100’s uF pF’s-100 nFVRM (DC/DC converter)
Pin interconnect
BGA packagedIC pins
Speed of charge delivery
Amount of charge available for delivery
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PDN Impedance Profile
10-4
10-2
100
-50
-40
-30
-20
-10
0
10
20
F req uency [G Hz]
Z 21 [d
BΩ]
Power plane distributed behavior
Global decoupling
Bulk cap
Local decoupling
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Outline
Interconnect Inductance- shall be minimized in any situation
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Effect of Inductance
IC loadCLdecoupling capacitor
GNDVCC
Inductance associated with the capacitor: package parasitics, bonding pads, board traces, and vias
Inductance – impedes the flow of current (charge) to the load that will charge it to achieve a logic 1
CjLjRZ
ωω 1++=
LCf res π2
1=
L
C
t=0
v(0)=VS
i(t)
R
)sin()0(LCt
LCVti S=≥
L
C
L
C
R
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Minimizing Interconnect Inductance
SMT Decoupling Capacitor ESL
Loop 1 Lvia
Loop 2 Labove
The “Good”
The “Bad”
The “Ugly”Really “Ugly”
The “Better” The “Best”
Via
Capacitor Pads
SMT Capacitor
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Outline
Individual Capacitor Values- multiple values or maximum value? May not matter
LCf res π2
1=
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Two Common SMT Decoupling Strategies
● Use an array of capacitor values: – This may be the best known approach and is very popular in the
signal integrity design community (SI-LIST).– Rationale: to maintain a flat impedance profile below a target
impedance over a wide frequency range– Typically a logarithmically spaced (10, 22, 47, 100, 220, 470nF,
etc.) array of 3 values per decade.
● Use the largest capacitor value in the package size– This is less well-known, but a popular approach in the EMI
design community– Rationale: to keep impedance as low as possible, less emphasis
on a target impedance and a flat profile
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-20 dB/dec
+20 dB/dec
decoupling capacitors
distributed behavior
Different Approaches : ComparisonApproach A : values of decoupling capacitors logarithmically spaced, i.e. 3 values per decade : 10, 22, 47, 100, etc.
Approach B : largest values of decoupling available in two package sizes, i.e., 0603 and 0402
Approach B1 : largest values of decoupling available in one package size, i.e., 0402.
Target Impedance
Both approaches can meet the design specs relative to the target impedance
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Outline
Capacitor Location- could be important due to PCB geometry
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Background
• Does capacitor location matter?
• How close is close?• What if capacitors can’t
be placed close enough to the IC pwr/gnd pins?
• Is it worth sacrificing routing or using costly new technology in order to get capacitors closer?
• Need a way to facilitate engineering judgment
GND 1.5V 3.3V
decoupling caps
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Location Affects Interconnect Inductance
Interconnect inductance is reduced when the cap is moved closer to the IC ⇒ The capacitor becomes more effective
CjLjRZ
ωω 1++=
LCf res π2
1=
L
C
t=0
v(0)=VS
i(t)
R
)sin()0(LCt
LCVti S=≥
CAP IC
Lbelow
i(t)
P
G
Labove Labove
CAPS
S
L
C
L
C
R
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Equivalent Circuit Model
AB
CD
1L2L
2V1V
k
1IParallel Plate
Geometry
Port 1 Port 2
+ +
decCNI3 3L L ESL′= +
J. Fan, et. al., ``Quantifying SMT decoupling capacitor placement in DC power bus design for multi-layer PCBs,’’ IEEE Transactions on Electromagnetic Compatibility, Vol. 43, No. 4, pp. 588-599, November 2001.
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AB
CD
1L2L
2V1V
k
1IParallel Plate
Geometry
Port 1 Port 2
+ +
decCNI3 3L L ESL′= +
Local Decoupling Effect – Frequency-Domain
' 221
1
VZI
=
2
3
2
3
1132
321
21
21
1
)1(
)()(
LL
LLk
ZLLjMLLj
II
ZZ
N +
+−≈
′++−+==
′ ωω
Equivalent circuit at high frequencies2
21N
VZI
=
' 111
1
VZI
=AB
CD
1L M−
2V1V
1IParallel Plate
Geometry
Port 1 Port 2
+ +
NI
2 3L L M+ −
M
decC
AB
CD
1L M−
2V1V
1IParallel Plate
Geometry
Port 1 Port 2
+ +
NI
2 3L L M+ −
M
01 →decCω
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Local Decoupling Effect – Frequency-Domain
cfΔ
ZΔ
ZΔ
ZΔ
• Increasing series resonant frequency of decoupling capacitor
• Reducing impedance uniformly in a frequency-independent manner (approximately) for frequencies higher than the series resonant frequency
VCC
GND
VCC IC
VCC
VCC GND
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Local Decoupling Effect – Time-Domain
M
L2+L3-M
L1-M
iN(t)
i1(t)
i2(t)dt
tdiMLLdt
tdiM )()()( 232
1 −+≈
Equivalent circuit at early time (t≈0)
)()(32
321 ti
LLMLLti N+
−+=
)(1
)1(
2
3
2
3
ti
LL
LLk
N
+
+−≈1L M−
2V1V
1IParallel Plate
Geometry
Port 1 Port 2
+ +
NI
2 3L L M+ −
M
decC
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Local Decoupling Effect – Time Domain
Reducing initial noise voltage generated in the power bus due to logic transitions.
Time (ns)
Port
2 vo
ltage
(mV
)
VCC
GND
VCC IC
VCC
VCC GND
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Local Decoupling Effect – Charge Delivery
0 1 2 3 4 5 62
2.2
2.4
2.6
2.8
3
3.2
3.4
time [ns]
Vpl
ane
[V]
50 mils400 mils5000 milsNo decap
0 1 2 3 4 5 62
2.2
2.4
2.6
2.8
3
3.2
3.4
time [ns]
Vpl
ane
[V]
50 mils400 mils5000 milsNo decap
35 mil thick 10 mil thick
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Estimating Local Decoupling Effect
02 ln 0.75
2equivRdL Hr
μπ
⎡ ⎤⎛ ⎞= −⎢ ⎥⎜ ⎟
⎝ ⎠⎣ ⎦
ln 0 .75
ln 0 .75
equiv
equiv
Rs r
kR
r
⎛ ⎞−⎜ ⎟+⎝ ⎠≈
⎛ ⎞−⎜ ⎟
⎝ ⎠
4equiva bR +≈For a rectangular power bus,
s – spacing between two vias
d – PWR/GND plane pair spacing r – via radiusRequiv – equivalent radius of power bus
Closed-form expressions derived from the radial transmission-line theory :
assumption: vias are located close to the center of the planes
accounting for fringing effect
⎥⎥⎥⎥
⎦
⎤
⎢⎢⎢⎢
⎣
⎡
+
+−≈Δ
2
3
2
3
1021
1
)1(log20)(||
LL
LLk
dBZ
J. Fan, et. al., “Estimating the noise mitigation effect of local decoupling in printed circuit boards,” IEEE Transactions on Advanced Packaging, Vol. 25, No. 2, pp. 154-165, May 2002.
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220 ln 1 1
2s s
ps ss s
h h l lM hl l h h
μπ
⎧ ⎫⎡ ⎤ ⎛ ⎞⎪ ⎪⎛ ⎞⎢ ⎥= + + + − +⎨ ⎬⎜ ⎟⎜ ⎟⎢ ⎥⎝ ⎠ ⎝ ⎠⎪ ⎪⎣ ⎦⎩ ⎭
220 ln 1 1
2s s
ps ss s
h h r rL hr r h h
μπ
⎧ ⎫⎡ ⎤ ⎛ ⎞⎪ ⎪⎛ ⎞⎢ ⎥= + + + − +⎨ ⎬⎜ ⎟⎜ ⎟⎢ ⎥⎝ ⎠ ⎝ ⎠⎪ ⎪⎣ ⎦⎩ ⎭
( )2via ps psL L M= −
( )
( )
( )
( )
2 2 2 2 2 2
2 2 22 2
2 2 2
2 2 2 2
2 2 272
2 2 2 2
1
2 2 2
2 2 2 2
2 2 32
2 2
2 23
ln
2 23
10 ln
4 tan
2 23
4ln3
t
w p l w p l
l w p ll w p
l w p l
w p w p
w w p lM l w
w w w p l
wlwplp w p l
l p l p
l l p pp ll l p
−
−
⎡⎢− − + + +⎢⎢
⎛ ⎞⎢ + + +⎜ ⎟+ −⎜ ⎟− + + +⎝ ⎠
+ − +
⎛ ⎞+ + +⎜ ⎟= +⎜ ⎟− + + +⎝ ⎠
⎛ ⎞⎜ ⎟− ⋅⎜ ⎟+ +⎝ ⎠
+ − +
⎛ ⎞+ +⎜ ⎟+ +⎜ ⎟− + +⎝ ⎠⎣
⎤⎥⎥⎥⎥
⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎢ ⎥⎦
( )
2 2 2 22 2
7
23/ 22 2 3 3
3 ln 3 ln2 10 3t
l w l w w lw l l ww lL
ww l l w
−
⎡ ⎤⎛ ⎞ ⎛ ⎞+ + + +⎢ ⎥+⎜ ⎟ ⎜ ⎟× ⎜ ⎟ ⎜ ⎟⎢ ⎥= ⎝ ⎠ ⎝ ⎠⎢ ⎥⎢ ⎥− + + +⎣ ⎦
trace t tL L M= −3 via traceL L L′ = + where and
l: separation between two via; hs: height of the capacitor from the nearest power
or ground plane ;w: width of the trace connecting two vias, or width
of the capacitor package if no trace ;r: radius of the via ; and p = 2hs.
Estimating Local Decoupling Effect
C. Wang, et. al., “An efficient approach for power delivery network design with closed-form expressions for parasitic interconnect inductances,” IEEE Transactions on Advanced Packaging, Vol. 29, No. 2, pp. 320-334, May 2006.
Validated with
CEMPIE
cap
pad
via
trace
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ICCAP
GNDPWR
IC
CAP
GNDPWR
Dos Don’ts
Design Implications
J. L Knighten, B. Archambeault, J. Fan, et. al., “PDN Design Strategies: II. Ceramic SMT Decoupling Capacitors – Does Location Matter?,” IEEE EMC Society Newsletter, Issue No. 207, Winter 2006, pp. 56-67.
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Design ImplicationsThe ratio of L3/L2• A good indicator of the ability of a power bus to support local decoupling • The lower, the better• Can be greatly increased by even very short trace length from via to pad
PWR/GND pair
thickness L3' (nH)
L3/L2with no
trace
L3/L2 w/extra 100 mil trace
length
L3/L2 w/extra 200 mil trace
length
L3/L2 w/extra 300 mil trace
length10 mils 1.66 6.75 9.13 11.50 13.8835 mils 0.92 1.29 1.98 2.67 3.36
Tabulated values for • Centered power bus in a 62-mil PCB
stack-up• 10 mil via diameter• 0603 SMT capacitor package
cap
pad
via
trace
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Design Implications
SMT de-cap Placement L3' (nH) L3/L2
Obverse 0.45 2.96875
Reverse 2.56 9.5625
The placement of de-cap (obverse or reverse), can be highly dependent on the position of power bus in PCB stack-up.
Always LEAST above-plane inductance!
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Power/Ground Plane Pair- thin is always better
Outline
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Pi Pj
Impedance of the Power/Ground Plane Pair
dZij ∝)(ωC. Wang, et. al., “An efficient approach for power delivery network design with closed-form expressions for parasitic interconnect inductances,” IEEE Transactions on Advanced Packaging, Vol. 29, No. 2, pp. 320-334, May 2006.
( ) ( ) ( ) ( )⎥⎥⎥
⎦
⎤
⎢⎢⎢
⎣
⎡+
−+=
≠== =∑∑ HM
ijyjyixjxi
nm
M
mjiji
N
n nm
mn
planeij LjLLLLpyyxxf
abkkdj
CjZ ωεεωμ
ωω ,,,,,,1
00 0
22
22
Location Port Dimensions
Higher order mode L
Resonant frequency contributions
Low frequency capacitive term
exists only when i = j
Based on the cavity method:
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Power Bus Thickness Effects on Decoupling
1 10 100 1000 5000-50
-40
-30
-20
-10
0
10
20
30
40
Frequency (MHz)
|Z11
| (dB
Ω)
d = 1 mild = 5 milsd = 10 milsd = 40 mils
1 10 100 1000 5000-80
-60
-40
-20
0
20
40
Frequency (MHz)
|Z21
| (dB
Ω)
d = 1 mild = 5 milsd = 10 milsd = 40 mils
Frequency domain: impedance
a = 12 in
b = 10 in Port 2
Port 1(3 in, 2 in)
(7 in, 4 in)d
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a = 12 in
b = 10 in Port 2
Port 1(3 in, 2 in)
(7 in, 4 in)d
Power Bus Thickness Effects on Decoupling
0 0.5 1 1.5 2 2.5 3-100
0
100
200
300
400
500
Time (ns)
Port
2 vo
ltage
(mV
)
d = 1 mild = 5 milsd = 10 milsd = 40 mils
0 0.5 1 1.5 2 2.5 3-4000
-3000
-2000
-1000
0
1000
2000
3000
4000
Time (ns)
Port
1 vo
ltage
(mV
)
d = 1 mild = 5 milsd = 10 milsd = 40 mils
Time domain: port voltages
0 0.2 0.4 0.6 0.8 10
0.2
0.4
0.6
0.8
1
Time (ns)
Cur
rent
(A)
port 1 current
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0 1 2 3 4 5 62
2.2
2.4
2.6
2.8
3
3.2
3.4
time [ns]
Vpl
ane
[V]
L3 0 .5nHL3 1nHL3 2nHL3 3nHNo decap
0 1 2 3 4 5 62
2.2
2.4
2.6
2.8
3
3.2
3.4
time [ns]
Vpl
ane
[V]
L3 0.5nHL3 1nHL3 2nHL3 3nHNo decap
Power Bus Thickness Effects on Decoupling
35 mil thick 10 mil thick
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Effect of Typical Laminates on Decoupling
Dk Loss Tangent
Thick-ness, d
Comment
4.4 0.02 4, 10, 35 mils
better quality fiberglass/ epoxy resin material
4 0.02 2 mils patented thin FR4 core, widely used.
16 0.006 16 microns (0.63 mils)
ultra-thin material commercially available
20 0.015 0.2 mils ultra-thin material from [7, 8].
10 100 1000 5000-60
-50
-40
-30
-20
-10
0
10
20
30
40
Frequency (MHz)
|Z11
| (dB
Ω)
εr = 4.4, tanδ = 0.02, d = 4 milsεr = 4.4, tanδ = 0.02, d = 10 milsεr = 4.4, tanδ = 0.02, d = 35 milsεr = 4, tanδ = 0.018, d = 2 milsεr = 16, tanδ = 0.006, d = 0.63 milsεr = 20, tanδ = 0.015, d = 0.2 mils
Embedded capacitance (thin laminate) has superior electrical performance
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Summary
Interconnect inductance shall be minimized in any situation: inductance limits the effectiveness of decoupling
Individual capacitor values may not matter: using an array of capacitor values or the largest value in a package size meetsdesign specifications
Capacitor location could be important due to PCB geometry: thin power bus structures (< 10 mils) usually result in a larger L3/L2 value; thus capacitor location is relatively unimportant. Thick power bus structures usually result in a smaller (L3/L2) value, and capacitor placement can be an important design factor
Thin power/ground plane pair is always better: low impedance and high charge availability