precision digital power monitor - farnell element14 · 2015. 12. 2. · vinm vinp 1 3 4 15 a1 a0...

Precision Digital Power MonitorISL28022The ISL28022 is a bi-directional high-side and low-side digital current sense and voltage monitor with serial interface. The device monitors current and voltage and provides the results digitally along with calculated power. The ISL28022 provides tight accuracy of less than 0.3% for both voltage and current monitoring over the entire input range. The digital power monitor has configurable fault thresholds and measurable ADC gain ranges.

The ISL28022 handles common-mode input voltage ranging from 0V to 60V. The wide range permits the device to handle telecom, automotive and industrial applications with minimal external circuitry. Both high and low-side ground sensing applications are easily handled with the flexible architecture.

The ISL28022 consumes an average current of just 700µA and is available in the 10 Ld MSOP package. The ISL28022 is also offered in a space saving 16 Ld QFN package. The part operates over the extended temperature range from -40°C to +125°C.

Features• Bus voltage sense range . . . . . . . . . . . . . . . . . . . . . . 0V to 60V

• 16-bit ∑∆ADC monitors current and voltage

• Voltage measuring error . . . . . . . . . . . . . . . . . . . . . . . . .

ISL28022

Table of ContentsBlock Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

Pin Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Ordering Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

Recommended Operating Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

Typical Performance Curves . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

Functional Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12Detailed Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12Pin Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12Register Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

I2C Serial Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19Protocol Conventions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19SMBus Support . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Device Addressing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Write Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Read Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Broadcast Addressing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21Measurement Stability vs Acquisition Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22Fast Transients. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23External Clock. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23Over-ranging . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24Shunt Resistor Selection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25A Trace as a Sense Resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Typical Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28Point Of Load Power Monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28Adjustable Point Of Load Monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28Power Monitor Boost Regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29Floating Supply DPM (>60V or

ISL28022

Block Diagram

Digital ControlLogic

I2CSM BUS

A0

16

A1

SMBCLK

SMBDAT

OSC

VINM

VINP

VCC

GND

ADC 16-Bit

REG MAP

REF

CLOCK

DIV

SW M

ux

CM = 0 TO 60V

ECLK/INT

VBUS

Pin ConfigurationsISL28022

(10 LD MSOP)TOP VIEW

ISL28022(16 LD QFN)

TOP VIEW

A1

A0

EXT_CLK/INT

SDA/SMBDAT

SCL/SMBCLK

1

2

3

4

5

10

9

8

7

6 VCC

GND

VBUS

VINM

VINP

1

3

4

15

A1

A0

EXT_CLK/INT

SDA/SMBDAT

NC

NC

NC

VIN

P

16 14 13

2

12

10

9

11

65 7 8

VINM

VBUS

GND

VCC

SCL/

SMB

CLK N

C

NC

NC

GND

3 FN8386.2October 10, 2013

ISL28022

Pin DescriptionsMSOP

PIN NUMBER

QFN PIN NUMBER

PIN NAME DESCRIPTION

1 1 A1 I2C address, bit 1

2 2 A0 I2C address, bit 0

3 3 EXT_CLK/INT External ADC clock input or CPU interrupt output signal. When the pin is configured as an interrupt, the output is an open drain

4 4 SDA/SMBDAT I2C serial data input/output

5 5 SCL/SMBCLK I2C clock input

6 9 VCC Positive power pin. The positive power supply to the part

7 10 GND Negative power pin. Can be connected to ground or a negative voltage

8 11 VBUS VBUS power voltage sense

9 12 VINM Current sense minus input

10 13 VINP Current sense plus input

6, 7, 8, 14, 15, 16

NC No connect. No internal connection

Epad GND Negative power pin. Can be connected to ground or a negative voltage

Ordering InformationPART NUMBER(Notes 1, 2, 3)

PART MARKING

PACKAGE (Pb-Free)

PKG. DWG. #

ISL28022FUZ 8022F 10 Ld MSOP M10.118

ISL28022FRZ 022F 16 Ld QFN L16.3x3B

ISL28022EV1Z Evaluation Board

NOTES:

1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.

2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.

3. For Moisture Sensitivity Level (MSL), please see device information page for ISL28022. For more information on MSL please see tech brief TB363.

4 FN8386.2October 10, 2013

http://www.intersil.com/data/tb/tb347.pdfhttp://www.intersil.com/products/ISL28022#packaginghttp://www.intersil.com/data/tb/tb363.pdf

ISL28022

Absolute Maximum Ratings Thermal InformationVCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6.0VVBUS Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .63VCommon Mode Input Voltage (VINP, VINM) . . . . . . . . . . . . . . . . . . . . . . . 63VDifferential Input Voltage (VINP, VINM) . . . . . . . . . . . . . . . . . . . . . . . . . .±63VInput Voltage (Digital Pins) . . . . . . . . . . . . . . . . . . . . . . . . .GND-0.3V to 5.5VOutput Voltage (Digital Pins) . . . . . . . . . . . . . . . . . . . . GND-0.3 to VCC+0.3VOpen Drain Output Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10mAOpen Drain Voltage (Interrupt) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24VESD Rating

Human Body Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8kVMachine Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300VCharged Device Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2.0kV

Latch up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60V @ + 125°C

Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W)16 Ld QFN (Notes 4, 5) . . . . . . . . . . . . . . . . 52 6.510 Ld MSOP (Notes 6, 7) . . . . . . . . . . . . . . . 150 55

Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°CMaximum Junction Temperature (TJMAX) . . . . . . . . . . . . . . . . . . . . .+150°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below

http://www.intersil.com/pbfree/Pb-FreeReflow.asp

Recommended Operating ConditionsAmbient Temperature Range (TA) . . . . . . . . . . . . . . . . . . .-40°C to +125°C

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact productreliability and result in failures not covered by warranty.

NOTES:

4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379.

5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.6. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.7. For θJC, the “case temp” location is taken at the package top center.

Electrical Specifications TA = +25°C, VCC = 3.3, VINP = VBUS = 12V, VSENSE = VINP-VINM = 32mV, unless otherwise specified. All voltages with respect to GND pin.

PARAMETER DESCRIPTION CONDITIONSMIN

(Note 8) TYPMAX

(Note 8) UNIT

INPUTS

VSENSEDIFF Useful Full Scale Current Sense Differential Voltage Range (VINP-VINM)

PGA Gain = /1 0 ±40 mV

PGA Gain = /2 0 ±80 mV

PGA Gain = /4 0 ±160 mV

PGA Gain = /8 0 ±300 mV

VSHUNT_step LSB Step Size, Shunt Voltage 10 µV

VCMSENSE Current Sense Common Mode(VINP, VINM)

0 VBUS V

VOS VSENSE Offset Voltage PGA Gain = /1, /2, /4, /8; ADC Setting = 1111

±10 ±75 µV

VOSTC VSENSE Offset Voltage Temperature Coefficient

0.15 µV/°C

CMRR VSENSE VOS vs Common Mode VBUS = 0 to 60V; BRNG = 2, 3 110 130 dB

PSRR VSENSE VOS vs Power Supply VCC = 3V to 5V 105 dB

ACS Current Sense Gain Error ±40 m%

ACSTC Current Sense Gain Error Temperature Coefficient

±1 m%/°C

IVINACT Input Leakage, VIN Pins Active Mode (for both VINP and VINM pins)

±20 µA

IVINACT Input Leakage, VIN Pins Power Down Mode (for both VINP and VINM pins)

±0.1 ±0.5 µA

5 FN8386.2October 10, 2013

http://www.intersil.com/pbfree/Pb-FreeReflow.asphttp://www.intersil.com/data/tb/tb379.pdfhttp://www.intersil.com/data/tb/tb379.pdf

ISL28022

VBUS Useful Bus Voltage Range BRNG = 0 0 16 V

BRNG = 1 0 32 V

BRNG = 2, 3 0 60 V

VBUS_Step LSB Step Size, Bus Voltage BRNG = 0 4 mV

VBUS_VCO VBUS Voltage Coefficient 50 ppm/V

RVBACT Input Impedance, VBUS Pin Active Mode 600 kΩ

DC ACCURACY

ADC Resolution (Native) PGA gain = /1, VSENSE = ±300mV

16 Bits

Current Measurement Error TA = +25°C ±0.2 ±0.3 %

Current Measurement Error Over Temperature

TA = -40°C to +85°C ±0.5% %

TA = -40°C to +125°C ±1% %

Bus Voltage Measurement Error TA = +25°C ±0.2 ±0.3 %

Bus Voltage Measurement Error Over Temperature

TA = -40°C to +85°C ±0.5% %

TA = -40°C to +125°C ±1% %

ADC TIMING SPECS

ts ADC Conversion Time ADC setting = 0000 72 79.2 µs

ADC setting = 0001 132 145.2 µs

ADC setting = 0010 258 283.8 µs

ADC setting = 0011 508 558.8 µs

ADC setting = 1001 1.01 1.11 ms

ADC setting = 1010 2.01 2.21 ms

ADC setting = 1011 4.01 4.41 ms

ADC setting = 1100 8.01 8.81 ms

ADC setting = 1101 16.01 17.61 ms

ADC setting = 1110 32.01 35.21 ms

ADC setting = 1111 64.01 70.41 ms

I2C INTERFACE SPECIFICATIONS

VIL SDA and SCL Input Buffer LOW Voltage -0.3 0.3 x VCC V

VIH SDA and SCL Input Buffer HIGH Voltage 0.7 x VCC VCC + 0.3 V

Hysteresis SDA and SCL Input Buffer Hysteresis 0.05 x VCC V

VOL SDA Output Buffer LOW Voltage, Sinking 3mA

VCC = 5V, IOL = 3mA 0 0.02 0.4 V

CPIN SDA and SCL Pin Capacitance TA = +25°C, f = 1MHz, VCC = 5V, VIN = 0V, VOUT = 0V

10 pF

fSCL SCL Frequency 400 kHz

Electrical Specifications TA = +25°C, VCC = 3.3, VINP = VBUS = 12V, VSENSE = VINP-VINM = 32mV, unless otherwise specified. All voltages with respect to GND pin. (Continued)


(Note 8) TYPMAX

(Note 8) UNIT

6 FN8386.2October 10, 2013

ISL28022

tIN Pulse Width Suppression Time at SDA and SCL Inputs

Any pulse narrower than the max spec is suppressed.

50 ns

tAA SCL Falling Edge to SDA Output Data Valid

SCL falling edge crossing 30% of VCC, until SDA exits the 30% to 70% of VCC window.

900 ns

tBUF Time the Bus Must be Free Before the Start of a New Transmission

SDA crossing 70% of VCC during a STOP condition, to SDA crossing 70% of VCC during the following START condition.

1300 ns

tLOW Clock LOW Time Measured at the 30% of VCC crossing. 1300 ns

tHIGH Clock HIGH Time Measured at the 70% of VCC crossing. 600 ns

tSU:STA START Condition Setup Time SCL rising edge to SDA falling edge. Both crossing 70% of VCC.

600 ns

tHD:STA START Condition Hold Time From SDA falling edge crossing 30% of VCC to SCL falling edge crossing 70% of VCC.

600 ns

tSU:DAT Input Data Setup Time From SDA exiting the 30% to 70% of VCC window, to SCL rising edge crossing 30% of VCC.

100 ns

tHD:DAT Input Data Hold Time From SCL falling edge crossing 30% of VCC to SDA entering the 30% to 70% of VCC window.

20 900 ns

tSU:STO STOP Condition Setup Time From SCL rising edge crossing 70% of VCC, to SDA rising edge crossing 30% of VCC.

600 ns

tHD:STO STOP Condition Hold Time From SDA rising edge to SCL falling edge. Both crossing 70% of VCC.

600 ns

tDH Output Data Hold Time From SCL falling edge crossing 30% of VCC, until SDA enters the 30% to 70% of VCC window.

0 ns

tR SDA and SCL Rise Time From 30% to 70% of VCC 20 + 0.1 x Cb

300 ns

tF SDA and SCL Fall Time From 70% to 30% of VCC 20 + 0.1 x Cb

300 ns

Cb Capacitive Loading of SDA or SCL Total on-chip and off-chip 75 pF

RPU SDA and SCL Bus Pull-up ResistorOff-chip

Maximum is determined by tR and tF. For Cb = 400pF, max is about 2kΩ~2.5kΩ.For Cb = 40pF, max is about 15kΩ~20kΩ

1 k



(Note 8) TYPMAX

(Note 8) UNIT

7 FN8386.2October 10, 2013

ISL28022

POWER SUPPLY

Operating Supply Voltage Range 3 5.5 V

ICCEXT Power Supply Current On VCC Pin, Active Mode

External power supply mode, VCC = 5V

0.7 1.0 mA

ICCPD Power Supply Current On VCC Pin, Power-Down Mode

External power supply mode, VCC = 5V

5 15 µA

NOTE:8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization

and are not production tested.



(Note 8) TYPMAX

(Note 8) UNIT

Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC =15; unless otherwise specified.

FIGURE 2. VSHUNT VOS FIGURE 3. VSHUNT VOS vs TEMPERATURE

FIGURE 4. VSHUNT MEASUREMENT ERROR FIGURE 5. VSHUNT MEASUREMENT ERROR vs VSHUNT INPUT

0

5

10

15

20

25

-75 -60 -45 -30 -15 0 15 30 45 60 75

HIT

S

VSHUNT VOS (µV)

-0.0750 -0.0625 -0.0500 -0.0375 -0.0250 -0.0125

0 0.0125 0.0250 0.0375 0.0500 0.0625 0.0750

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

V OS

(mV)

VCC = 3.3V

VCC = 3V VCC = 5V

SADC = 15

0

5

10

15

20

25

30

35

40

-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30

HIT

S

VSHUNT MEASUREMENT ERROR (%)

-0.3

-0.2

-0.1

0

0.1

0.2

0.3

-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30 VSHUNT (V)

V SH

UN

T M

EASU

REM

ENT

ERR

OR

(%)

VCC = 5.5V

VCC = 3.3V VCC = 3V

T = +25°CVSHUNT (CMV) = 12V SADC = 15

8 FN8386.2October 10, 2013

ISL28022

FIGURE 6. VSHUNT GAIN vs TEMPERATURE FIGURE 7. VBUS MEASUREMENT ERROR DISTRIBUTION

FIGURE 8. VBUS MEASUREMENT ERROR vs VBUS (TA = +25°C) FIGURE 9. VBUS MEASUREMENT ERROR vs TEMPERATURE

FIGURE 10. CMRR vs TEMPERATURE FIGURE 11. SUPPLY CURRENT vs MODE vs TEMPERATURE

Typical Performance Curves TA = +25°C, VCC = 3.3V, VINP = VBUS = 12V, S(B)ADC =15; unless otherwise specified. (Continued)

-1.0 -0.8 -0.6 -0.4 -0.2

0 0.2 0.4 0.6 0.8 1.0

-50 -25 0 25 50 75 100 125

TEMPERATURE (°C)

GA

IN E

RR

OR

(%)

VCC = 5.5V

VCC = 3.3V

VCC = 3V

VSHUNT (DIFF) = 32mV VSHUNT (CMV) = 12V SADC = 15

0

10

20

30

40

50

60

70

-0.30 -0.25 -0.20 -0.15 -0.10 -0.05 0 0.05 0.10 0.15 0.20 0.25 0.30

HIT

S

VBUS MEASUREMENT ERROR (%)

-1.0 -0.8 -0.6 -0.4 -0.2

0 0.2 0.4 0.6 0.8 1.0

0 8 16 24 32 40 48 56 64 VBUS (V)

V BU

S M

EASU

REM

ENT

ERR

OR

(%)

VCC = 5.5V

VCC = 3.3V VCC = 3V

-1.0 -0.8 -0.6 -0.4 -0.2

0 0.2 0.4 0.6 0.8 1.0

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

V BU

S M

EASU

REM

ENT

ERR

OR

(%)

VCC = 5.5V

VCC = 3.3V VCC = 3V

120

125

130

135

140

145

150

155

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

VCC = 3V VCC = 3.3V

VCC = 5V

VSHUNT (dCMV) = 0V TO 60V SADC = 15

CM

RR

(dB

)

300 350 400 450 500 550 600 650 700 750 800

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

SUPP

LY C

UR

REN

T (µ

A)

MODE = 7MODE = 4

9 FN8386.2October 10, 2013

ISL28022

FIGURE 12. SUPPLY CURRENT vs MODE vs VCC FIGURE 13. SUPPLY CURRENT vs MODE 0 vs TEMPERATURE

FIGURE 14. SUPPLY CURRENT vs MODE 0 vs VCC FIGURE 15. SHUNT IVIN vs TEMPERATURE (MODE 5)

FIGURE 16. SHUNT IVIN vs COMMON MODE VOLTAGE (MODE 5) FIGURE 17. SHUNT IVIN vs TEMPERATURE (MODE 0, 4)


300 350 400 450 500 550 600 650 700 750 800

3.0 3.5 4.0 4.5 5.0 5.5 6.0 VCC (V)

SUPP

LY C

UR

REN

T (µ

A)

MODE = 7

MODE = 4

0 2 4 6 8

10 12 14 16 18 20

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

SUPP

LY C

UR

REN

T (µ

A)

0 2 4 6 8

10 12 14 16 18 20

3.0 3.5 4.0 4.5 5.0 5.5 6.0 VCC (V)

SUPP

LY C

UR

REN

T (µ

A)

5

7

9

11

13

15

17

19

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

I VIN

(µA

)

5 6 7 8 9

10 11 12 13 14 15

0 8 16 24 32 40 48 56 64 Vcm (V)

I VIN

(µA

)

0

0.005

0.010

0.015

0.020

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

I VIN

(µA

)

MODE = 4

MODE = 0

10 FN8386.2October 10, 2013

ISL28022

FIGURE 18. SHUNT IVIN vs COMMON MODE VOLTAGE (MODE 0, 4) FIGURE 19. SHUNT IOS vs TEMPERATURE (MODE 5)

FIGURE 20. SHUNT IOS vs COMMON MODE VOLTAGE (MODE 5) FIGURE 21. SHUNT IOS vs TEMPERATURE (MODE 0, 4)

FIGURE 22. SHUNT IOS vs COMMON MODE VOLTAGE (MODE 0, 4) FIGURE 23. VSHUNT BANDWIDTH vs SADC MODE


0

0.005

0.010

0.015

0.020

0 8 16 24 32 40 48 56 64 Vcm (V)

MODE = 0MODE = 4

I VIN

(µA

)

-0.5 -0.4 -0.3 -0.2 -0.1

0 0.1 0.2 0.3 0.4 0.5

-50 -25 0 25 50 75 100 125

TEMPERATURE (°C)

I OS

(µA

)

-0.5 -0.4 -0.3 -0.2 -0.1

0 0.1 0.2 0.3 0.4 0.5

0 8 16 24 32 40 48 56 64 Vcm (V)

I OS

(µA

)

-0.0020

-0.0015

-0.0010

-0.0005

0

0.0005

0.0010

0.0015

0.0020

-50 -25 0 25 50 75 100 125 TEMPERATURE (°C)

I OS

(µA

)

MODE = 4MODE = 0

-0.0020

-0.0015

-0.0010

-0.0005

0

0.0005

0.0010

0.0015

0.0020

0 8 16 24 32 40 48 56 64 Vcm (V)

I OS

(µA

)

MODE = 4MODE = 0

-50

-40

-30

-20

-10

0

10

10 100 1k 10kFREQUENCY (Hz)

GA

IN (d

B)

SADC = 1

SADC = 0

SADC = 2

SADC = 3

VIN = 200mVP-P SINE WAVE

11 FN8386.2October 10, 2013

ISL28022

Functional DescriptionOverviewThe ISL28022 is a digital power monitor (DPM) device that is capable of measuring bi-directional currents while monitoring the bus voltage.

The DPM requires an external shunt resistor to enable current measurements. The shunt resistor translates the bus current to a voltage. The DPM measures the voltage across the shunt resistors and reports the measured value out digitally via an I2C interface. A register within the DPM is reserved to store the value of the shunt resistor. The stored current sense resistor value allows the DPM to output the current value to an external digital device.

The ISL28022 measures bus voltage and current sequentially. The device has a power measurement functionality that multiplies current and voltage measured values. The power calculation is stored in a unique register. The power measurement allows the user to monitor power to or from the load in addition to current and voltage.

The ISL28022 can monitor supplies from 0 to 60V while operating on a chip supply ranging from 3V to 5.5V.

The ISL28022 ADC sample rate can be configured to an internal oscillator (500kHz) or a user can provide a synchronized clock.

Detailed DescriptionThe ISL28022 consists of a two channel analog front end multiplexer, a 16-bit sigma delta ADC and digital signal processing/serial communication circuitry.

The main block within the device is a 3rd order Sigma Delta ADC. The input signal bandwidth is 1kHz, wide enough for power monitoring applications. The main block includes an internal 1.2V band gap voltage reference that is used to drive the ADC.

The analog front end multiplexer selects the input to the ADC. The selection to the input of the ADC is either a single ended VBUS measurement or a fully differential measurement across a shunt resistor.

The digital block contains controllable registers, I2C serial communication circuitry and a state machine. The state machine controls the behavior of the ADC acquisition, whether the acquisition is triggered or continuous. A more detailed description of the state machine states can be found in “MODE: Operating Mode” on page 14.

Pin Descriptions

A1A1 is the address select pin. A1 is one of two I2C/SMBus slave address select pins that are multi-logic programmable for a total of 16 different address combinations.

There are four selectable levels for A1, VCC, GND, SCL/SMBCLK, and SDA/SMBDAT. See Table 21 for more details in setting the slave address of the device.

A0A0 is the address select pin. A0 is one of two I2C/SMBus slave address select pins that are multi-logic programmable for a total of 16 different address combinations.

There are four selectable levels for A0, VCC, GND, SCL/SMBCLK, and SDA/SMBDAT. See Table 21 for more details in setting the slave address of the device.

EXT_CLK/INTEXT_CLK/INT is the External/Interrupt clock pin. EXT_CLK/INT is a bi-directional pin. The pin provides a connection to the system clock. The system clock is connected to the ADC. The acquisitions rate of the ADC can be varied through the EXT_CLK/INT pin. The pin functionality is set through a control register bit.

When the EXT_CLK/INT pin is configured as an output, the pin functionality becomes an interrupt flag to connecting devices. EXT_CLK/INT pin as an output requires a pull up resistor to a power supply, up to 20V, for proper operation.The internal threshold detectors (OVsh/UVsh/OVb/UVb) signal level relative to the measured value determines the state of the INT pin.

FIGURE 24. VSHUNT BANDWIDTH vs EXTERNAL CLOCK FREQUENCY


-50

-40

-30

-20

-10

0

10

10 100 1k 10kFREQUENCY (Hz)

SADC = 3F_EXTCLK = OFF

SADC = 3F_EXTCLK = 768kHz

SADC = 3F_EXTCLK = 384kHz

VIN = 200mVP-P SINE WAVE

GA

IN (d

B)

12 FN8386.2October 10, 2013

ISL28022

SDA/SMBDATSDA/SMBDAT is the serial data input/output pin. SDA/SMBDAT is a bi-directional pin used to transfer data to and from the device. The pin is an open drain output and may be wired with other open drain/collector outputs. The open drain output requires a pull-up resistor for proper functionality. The pull-up resistor should be connected to VCC of the device.

SCL/SMBCLKSCL/SMBCLK is the serial clock input pin. The SCL/SMBCLK input is responsible for clocking in all data to and from the device.

VCCVCC is the positive supply voltage pin. VCC is an analog power pin. VCC supplies power to the device.

GNDGND is the ground pin. All voltages internal to the chip are referenced to ground. GND should be tied to 0V for single supply applications. For dual supply applications, the pin should be connected to the most negative voltage in the application.

VBUSVBUS is the power bus voltage input pin. The pin should be connected to the desired power supply bus to be monitored.

VINPVINP is the shunt voltage monitor positive input pin. The pin connects to the most positive voltage of the current shunt resistor.

VINMVINM is the shunt voltage monitor negative input pin. The pin connects to the most negative voltage of the current shunt resistor.

Register DescriptionsTable 1 is the register map for the device. The table describes the function of each register and its respective value. The addresses are sequential and the register size is 16 bits (2 bytes) per address.

CONFIGURATION REGISTERThe configuration register (Table 2) controls the functionality of the chip. ADC measurable range, converter acquisition times, converter resolution and state machine modes are configurable bits within this register.

RST: Reset Bit

Configuring the reset bit (bit15) to a 1 generates a system reset that initializes all registers to their default values and performs a system calibration.

BRNG: Bus Voltage Range

Bits 13 and 14 of the configuration register sets the bus measurable voltage range. Table 3 shows the BRNG bit configurations versus the allowable full scale measurement range. The shaded row is the power-up default.

TABLE 1. ISL28022 REGISTER DESCRIPTIONS

REGISTER ADDRESS (HEX) REGISTER NAME FUNCTION

POWER-ON RESET VALUE (HEX) ACCESS

00 Configuration Power On Reset, Bus and Shunt ranges, ADC acquisition times, Mode configuration

799F R/W

01 Shunt Voltage Shunt voltage measurement value 0000 R

02 Bus Voltage Bus voltage measurement value 0000 R

03 Power Power measurement value 0000 R

04 Current Current measurement value 0000 R

05 Calibration Register Register used to enable current and power measurements.

0000 R/W

06 Shunt Voltage Threshold Min/Max Shunt thresholds 7F81 R/W

07 Bus Voltage Threshold Min/Max VBUS thresholds FF00 R/W

08 DCS Interrupt Status Threshold interrupts 0000 R/W

09 Aux Control Register Register to control the interrupts and external clock functionality

0000 R/W

TABLE 2. CONFIGURATION REGISTER

Bit D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0

NAME RST BRNG1 BRNG0 PG1 PG0 BADC3 BADC2 BADC1 BADC0 SADC3 SADC2 SADC1 SADC0 MODE2 MODE1 MODE0

13 FN8386.2October 10, 2013

ISL28022

PG: PGA (Shunt Voltage Only)

Bits 11 and 12 of the configuration register determines the shunt voltage measurement range. Table 4 shows the PGA bit configurations versus the allowable full scale measurement range. The shaded row is the power up default.

BADC: Bus ADC Resolution/Averaging

Bits [10:7] of the configuration register sets the ADC resolution/ averaging when the ADC is configured in the VBUS mode. The ADC can be configured versus bit accuracy. The bit accuracy selections range from 12 to 15-bits. The ADC is configurable versus the number of averages. The selection ranges from 2 to 128 samples. Table 5 shows the breakdown of each BADC setting. The shaded row is the default setting upon power up.

SADC: Shunt ADC Resolution/Averaging

Bits [10:7] of the configuration register sets the ADC resolution/ Averaging when the ADC is configured in the VSHUNT mode. The ADC can be configured versus bit accuracy. The bit accuracy selections range from 12 to 15-bits. The ADC is configurable versus number of averages. The selection ranges from 2 to 128 samples. Table 5 shows the break down of each SADC setting. The shaded row is the default setting upon power-up.

MODE: Operating Mode

Bits [2:0] of the configuration register controls the state machine within the chip. The state machine globally controls the overall functionality of the chip. Table 6 shows the various states the chip can be configured to, as well as the mode bit definitions to achieve a desired state. The shaded row is the default setting upon power-up.

TABLE 3. BRNG BIT SETTINGS

BRNG1 BRNG0USABLE FULL

SCALE RANGE (V)

0 0 16

0 1 32

1 0 60

1 1 60

TABLE 4. PGA BIT SETTINGS

PG1 PG0 GAINRANGE

(mV)

0 0 1 ±40

0 1 ÷2 ±80

1 0 ÷4 ±160

1 1 ÷8 ±300

TABLE 5. ADC SETTINGS, APPLIES TO BOTH SADC AND BADC CONTROL

ADC3 ADC2 ADC1 ADC0 MODE/SAMPLES CONVERSION TIME

0 X 0 0 12-Bit 72µs

0 X 0 1 13-Bit 132µs

0 X 1 0 14-Bit 258µs

0 X 1 1 15-Bit 508µs

1 0 0 0 15-Bit 508µs

1 0 0 1 2 1.01ms

1 0 1 0 4 2.01ms

1 0 1 1 8 4.01ms

1 1 0 0 16 8.01ms

1 1 0 1 32 16.01ms

1 1 1 0 64 32.01ms

1 1 1 1 128 64.01ms

TABLE 6. OPERATING MODE SETTINGS

MODE2 MODE1 MODE0 MODE

0 0 0 Power-Down

0 0 1 Shunt Voltage, Triggered

0 1 0 Bus Voltage, Triggered

0 1 1 Shunt and Bus, Triggered

1 0 0 ADC Off (disabled)

1 0 1 Shunt Voltage, Continuous

1 1 0 Bus Voltage, Continuous

1 1 1 Shunt and Bus, Continuous

14 FN8386.2October 10, 2013

ISL28022

SHUNT VOLTAGE REGISTER 01H (READ-ONLY)The Shunt Voltage Register reports the measured value across the shunt pins (Vinp and Vinm) into the register. The shunt register LSB is independent of PGA range settings. The PGA setting for the shunt register masks the unused most significant bit with a sign bit. For lower range of PGA settings, multiple sign bits are returned by the DPM. Only one sign bit should be used to calculate the measured value.

Tables 7 through 10 show the weights of each bit for various PGA ranges. The tables should be used to calculate the measured value across the shunt pins from the binary to decimal domains.

To calculate the measured decimal value across the shunt, first read the shunt voltage register. Assume the PGA setting is set to the 80mV range. For this example, the reading output by the chip is 1111 1010 0000 0101. The 80mV range has three sign bits.

Only one sign bit needs to be used to calculate the measured decimal value. Bits 14 and 15 are omitted from the calculation. This leaves a binary reading of 11 1010 0000 0101.

Next, multiply each bit by its respective weight. Bit0 value would be multiplied by bit0 weight (1), Bit1 value * Bit1 weight (2), etc....

Add all the multiplied values to equate to a single number. For the binary reading 11 1010 0000 0101 this equates to -1531.

The LSB for a shunt register is 10µV. Multiplying the decimal value by the LSB weight yields the measured voltage across the shunt. A 1111 1010 0000 0101 reading equals -15.31mV measured across the shunt pins.

.

TABLE 7. SHUNT VOLTAGE REGISTER, PG GAIN = /8 (RANGE = 11), FULL SCALE = ±300MV, 15 BITS WIDE


NAME Sign Bit14 Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0

WEIGHT -32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1



NAME Sign Sign Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0

WEIGHT -16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1



NAME Sign Sign Sign Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0

WEIGHT -8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 10. SHUNT VOLTAGE REGISTER, PG GAIN = /1 (RANGE = 00), FULL SCALE = ± 40MV, 12 BITS WIDE


NAME Sign Sign Sign Sign Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0

WEIGHT -4096 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 11. BUS VOLTAGE REGISTER, BRNG = 10 OR 11, FULL SCALE = 60V, 14 BITS WIDE


NAME Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF

WEIGHT 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 12. BUS VOLTAGE REGISTER, BRNG = 01, FULL SCALE = 32V, 13 BITS WIDE


NAME Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF

WEIGHT 4096 2048 1024 512 256 128 64 32 16 8 4 2 1

15 FN8386.2October 10, 2013

ISL28022

BUS VOLTAGE REGISTER 02H (READ-ONLY)The Bus Voltage Register is where the DPM reports the measured value of the Vbus. There are three scale ranges possible depending on the BRNG setting controlled from the configuration register(00H).

Tables 11 through 13 are the weight bits for each BRNG setting. The binary value recorded in the Bus Voltage Register is translated to a decimal value in the same way as the shunt voltage register is converted to a decimal value.

Equation 1 is the mathematical equation for converting the binary Vbus value to a decimal value. N is the bit number. The LSB value for the Vbus measurement equals 4mV across all bus range (BRNG) settings.

CNVR: Conversion Ready (Bit 1)

The Conversion Ready Bit indicates when the ADC has finished a conversion and transferred the reading(s) to the appropriate register(s). The CNVR is only operable when the DPM is set to one of three trigger modes. The CNVR is at a high state when the conversion is in progress. The CNVR transitions and remains at a low state when the conversion is complete.

The CNVR bit is initialized or re-initialized in the following ways;

1. Writing to the configuration register.

2. Reading from Power Register

OVF: Math Overflow Flag (Bit 0)

The Math Overflow Flag (OVF) is a bit that is set to indicate the current or power data being read from the DPM is over ranged and meaningless.

CALIBRATION REGISTER 05H (READ/WRITE)To accurately read the current and power measurements from the chip, the calibration register needs to be programmed.

The calibration register value is calculated as follows:

1. Calculate the full scale current range that is desired. This is calculated using Equation 2. Rshunt is the value of the shunt resistor. VSHUNT is the full scale setting that is desired. In most cases, it is the PGA full scale range (300mV, 160mV, 80mV and 40mV) that the DPM is programmed to.

2. From the current full scale range, the current LSB is calculated using Equation 3. Current full scale is the outcome from Equation 2. ADCres is the resolution of shunt voltage reading. The value is determined by the SADC setting in configuration register. SADC setting equal to 3 and greater will have a 15-bit resolution. The ADCres value equals 2

15 or 32768.

3. From Equation 3, the calibration resister value is calculated using Equation 4. The resolution of the math that is processed internally in the DPM is 4096 or 12 bits of resolution. The VSHUNT LSB is set to 10µV. Equation 4 yields a 16-bit binary number that can be written to the calibration register. The calibration register format is represented in Table 14.

TABLE 13. BUS VOLTAGE REGISTER, BRNG = 00, FULL SCALE = 16V, 12 BITS WIDE


NAME Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0 CNVR OVF

WEIGHT 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 14. CALIBRATION REGISTER, 05h


NAME FS15 FS14 FS13 FS12 FS11 FS10 FS9 FS8 FS7 FS6 FS5 FS4 FS3 FS2 FS1 FS0

Vbus0

15

n

Bitn Bit_Weightn⋅( )∑=

⎡⎢⎢⎣

⎤⎥⎥⎦

Vbus LSB⋅(EQ. 1)

(EQ. 2)Current FS

Vshunt FSR shunt

(EQ. 3)Current LSB

Current FSADC res

(EQ. 4)

CalRegval integerMath res VshuntLSB⋅

CurrentLSB Rshunt⋅( )⎡⎢⎣

⎤⎥⎦

CalRegval integer0.04096

CurrentLSB Rshunt⋅( )⎡⎢⎣

⎤⎥⎦

16 FN8386.2October 10, 2013

ISL28022

CURRENT REGISTER 04H (READ-ONLY)Once the calibration register (05H) is programmed, the output current is calculated using Equation 5:

Bit is the returned value of each bit from the current register either 1 or a 0. The weight of each bit is represented in Table 15. N is the bit number. The current LSB is the value calculated from Equation 3.

POWER REGISTER 03H (READ-ONLY)The Power register only has meaning if the calibration register (05H) is programmed. The units for the power register are in watts. The power is calculated using Equation 6:

Bit is the returned value of each bit from the power register either 1 or a 0. The weight of each bit is represented in Table 16. N is the bit number. The power LSB is calculated from Equation 7:

If Vbus range, BRNG, is set to 60V, the power equation in Equation 6 is multiplied by 2.

THRESHOLD REGISTERSThe shunt voltage or Vbus threshold registers are used to set the Min/Max threshold limits that will be tested versus VSHUNT or Vbus readings. Measurement readings exceeding the respective VSHUNT or VBUS limits, either above or below, will set a register flag and perhaps an external interrupt depending on the

configuration of the interrupt enable bit (INTREN) in register 09H. The testing of the ADC reading versus the respective threshold limits occurs once per ADC conversion.

SHUNT VOLTAGE THRESHOLD REGISTER 06H (READ/WRITE)The VSHUNT minimum and maximum threshold limits are set using one register. The shunt value readings are either positive or negative. D15 and D7 bits of Table 17 are given to represent the sign of the limit. SMX bits represent the upper limit threshold. SMN represents the lower threshold limit. Equation 8 is the calculation used to convert the VSHUNT threshold binary value to decimal. Bit is the value of each bit set in the shunt threshold register. The value is either 1 or a 0. The weight of each bit is

represented in Table 17. N is the bit number. The shunt voltage threshold LSB is 2.56mV.

Current

0

15

n


⎡⎢⎢⎣

⎤⎥⎥⎦

Current LSB⋅

(EQ. 5)

(EQ. 6)

Power

0

15

n


⎡⎢⎢⎣

⎤⎥⎥⎦

Power LSB⋅ 5000⋅

Power LSB Current LSB Vbus LSB⋅ (EQ. 7)

TABLE 15. CURRENT REGISTER, 04h


NAME Bit 15 Bit14 Bit13 Bit12 Bit11 Bit10 Bit9 Bit8 Bit7 Bit6 Bit5 Bit4 Bit3 Bit2 Bit1 Bit0

WEIGHT -32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 16. POWER REGISTER, 03h


NAME PD15 PD14 PD13 PD12 PD11 PD10 PD9 PD8 PD7 PD6 PD5 PD4 PD3 PD2 PD1 PD0

WEIGHT 32768 16384 8192 4096 2048 1024 512 256 128 64 32 16 8 4 2 1

TABLE 17. SHUNT VOLTAGE THRESHOLD REGISTER, 06h


NAME Sign SMX6 SMX5 SMX4 SMX3 SMX2 SMX1 SMX0 Sign SMN6 SMN5 SMN4 SMN3 SMN2 SMN1 SMN0

WEIGHT -128 64 32 16 8 4 2 1 -128 64 32 16 8 4 2 1

Vs thresh0

7

n


⎡⎢⎢⎣

⎤⎥⎥⎦

VsThresh LSB⋅

(EQ. 8)

TABLE 18. BUS VOLTAGE THRESHOLD REGISTER, 07h


NAME BMX7 BMX6 BMX5 BMX4 BMX3 BMX2 BMX1 BMX0 BMN7 BMN6 BMN5 BMN4 BMN3 BMN2 BMN1 BMN0

WEIGHT 128 64 32 16 8 4 2 1 128 64 32 16 8 4 2 1

17 FN8386.2October 10, 2013

ISL28022

BUS VOLTAGE THRESHOLD REGISTER 07H (READ/WRITE)The Vbus minimum and maximum threshold limits are set using one register. The Vbus value readings range from 0 to 60V. Table 18 shows the register configuration and bit weights for the Vbus threshold register. BMX bits represent the upper limit threshold. BMN represents the lower threshold limit. Equation 9 is the calculation used to convert the Vbus threshold binary value to decimal. Bit is the value of each bit set in the Vbus threshold register. The value is either 1 or a 0. The weight of each bit is represented in table 18. N is the bit number. The Vbus voltage threshold LSB is 256mV.

INTERRUPT STATUS REGISTER 08H (READ/WRITE)The interrupt status register consists of a series of bit flags that indicate if an ADC reading has exceeded the readings respective limit. A 1 or high reading from a warning bit indicates the reading has exceeded the limit. To clear a warning, write a 1 or high to the set warning bit. Table 19 shows the definition of the interrupt status register.

BMNW is bus voltage minimum warning. A 1 reading for this bit indicates the bus reading is below the bus voltage minimum threshold limit.

BMXW is bus voltage maximum warning. A 1 reading for this bit indicates the bus reading is above the bus voltage maximum threshold limit.

SMNW is shunt voltage minimum warning. A 1 reading for this bit indicates the shunt reading is below the shunt voltage minimum threshold limit.

SMXW is shunt voltage maximum warning. A 1 reading for this bit indicates the shunt reading is above the shunt voltage maximum threshold limit.

AUX CONTROL REGISTER 09H (READ/WRITE)The aux control register controls the functionality of the EXTCLK/INT pin of the ISL28022. Table 20 shows the definition of the register.

FORCEINTR is the force interrupt bit. Programming a 1 to the bit will force a 0 or a low at the EXTCLK/INT pin.

INTREN is the interrupt enable bit. Programming a 1 to the bit will allow for a threshold measurement violation to set the state of the EXTCLK/INT pin. With the INTREN set, any flag set from the interrupt status register will change the state of the EXTCLK/INT pin from 1 to a 0.

EXCLKEN is the external clock enable bit. Setting the bit enables the external clock. This also changes the EXTCLK/INT pin from an output to an input. The internal oscillator will shut down when the bit is enabled.

EXTCLKDIV are the external clock divider bits. The bits control an internal clock divider that are useful for fast system clocks. The internal clock frequency from pin to chip is represented in Equation 10:

fEXTCLK is the frequency of the signal driven to the EXTCLK/INT pin. EXTCLKDIV is the decimal value of the clock divide bits.

(EQ. 9)

Vb thresh0

7

n


⎡⎢⎢⎣

⎤⎥⎥⎦

VbThresh LSB⋅

TABLE 19. INTERRUPT STATUS REGISTER, 08h


NAME NA NA NA NA NA NA NA NA NA NA NA NA SMXW SMNW BMXW BMNW

WEIGHT 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

TABLE 20. AUX CONTROL REGISTER, 09h


NAME NA NA NA NA NA NA NA FORCEINTR INTREN ExtClkEn ExtCLKDiv[5:0]

WEIGHT 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

freq internalf EXTCLK

EXTCLKDIV 1+( ) 2⋅ (EQ. 10)

18 FN8386.2October 10, 2013

ISL28022

I2C Serial Interface The ISL28022 supports a bi-directional bus oriented protocol. The protocol defines any device that sends data onto the bus as a transmitter and the receiving device as the receiver. The device controlling the transfer is the master and the device being controlled is the slave. The master always initiates data transfers and provides the clock for both transmit and receive operations. Therefore, the ISL28022 operates as a slave device in all applications.

The ISL28022 uses two bytes to transfer all reads and writes. All communication over the I2C interface is conducted by sending the MSByte of each byte of data first, followed by the LSByte.

Protocol ConventionsFor normal operation, data states on the SDA line can change only during SCL LOW periods. SDA state changes during SCL HIGH are reserved for indicating START and STOP conditions (see Figure 25). On power-up of the ISL28022, the SDA pin is in the input mode.

All I2C interface operations must begin with a START condition, which is a HIGH to LOW transition of SDA while SCL is HIGH. The ISL28022 continuously monitors the SDA and SCL lines for the START condition and does not respond to any command until this condition is met (see Figure 25). A START condition is ignored during the power-up sequence.

All I2C interface operations must be terminated by a STOP condition, which is a LOW to HIGH transition of SDA while SCL is HIGH (see Figure 25). A STOP condition at the end of a read operation or at the end of a write operation places the device in its standby mode.

FIGURE 25. VALID DATA CHANGES, START AND STOP CONDITIONS

FIGURE 26. ACKNOWLEDGE RESPONSE FROM RECEIVER

FIGURE 27. BYTE WRITE SEQUENCE (SLAVE ADDRESS INDICATED BY nnnn)

SDA

SCL

START DATA DATA STOPSTABLE CHANGEDATA

STABLE

SDA OUTPUT FROMTRANSMITTER

SDA OUTPUT FROMRECEIVER

81 9

START ACK

SCL FROMMASTER

HIGH

HIGH IMPEDANCE

START

IDENTIFICATIONBYTE

DATABYTE

ACK

SIGNALS FROM THEMASTER

SIGNALS FROMTHE ISL28022

ACK

1 000 n

WRITE

SIGNAL AT SDA 0 0 0 0n n n

ADDRESSBYTE

STOP

DATABYTE

ACK

ACK

19 FN8386.2October 10, 2013

ISL28022

SMBus SupportThe ISL28022 supports SMBus protocol, which is a subset of the global I2C protocol. SMBCLK and SMBDAT have the same pin functionality as the SCL and SDA pins, respectively. The SMBus operates at 100kHz.

Device AddressingFollowing a start condition, the master must output a slave address byte. The 7 MSB’s are the device identifiers. The A0 and A1 pins control the bus address. These bits are shown in Table 21, there are 16 possible combinations depending on the A0/A1 connections.

TABLE 21. I2C SLAVE ADDRESSES

A1 A0 SLAVE ADDRESS

GND GND 1000 000

GND VCC 1000 001

GND SDA 1000 010

GND SCL 1000 011

VCC GND 1000 100

VCC VCC 1000 101

VCC SDA 1000 110

VCC SCL 1000 111

SDA GND 1001 000

SDA VCC 1001 001

SDA SDA 1001 010

SDA SCL 1001 011

SCL GND 1001 100

SCL VCC 1001 101

SCL SDA 1001 110

SCL SCL 1001 111

Broadcast Address 0111 111

20 FN8386.2October 10, 2013

ISL28022

The last bit of the slave address byte defines a read or write operation to be performed. When this R/W bit is a “1”, a read operation is selected. A “0” selects a write operation (refer to Figure 27).

After loading the entire slave address byte from the SDA bus, the ISL28022 compares the loaded value to the internal slave address. Upon a correct compare, the device outputs an acknowledge on the SDA line.

Following the slave byte is a one byte word address. The word address is either supplied by the master device or obtained from an internal counter. On power-up, the internal address counter is set to address 00h, so a current address read starts at address 00h. When required, as part of a random read, the master must supply the one word address byte, as shown in Figure 28.

In a random read operation, the slave byte in the “dummy write” portion must match the slave byte in the “read” section. For a random read of the registers, the slave byte must be “100nnnnx” in both places.

Write OperationA write operation requires a START condition, followed by a valid identification byte, a valid address byte, two data bytes, and a STOP condition. The first data byte contains the LSB of the data, the second contains the MSB. After each of the four bytes, the ISL28022 responds with an ACK. At this time, the I2C interface enters a standby state.

Read OperationA read operation consists of a three byte instruction, followed by two data bytes (see Figure 28). The master initiates the operation issuing the following sequence: A START, the identification byte with the R/W bit set to “0”, an address byte, a second START, and a second identification byte with the R/W bit set to “1”. After each of the three bytes, the ISL28022 responds with an ACK. Then the ISL28022 transmits two data bytes as long as the master responds with an ACK during the SCL cycle following the eighth bit of the first byte. The master terminates the read operation (issuing no ACK then a STOP condition) following the last bit of the second data byte (see Figure 28).

The data bytes are from the memory location indicated by an internal pointer. This pointer’s initial value is determined by the address byte in the read operation instruction, and increments by one during transmission of each pair of data bytes. The highest

valid memory location is 09h, reads of addresses higher than that will not return useful data.

Broadcast AddressingThe DPM has a feature that allows the user to configure the settings of all DPM chips at once. For example, a system has 16 DPM chips connected to an I2C bus. A user can set the range or initiate a data acquisition in one I2C data transaction by using a slave address of 0111 111. The broadcast feature saves time in configuring the DPM as well as measuring signal parameters in time synchronization. The broadcast should not be used for DPM read backs. This will cause all devices connected to the I2C bus to talk to the master simultaneously.

I2C Clock SpeedThe device supports high-speed digital transactions up to 3.4Mbs. To access the high speed I2C feature, a master byte code of 0000 1xxx is attached to the beginning of a standard frequency read/ write I2C protocol. The x in the master byte signifies a do not care state. X can either equal a 0 or a 1. The master byte code should be clocked into the chip at frequencies equal or less than 400kHz. The master code command configures the internal filters of the ISL28022 to permit data bit frequencies greater than 400kHz. Once the master code has been clocked into the device, the protocol for a standard read/ write transaction is followed. The frequency at which the standard protocol is clocked in at can be as great as 3.4MHz. A stop bit at the end of a standard protocol will terminate the high speed transaction mode. Appending another standard protocol serial transaction to the data string without a stop bit, will resume the high speed digital transaction mode. Figure 30 illustrates the data sequence for the high speed mode.

FIGURE 28. READ SEQUENCE (SLAVE ADDRESS SHOWN AS nnnn)

SIGNALSFROM THE

MASTER

SIGNALS FROMTHE SLAVE

SIGNAL ATSDA

START

IDENTIFICATION BYTE WITH

R/W = 0ADDRESS

BYTE

ACK

ACK

0

STOP

1

IDENTIFICATION BYTE WITH

R/W = 1

ACK

START

SECOND READ DATA BYTE

FIRST READ DATA BYTE

ACK

1 00 n n n n 1 00 n n n n

FIGURE 29. SLAVE ADDRESS, WORD ADDRESS, AND DATA BYTES

D15 D14 D13 D10D12 D11 D9 D8

A0A7 A2A4 A3 A1

DATA BYTE 1

A6 A5

1 0 0 nn n R/Wn

WORD ADDRESS

D7 D6 D5 D2D4 D3 D1 D0

SLAVE ADDRESS BYTE

DATA BYTE 2

21 FN8386.2October 10, 2013

ISL28022

Signal IntegrityThe purity of the signal being measured by the ISL28022 is not always ideal. Environmental noise or noise generated from a regulator can degrade the measurement accuracy. The ISL28022 maintains a high CMRR ratio from DC to approximately 10kHz, as shown in Figure 31.

The CMRR vs Frequency graph best represents the response of the ISL28022 when an aberrant signal is applied to the circuit. The normal state of the measured signal is DC and is the state at which the ISL28022 internal calibration is performed.

The graph was generated by shorting the ISL28022 input without any filtering and applying a 0 to 10V triangle wave to the shunt inputs, Vinp and Vinm. The voltage shunt measurement was recorded for each frequency applied to shunt input.

The CMRR of can be improved by designing an filter stage for the of the ISL28022. The purpose of the filter stage is to attenuate the amplitude of the unwanted signal to the noise level of the ISL28022. Figure 32 is a simple filter example to attenuate unwanted signals.

The filter circuit in Figure 32 attenuates unwanted signals. CSH and RSH is single pole RC filter that differentially attenuates unwanted signals to the ISL28022. Most power monitoring applications require a shunt resistor to be low in value to measure large currents. For small shunt resistors, a large value capacitor is required to attenuate low frequency signals. Most large value capacitors are not offered in space saving packages. The corner frequency of the differential filter, CSH and RSH, should be designed for higher value frequency filtering.

R1 and C1 for both inputs are the single ended filter to the ISL28022. The value of the series resistor to the ISL28022 can be a larger value than the shunt resistor, RSH. A larger series resistor to the input allows for a lower cutoff frequency filter design to the ISL28022. The ISL28022 can source up to 20µA of transient current in the measurement mode. The transient or switching offset current can be a large as 10µA. The switching offset current combined with the series resistance, R1, creates an error offset voltage. A balance of the value of R1 and the shunt measurement error should be achieved for this filter design.

The common mode voltage of the shunt input stage ranges from 0V to 60V. The capacitor voltage rating for C1 and CSH should comply with the nominal voltage being applied to the input.

Measurement Stability vs Acquisition TimeThe BADC and SADC bits within the Configuration register configures the conversion time and accuracy for the bus and shunt inputs respectively. The faster the conversion time the less accuracy and more noise introduced into the measurement. Figure 33 is a graph that illustrates the shunt measurement variability versus a set SADC mode. The standard deviation of 2048 shunt VOS measurements is used to quantify the measurement variability of each mode.

FIGURE 30. BYTE TRANSACTION SEQUENCE FOR INITIATING DATA RATES ABOVE 400Kbs

FIGURE 31. CMRR vs FREQUENCY

80 85 90 95

100 105 110 115 120 125 130

10 100 1k 10k 100k 1MFREQUENCY (Hz)

CM

RR

(dB

)

FIGURE 32. SIMPLIFIED FILTER DESIGN TO IMPROVE NOISE PERFORMANCE TO THE ISL28022

LOA

DR

SH

FROM SOURCE

ISL28022

R1

R1

CSH

C1

C1

22 FN8386.2October 10, 2013

ISL28022

Fast Transients An small isolation resistor placed between ISL28022 inputs and the source is recommended. In hot swap or other fast transient events, the amplitude of a signal can exceed the recommended operating voltage of the part due to the line inductance. The isolation resistor creates a low pass filter between the device and the source. The value of the isolation resistor should not be too large. A large value isolation resistor can effect the measurement accuracy. The offset current for shunt input can be as large as 10µA. The value of the isolation resistor combined with the offset current creates an error offset voltage at the shunt input. The input of the Bus channel is connected to the top of a precision resistor divider. The accuracy of the resistor divider determines the gain error of the Bus channel. The input resistance of the Bus channel is 600kΩ. Placing an isolation resistor of the 10Ω will change the gain error of the Bus channel by 0.0016%.

External Clock

An externally controlled clock allows measurements to be synchronized to an event that is time dependent. The event could be application generated, such as timing a current measurement to a charging capacitor in a switch regulator application or the event could be environmental. A voltage or current measurement may be suspectable to crosstalk from a controlled source. Instead of filtering the environmental noise from the measurement, another approach would be to synchronize the

measurement to the source. The variability and accuracy of the measurement will improve.

The ISL28022 has the functionality to allow for synchronization to an external clock. The speed of the external clock combined with the choice of the internal chip frequency division value determines the acquisition times of the ADC. The internal system clock frequency is 500kHz. The internal system clock is also the ADC sampling clock. The acquisition times scale linearly from 500kHz. For example, an external clock frequency of 1.0MHz with a frequency divide setting of 2 results in acquisition times that equals the internal oscillator frequency when enabled. The internal clock frequency of the ISL28022 should not exceed 500kHz. The ADC modulator is optimized for frequencies of 500kHz and below. Operating internal clock frequencies above 500kHz result in measurement accuracy errors due to the modulator not having enough time to settle.

Suppose an external clock frequency of 1.0MHz is applied with a divide by 8 internal frequency setting, the system clock speed is 125kHz or 4x slower than internal system clock. The acquisition times for this example will increase by 4. For a S(B)ADC setting of 3, the ISL28022 will have an acquisition time of 2.032ms instead of 508µs.

The ECLK/INT pin connects to a buffer that drives a D-flip flop. Figure 35 illustrates a simple schematic of the ECLK/INT pin internal connection. The series of divide by 2 configured D-flip flops are control by the CLKDIV bits from the Aux Control Register. The buffer is a Schmitt triggered buffer. The bandwidth of the buffer is 4MHz. Figure 36 shows the bandwidth of the ECLK/INT pin.

FIGURE 33. MEASUREMENT STABILITY vs SADC MODE

FIGURE 34. SIMPLIFIED SCHEMATIC OF THE ISL28022 SYNCHRONIZED TO A PWM SOURCE

0 0.02 0.04 0.06 0.08 0.10 0.12 0.14 0.16 0.18 0.20

0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 SADC MODE

V SH

UN

T V O

S SI

GM

A (m

V)

I2 CSM

BU

S

A1

SCL

SDA

VINP

VINM

GNDR

SHADC

16-BIT

SW M

UX

ECLK

REG MAP

VCC

VBUS

A0

ISL28022 DPM

LOA

D

TO M

CUFUNCTION

GENERATOR

3.3V

+-

VTH

FIGURE 35. SIMPLIFIED INTERNAL BLOCK CONNECTION OF THE ECLK/INT PIN

FIGURE 36. EXTERNAL CLOCK BANDWIDTH vs MEASUREMENT ACCURANCY

ECLK/INT

+1

÷ Fclk_Sys

2X

-1

0

1

2

3

4

5

1 10 EXTERNAL CLOCK FREQUENCY (MHz)

V SH

UN

T M

EASU

REM

ENT

ER

RO

R N

OR

MA

LIZE

D (%

)

CLKDIV = 5 (÷ 12) SADC = 3

23 FN8386.2October 10, 2013

ISL28022

The VSHUNT measurement error degrades at ECLK frequencies above 4MHz. It is recommended that the ECLK does not exceed 4MHz. At ECLK frequencies below 2.5MHz or internal clock frequencies of 208kHz, the clock frequency to modulator is to slow allowing the charged capacitors to discharge due to parasitic leakages. The capacitor discharge results in a measurement error.

Over-ranging It is not recommended to operate the ISL28022 outside the set voltage range. In the event of measuring a shunt voltage beyond the maximum set range (300mV) and lower than the clamp voltage of the protection diode (1V), the measured output reading may be within the accepted range but will be incorrect.

Shunt Resistor Selection In choosing a sense resistor, the following resistor parameters needs to be considered; the resistor value, the resistor temperature coefficient and the resistor power rating.

The sense resistor value is a function of the full scale voltage drop across the shunt resistor and the maximum current measured for the application. The ISL28022 has 4 voltage ranges that are controlled by programming the PGA bits within the configuration register. The PGA bits control the voltage range for the VSHUNT input (VINP-VINM) of the ISL28022. Once the voltage range for the input is chosen and the maximum measurable current is known, the sense resistor value is calculated using Equation 11:

In choosing a sense resistor, the sense resistor power rating should be taken into consideration. The physical size of a sense resistor is proportional to the power rating of the resistor. The maximum power rating for the measurement system is calculated as the Vshunt_range multiplied by the maximum measurable current expected. The power rating equation is represented by Equation 12:

A general rule of thumb is to multiply the power rating calculated in Equation 12 by 2. This allows the sense resistor to survive an event when the current passing through the shunt resistor is greater than the measurable maximum current. The higher the ratio between the power rating of the chosen sense resistor and the calculated power rating of the system (Equation 12), the less the resistor will heat up in high-current applications.

The temperature coefficient (TC) of the sense resistor directly degrades the current measurement accuracy. The surrounding temperature of the sense resistor and the power dissipated by the resistor will cause the sense resistor value to change. The change in resistor temperature with respect to the amount of current that flows through the resistor, is directly proportional to the ratio of the power rating of the resistor versus the power being dissipated. A change in sense resistor temperature results in a change in sense resistor value. Overall, the change in sense

resistor value contributes to the measurement accuracy for the system. The change in a resistor value due to a temperature rise can be calculated using Equation 13:

ΔTemperature is the change in temperature in Celsius. RsenseTC is the temperature coefficient rating for a sense resistor. Rsense is the resistance value of the sense resistor at the initial temperature.

Table 22 is a shunt resistor reference table for select full scale current measurement ranges (ImeasMax). The table also provides the minimum rating for each shunt resistor.

It is often hard to readily purchase shunt resistor values for a desired measurable current range. Either the value of the shunt resistor does not exist or the power rating of the shunt resistor is too low. A means of circumventing the problem is to use two or more shunt resistors in parallel to set the desired current measurement range. For example, an application requires a full scale current of 50A with a maximum voltage drop across the shunt resistor of 40mV. Table 22 shows, this requires a sense resistor of 0.8mΩ, 2W resistor. Assume the power ratings and the shunt resistor values to choose from are 1mΩ/ 1W, 2mΩ /1W and 4mΩ /1W.

(EQ. 11)

R senseV shunt_rangeImeas Max

(EQ. 12)P res_rating Vshunt_range Imeas Max⋅

TABLE 22. SHUNT RESISTOR VALUES AND POWER RATINGS FOR SELECT MEASURABLE CURRENT RANGES

Rsense/Prating VSHUNT RANGE (PGA SETTING)

ImeasMax

(PGA 00)40mV

(PGA 01)80mV

(PGA 10)160mV

(PGA 11)300mV

100µA 400Ω/4µW 800Ω/8µW 1.6kΩ/16uW 3kΩ/30uW

1mA 40Ω/40µW 80Ω/80µW 160Ω/160uW 300Ω/300uW

10mA 4Ω/400µW 8Ω/800µW 16Ω/1.6mW 30Ω/3mW

100mA 400mΩ/4mW

800mΩ/8mW 1.6Ω/16mW 3Ω/30mW

500mA 80mΩ/20mW

160mΩ/40mW 320mΩ/80mW

600mΩ/150mW

1A 40mΩ/40mW

80mΩ/80mW 160mΩ/160mW

300mΩ/300mW

5A 8mΩ/200mW

16mΩ/400mW 32mΩ/800mW

60mΩ/1.5W

10A 4mΩ/400mW

8mΩ/800mW 16mΩ/1.6W 30mΩ/3W

50A 0.8mΩ/2W 1.6mΩ/4W 3.2mΩ/8W 6.0mΩ/15W

100A 0.4mΩ/4W 0.8mΩ/8W 1.6mΩ/16W 3mΩ/30W

500A 0.08mΩ/20W 0.16mΩ/40W 0.32mΩ/80W 0.6mΩ/150W

(EQ. 13)ΔR sense R sense Rsense TC⋅ ΔTemperature⋅

24 FN8386.2October 10, 2013

ISL28022

Let’s use a 1mΩ and a 4mΩ resistor in parallel to create the shunt resistor value of 0.8mΩ. Figure 37 shows an illustration of the shunt resistors in parallel.

The power to each shunt resistor should be calculated before calling a solution complete. The power to each shunt resistor is calculated using Equation 14:

The power dissipated by the 1mΩ resistor is 1.6W. 400mW is dissipated by the 4mΩ resistor. 1.6W exceeds the rating limit of 1W for the 1mΩ sense resistor. Another approach would be to use three shunt resistors in parallel as illustrated in Figure 38.

Using Equation 14, the power dissipated to each shunt resistor yields 0.8W for the 2mΩ shunt resistors and 0.4W for the 4mΩ shunt resistor. All shunt resistor are within the specified power ratings.

LayoutThe layout of a current measuring system is equally important as choosing the correct sense resistor and the correct analog converter. Poor layout techniques can result in severed traces, signal path oscillations, magnetic contamination, which all contribute to poor system performance.

TRACE WIDTHMatching the current carrying density of a copper trace with the maximum current that will pass through is critical in the performance of the system. Neglecting the current carrying capability of a trace will result in a large temperature rise in the trace, and the loss in system efficiency due to the increase in resistance of the copper trace. In extreme cases, the copper trace could be severed because the trace could not pass the

current. The current carrying capability of a trace is calculated using Equation 15:

Imax is the largest current expected to pass through the trace. ΔT is the allowable temperature rise in Celsius when the maximum current passes through the trace. TraceThickness is the thickness of the trace specified to the PCB fabricator in mils. A typical thickness for general current carrying applications (

ISL28022

Orthogonal routing for high current flow traces will result in current crowding, localized heating of the trace and a change in trace resistance (see Figure 40).

The utilization of arcs and 45° traces in routing large current flow traces will maintain uniform current flow throughout the trace. Figure 41 illustrates the routing technique.

CONNECTING SENSE TRACES TO THE CURRENT SENSE RESISTORIdeally, a 4 terminal current sense resistor would be used as the sensing element. Four terminal sensor resistors can be hard to find in specific values and in sizes. Often a two terminal sense resistor is designed into the application.

Sense lines are high impedance by definition. The connection point of a high impedance line reflects the voltage at the intersection of a current bearing trace and a high impedance trace. The high impedance trace should connect at the intersection where the sense resistor meets the landing pad on the PCB. The best place to make current sense line connection is on the inner side of the sense resistor footprint. The illustration of the connection is shown in FIgure 42. Most of the current flow is at the outer edge of the footprint. The current ceases at the point the sense resistor connects to the landing pad. Assume the sense resistor connects at the middle of the each landing pad, this leaves the inner half of the each landing pad with little current flow. With little current flow, the inner half of each landing pad is classified as high impedance and perfect for a sense connection.

Current sense resistors are often smaller than the width of the traces that connect to the footprint. The trace connecting to the footprint is tapered at a 45° angle to control the uniformity of the current flow.

MAGNETIC INTERFERENCEThe magnetic field generated from a trace is directly proportional to the current passing through the trace and the distance from the trace the field is being measured at. Figure 43 illustrates the direction the magnetic field flows versus current flow.

CU

RR

ENT

FLO

W

FIGURE 40. AVOID ROUTING ORTHOGONAL CONNECTIONS FOR TRACES THAT HAVE HIGH CURRENT FLOWS

CURR

ENT F

LOW CURRENT FLOW

FIGURE 41. AVOID ROUTING ORTHOGONAL CONNECTIONS FOR TRACES THAT HAVE HIGH CURRENT FLOWS

SEN

SE R

ESIS

TOR

SENSE TRACE

SENSE TRACELANDING PAD

LANDING PAD

CURRENT BEARING TRACE

CURRENT BEARING TRACE

FIGURE 42. CONNECTING THE SENSE LINES TO A CURRENT SENSE RESISTOR

Bμ o I⋅

2 π⋅ r⋅FIGURE 43. THE CONDUCTOR ON THE LEFT SHOWS THE MAGNETIC

FIELD FLOWING IN A CLOCKWISE DIRECTION FOR CURRENTS FLOWING INTO THE PAGE. CURRENT FLOW OUT OF THE PAGE HAS A COUNTER CLOCKWISE MAGNETIC FLOW

26 FN8386.2October 10, 2013

ISL28022

The Equation in Figure 43 determines the magnetic field, B, the trace generates in relation to the current passing through the trace, I, and the distance the magnetic field is being measured from the conductor, r. The permeability of air, μo, is 4π*10

7 H/m.

When routing high-current traces, avoid routing high impedance traces in parallel with high-current bearing traces. A means of limiting the magnetic interference from high-current traces is to closely route the paths connected to and from the sense resistor. The magnetic fields will cancel outside the two traces and add between the two traces. Figure 44 illustrates a magnetic field insensitive layout.

If possible, do not cross traces with high-current. If a trace crossing cannot be avoided, cross the trace in an orthogonal manor and the furthest layer from the current bearing trace. The inference from the current bearing trace will be limited.

A Trace as a Sense ResistorIn previous sections, the resistance and the current carrying capabilities of a trace were discussed. In high current sense applications, a design may utilize the resistivity of a current sense trace as the sense resistor. This section will discuss how to design a sense resistor from a copper trace.

Suppose an application needs to measure current up to 200A. The design requires the least amount of voltage drop for maximum efficiency. The full scale voltage range of 40mV (PGA 00) is chosen. From Ω law, the sense resistor is calculated to be 200µΩ. The power rating of the resistor is calculated to be 8W. Assume the PCB trace thickness of the board equals 2oz/2.8mils and the maximum temperature rise of the trace is 20°C. Using Equation 15, the calculated trace width is 2.192in. The trace width, thickness and the desired sense resistor value is known. Utilizing Equation 16, the trace length is calculated to be 1.832in.

Figure 45 illustrates a layout example of a current sense resistor defined by a PCB trace. The serpentine pattern of the resistor reduces current crowding as well as limiting the magnetic interference caused by the current flowing through the trace.

The width of the trace in Figure 45 illustration would equal 2.192in and the length between the sense lines equals 1.832in.

The width of the resistor is long for some applications. A means of shortening the trace width is to connect two traces in parallel. For calculation ease, assume the resistive traces are routed on the outside layers of a PCB. Using Equations 15 and 16, the width of the trace is reduced from 2.192in to 1.096in.

When using multiple layers to create a trace resistor, use multiple vias to keep the trace potentials between the two conductors the same. Vias are highly resistive compared to a copper trace. Multiple vias should be employed to lower the voltage drop due to current flowing through resistive vias. Figure 46 illustrates a layout technique for a multiple layered trace sense resistor.

FIGURE 44. CLOSELY ROUTED TRACES THAT CONNECT TO THE SENSE RESISTOR REDUCES THE MAGNETIC INTERFERENCE SOURCED FROM THE CURRENT FLOWING THROUGH THE TRACES.

SENSE RESISTOR

SEN

SE T

RA

CE

SEN

SE T

RA

CE

LAN

DIN

G P

AD

LAN

DIN

G P

AD

CU

RR

ENT

FLO

W

CU

RR

ENT

FLO

W

TO TH

E SEN

SE

RESIS

TOR

FROM THE SENSE

RESISTOR

B to B from− B to B from−B to B from+

TO SENSE CIRCUITRY

FIGURE 45. ILLUSTRATES A LAYOUT EXAMPLE OF A CURRENT SENSE RESISTOR MADE FROM A PCB TRACE.

FIGURE 46. ILLUSTRATES A LAYOUT EXAMPLE OF A MULTIPLE LAYER TRACE RESISTOR.

CU

RR

ENT

FLO

W IN

CU

RR

ENT FLO

W O

UT

SENSE N

EG(-)

SENSE PO

S(+)

THE LENGTH OF THE TRACE BETWEEN THE TWO SENSE

LINES DEFINES THE SENSE RESISTOR

VALUE.

TOP

BOTTOM

TRACE

TRACE

VIA

VIA

TRACEVIA

(A) CROSS SECTION VIEW (B) TOPVIEW

PCB

PCB

27 FN8386.2October 10, 2013

ISL28022

Typical Applications

Point Of Load Power MonitorThe circuit illustrated in Figure 47 is a solution that can be used to monitor a load’s performance. The voltage regulator regulates to a point of load (POL) voltage. 5V, 3.3V, 2.5V and 1.8V are examples of POL voltages.

The main bus voltage applied to the voltage regulator regulates the voltage to the load at the VINM, VBUS and Sense node for the configuration shown in Figure 47. The shunt resistor in the circuit allows the current to be monitored while regulating the voltage to the load. The maximum shunt voltage, the ISL28022 is able to measure is ±300mV. The shunt resistor value is determined by the Equation 17:

CurrentFS is the maximum current to be measured through the load. This is chosen by the user.

The ISL28022 has over/undervoltage (OV/UV) sensing circuitry for the Bus and Shunt inputs. The levels of the error detection circuitry are controlled digitally via an I2C/SMBus interface. The status of each inputs’ error detection can be read digitally via a register. The ISL28022 allows for the summation of error detection bits to be routed to an interrupt pin. For the Point Of Load Monitoring circuit shown in Figure 47, the interrupt pin is connected to the enable pin of the regulator. In a fault condition, the ISL28022 will trigger an interrupt causing the voltage regulator to shut down. When a fault always exist, the ISL28022 interrupt pin output state can be digitally programmed.

The ISL28022 calculates the power and current internally and stores the results in an internal register. The VBUS connected directly to the load, enables a measurement system that monitors power to the load.

Adjustable Point Of Load Monitor Many applications require unique voltages to optimize a circuit’s potential. Figure 48 is a microcontroller selectable point of load (POL) circuit. The circuit is very similar to the POL monitor discussed previously. The general purpose input output bits (GPIO) of the microcontroller controls a multiplexer which connects a gain setting resistor to the adjust (ADJ) pin of the regulator. The feedback resistance (RF), the multiplexer switch resistance and the value of the gain setting resistor (R1 to R8) determine the regulated output voltage (POL V) to the load. Equation 18 is a generic formula to determine the regulated POLV.

The coefficient alpha, α, is dependent on the designed in regulator. For the ISL80101-ADJ, α equals 0.5. Rx is the value of the gain resistor (R1 to R8) selected by the microcontroller. Rmux is multiplexer switch resistor value. The multiplexer switch resistance is a function of the current flowing through the switch. A general practice is to choose resistor values such that current flowing through the multiplexer is small. The ISL84781 has an on- resistance of 0.4Ω.

I2 CSM

BU

SA1

SCL

SDA

VINP

VINMGND

RSH ADC

16-Bit

SW M

UX

VOUT

EN

ECLK/INT

VRAIL = 0 TO 60V

REG MAP

VCC

VBUS

A0

ISL28022 DPM

SENSE

VOUT

VOLTAGE REGULATOR

LOA

D

TO µ

CFIGURE 47. POINT OF LOAD MONITORING DESIGN IDEA

(EQ. 17)R shunt RSH

0.30Current FS

FIGURE 48. ADJUSTABLE POINT OF LOAD MONITORING DESIGN IDEA

I2CSMBUS

A1

SCL

SDA

VINP

VINM

GND

RSH ADC 16-Bit

SW M

UX

VOUT

EN

VRAIL = 0 TO 60V

REG MAP

VCC

VBUS A0

ISL28022 DPM

ADJ

ADJUSTABLEVOLTAGE

REGULATOR

LOA

D

ISL8

4781

A0

A1

A2

R1

R2 R3

R4 R5 R6

R7

R8

C

RF

SCL

SDA

VCC

GNDVCC

ECLK/INT

GPIOGPIOGPIOGPIO

POL V

To ECLK/INT

VCC

μ

(EQ. 18)POL_V αR F

R x R mux+1+

⎛⎜⎝

⎞⎟⎠

⋅

28 FN8386.2October 10, 2013

ISL28022

Power Monitor Boost RegulationThe Power Monitor Boost Regular application is an example of the ISL28022 used as a digital helper, (Figure 49). With minimal circuitry, the ISL28022 enables smart designs that digitally monitor the electrical parameters to a load. Alternative designs require a current amplifier paired with an ADC. The ADC chosen is often not compliant to common communication standards, such as I2C. The ISL28022 solves this problem and allows for 16 devices on a single I2C bus.

The ISL97516 chip is a high efficiency step-up voltage regulator. The maximum peak inductor current the regulator can deliver is 2.0A. If more output current is needed, the ISL97656 is rated for 4.0A maximum peak inductor current. For this particular application, the ISL97516 is configured to step up the voltage at the VDD pin to 12V. The voltage at VDD can range from 2.3V to 5.5V for normal 12V regulated operation. A USB power pin could be used to drive the ISL97516.

The regulation node of the circuit, shown in Figure 49 is at VOUT. The ISL97516 has feedback circuitry that removes the current sense resistor, RSH, from impacting the regulation voltage. The current sense resistor is calculated using Equation 17. Equation 17 on page 28 shows the formula used to calculate Rshunt.

The ISL28022 interrupt pin is connected to the Enable pin of theregulator. The ISL28022 has OV/UV alerts for both the Bus andShunt channels. A fault condition from either channel powersdown the voltage regulator.

Floating Supply DPM (>60V or

ISL28022

Force Voltage Measure Current and VoltageMany applications require testing of components or validating a system’s performance by utilizing circuits that force voltage and measure current and voltage to a specific connection point. The circuit in FIgure 51 is a force voltage measure current and voltage circuit.

The digital-to-analog converter, DAC, is configured by the microcontroller. The DAC is either integrated into the microcontroller or a standalone discrete device depending on the level of precision needed. The output of the DAC is fed into the non-inverting input of the ISL28534. The ISL28534 is a chopper-stabilized single output instrumentation amplifier with an additional operational amplifier (op amp) integrated into the die. The allowable voltages that can be fed to the op amp are 0.1V to V+ - 0.1V. The allowable supply voltages to the circuit ranges from 3.1 to 5.5V. The integrated op amp regulates the voltage to the load while delivering current. The op amp can successfully regulate the programmed voltage up to 1mA. For greater than 1mA drive currents, an external operational amplifier with higher current may be needed. Another approach is to add a push-pull output stage between the output of the integrated op amp and the sense resistor. The push-pull output stage that enables higher drive capability for the circuit is shown in Figure 52.

The addition of the power MOSFETS between the shunt resistor, RSH, and the load increases the drive current to the load. The MOSFETS are labeled M1 and M2 in Figure 52.

The feedback loop of the shunt resistor, RSH, and the power MOSFETS, M1 and M2, between the IN- terminal and Out terminal of the op amp, enhances the drive capability to the load. The IN-terminal regulates the voltage to the load, while the current is steered from 3.1V to 5V supply though the MOSFET, M1, to the load. The additional power stage limits the voltage delivered to the load.

Integrated into the ISL28534 is an instrumentation amplifier, IA, that can be configured to 1 of 9 gain ranges. The ISL28534 is one product of a family of six products. The products are differentiated by gain values and the number of outputs. The purpose of the instrumentation amplifier, IA, is to extend the measurable current range by using one sense resistor.

The ISL28534 gain selection spans from 1 to 1000. The voltage noise floor of the IA is 0.5µV at a high gain. The lowest voltage the IA can resolve is 1µV as a conservative value. The ISL28534 has an input offset current of 300pA at room temperature. The lowest current that can be measured is 500pA. The current value is dependent on the operational temperature range of the circuit.

The output of the IA connects to the shunt input (VINP, VINM) of the ISL28022. The full scale measurable range of the shunt input is ±300mV. Voltage readings that exceed 300mV in magnitude, require a reduction of gain in the IA setting. The ISL28022 digital comparators can be set to fire an interrupt for readings above 300mV. The interrupt can be routed to either the interrupt or a GPIO pin of the microcontroller. The connection allows the ISL28022 to notify the microcontroller to decrement the gain of the IA when the output reading of the IA exceeds 300mV.

FIGURE 51. A SIMPLIFIED SCHEMATIC OF A FORCE VOLTAGE MEASURE V

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