Vector Near-Field Measurement System Using an Electro-OpticMicrocavity and Electrical Downconversion

Dong-Joon Lee!, Jeong-Jin Kang2, Chia-Chu Chenl and John F. Whitakerl

lCenter for Ultrafast Optical Science and Department ofElectrical Engineering and Computer ScienceUniversity ofMichigan, 2200 Bonisteel Boulevard, Ann Arbor, Michigan 48109-2099, USA

2 Department of Information and CommunicationDong Seoul College, 423 Bokjeong-Dong, Sujeong-Gu, Seongnam, Gyunggi 461-714, Korea

Abstract - An electro-optic (EO) field-mapping system thatfeatures a continuous-wave laser-diode optical source, an entirelyfiber-coupled beam path, a resonance-assisted EO-microcavityprobe, and an RF-downconversion mixing circuit is shown to beeffective for extracting near-field vector RF information. Thesystem is the first of its kind to allow amplitude and phaseanalysis of signals interrogated with a continuous optical beamand no polarization components in the EO-modulation section. Acomplete tangential-electric-field characterization in the nearfield of an RFID antenna - a small planar loop intended forapplications in mobile-reader instruments - is presented, with a>35 dB signal-to-noise ratio attained.

Index Terms - Dipole antennas, distributed feedback lasers,electrooptic measurements, electrooptic modulation, Fabry-Perotresonators.

I. INTRODUCTION

In the design and analysis of microwave devices, it is oftendesirable, if not necessary, to have detailed information on thecharacteristics and potential effects of the localelectromagnetic radiation. While the far-field pattern is ofprimary concern due to the fact that the distance from theantenna to the observer is typically much greater than thelargest dimension of the transmitting antenna, in the near-fieldregion measurements are still useful in testing forelectromagnetic compatibility and interference (EMI),measuring specific absorption rate (SAR), creating far-fieldpatterns from a compact range environment, and diagnosingfailures in microwave integrated circuits [1,2].

One technique for nonintrusively pursuing near-fieldmeasurements of microwave devices and radiation sources,even down to their aperture plane, is that of electro-opticsensing (EOS) [3]. The outstanding spatial resolution (--5 Jlm)and scanning mobility of electro-optic (EO) field-mappinghave been demonstrated by several groups using optical-fiber­mounted, EO crystals as electric-field sensors [2, 4]. However,despite its advantages, EOS systems have relatively lowsensitivity and often require expensive pulsed or intensitymodulated lasers for high-speed sensing [1-6]. In addition,systems based on electro-optic intensity modulation require theuse of crossed polarizers and a quarter-wave retarder to createa modulation function that is linear with an applied RF field.

978-1-4244-1780-3/08/$25.00 © 2008 IEEE

While it is true that continuous-wave (cw) laser diodes havepreviously been employed to mitigate some ofthese issues withEOS, cw-laser-based systems have suffered from acorresponding loss in phase-measurement capability. Inaddition, even though resonant EO probes have also beendemonstrated, they have not taken advantage of a substantialsimplification in polarization control that can be predicted[6,7].

In this work, a variety of steps have been taken tosignificantly reduce the complexity of an EOS system whileboth maintaining a vector-signal-measurement capability andenhancing the electric-field sensitivity of the EOS technique.This has been accomplished first by using a modulation slope(i.e., modulation efficiency) that is controlled by tuning thewavelength of a diode-laser incident on an EO microcavity.Three optical-polarization elements were also eliminated fromthe optical beam path through use of this geometry. Secondly,the reflective, resonance-based system was operated at awavelength such that less noise and an enhanced EOmodulation signal were both realized compared to aconventional, single-pass EOS system. Finally, the EO­modulated signals detected by a photodetector were down­converted using a microwave mixer for sensing within thebandwidth of an RF lock-in amplifier. A 900-MHz mobileRFIO antenna was then used as an example device-under-test(OUT) to illustrate near-field measurements from the newEOS system.

II. PRINCIPLE AND EXPERIMENTAL SETUP

The electro-optic (EO) sensing mechanism relies on thePockels effect to modify the refractive indices of an EO crystal- and thus also the polarization of a probing light beam - withan applied electric field. In the conventional embodiment of anEOS field-measurement system, the probe light beam passesthrough the EO medium, situated between a pair of crossedpolarizers, and thus the light intensity varies with the strengthof the applied field. This polarizer/analyzer pair creates an EOefficiency slope for the modulator that follows the derivative ofthe function sin2(r/2), where r is the phase retardationinduced by the modulating field. A quarter waveplate shifts the

1589

Authorized licensed use limited to: University Town Library of Shenzhen. Downloaded on April 21,2010 at 05:49:03 UTC from IEEE Xplore. Restrictions apply.

operating point to an essentially linear operating regime at50% transmission [8].

Alternatively, it has been suggested that an EO-modulationslope can be achieved in a transmission modulator exclusivelythrough utilization of the Fabry-Perot effect in a thin EOmedium, i.e., without the need for the polarizer/analyzer pairor the quarter-waveplate [6,7]. We now use this concept topredict and then demonstrate such a microcavity, resonance­based system in an operational mode that exhibits greatercompatibility with practical near-field measurements, i.e., in areflection geometry, and specifically in an entirely fiber­coupled reflection geometry. As in transmission, the use of anoptical microcavity in reflection yields an EO-efficiency slopethat can be adjusted using the wavelength tunability of a laserdiode, while also enhancing the EO phase retardation byholding the photons in the EO medium for a longer interactiontime with the applied RF field [6,9]. As a result, the EOsignal, which is a product of the EO slope and field-inducedphase retardation, can reach a strength that exceeds that ofconventional, single-round-trip EOS.

The experimental schematic of the reflective, resonance­based EOS system is shown in Fig. 1. Based on computationsof the reflected spectrum from a Fabry-Perot etalon, the EOmedium, consisting of an x-cut, ---52-Jlm-thick LiTa03 plate,had a five-layer high-reflection (HR) coating deposited on itstop and bottom faces. A glass ferrule was used to mate theprobe tip with a single-mode optical fiber. A fiber-pigtailed,distributed-feedback (DFB) laser, manual polarizationcontroller, optical circulator, and fiber-pigtailed photodiodewere then spliced together with the probe to form acontinuous, low-loss, enclosed optical path.

a spectrum analyzer. Therefore, to also capture the DUT­signal phase, we have utilized an electronic mixing techniqueto downconvert the high frequency signal information into alow-frequency IF appropriate for the bandwidth of a lock-inamplifier. The measured amplitude and phase at the IF thendirectly correspond to those of the signal frequency from theOUT.

A qualitative description of the Fabry-Perot-based amplitudemodulation that arises from the EO-induced spectralperturbation on the static spectral bias is presented in Fig. 2.When the optical phase bias (i.e., the laser wavelength) is setalong the reflection-spectrum slope (by temperature-tuning thelaser diode), the reflected beam is modulated proportionallywith the slope due to the perturbation of the bias by an appliedRF field. It should be noted that, in EO sensing systems, thesignal-to-noise (SNR) ratio is more important than theabsolute size of the signal modulation. The signal modulationcan be increased at the highest slope of the reflectionspectrum, but the corresponding noise level is also typicallylarger when the reflected optical intensity onto the photodiodeis higher. Hence, for high SNR, the spectral bias is optimumwhen it is close to the destructive wavelength Ad in Fig. 2.

In our experiment, ---30 mW of optical power was outputfrom the DFB laser source. The laser wavelength was set at Aeo

(Fig. 2), where the reflected power is ---10% of the input beam,to maximize the modulation without exceeding the 3-mWsaturation level of the I-GHz-bandwidth photodiode. At thisoptical-bias point, compared to a conventional double-pass EOsensor utilizing the same crystal, an enhancement in thesystem SNR of~10 dB has been realized.

normalized

Fig. 2. Normalized EO-probe reflectance vs. tunable laser wavelength(solid line: experimental reflection; dashed line: fitted reflection withan equivalent Fresnel reflection coefficient of r = 0.81). A sinusoidalRF field modulates the refractive indices of the EO medium andhence also the optical phase delay in the microcavity. Analogous to asinusoidal variation in the wavelength, this effect of the RF fieldcauses a corresponding sinusoidal reflected-intensity modulation.

Fig.1. Experimental setup of the entirely fiber-coupled, resonance­based EO sensing system. (DFB LD: distributed feedback, cw laserdiode; PC: polarization controller; OSA: optical spectrum analyzer).The gray and black lines are optical fibers and electrical connections,respectively.

1 .0

0.8

0.6

0.4

0.2

0.0

r = 0.81

-......1-...................== amplitude modulated beam

1563.0 1563.5 (om)

As the frequency radiated by the DUT directly modulated theprobe optical beam in this case, the photodiode must havesufficient bandwidth to resolve the RF or microwave signal.While this is not a serious limitation up to and beyond evenKa-band, until now only scalar measurements have beenperformed by observing the detected RF-signal amplitude with

978-1-4244-1780-3/08/$25.00 © 2008 IEEE

III. RESULTS AND DISCUSSION

To demonstrate the operation of the new EOS system, wehave conducted near-field scans on an RFID antenna designedfor use in a mobile handheld scanner at 900 MHz. The RFIDDUT (Fig. 3(a)) is basically a folded ~/2 dipole antenna that istransformed to a rectangular structure so its size may be

1590

Authorized licensed use limited to: University Town Library of Shenzhen. Downloaded on April 21,2010 at 05:49:03 UTC from IEEE Xplore. Restrictions apply.

1

o

30

o

(mm)

-1[

o 10 20 30 (mm) (= n)--------...·x

20

30

20

y10

1

o

(a)

(b)

(b)

(a)

(mm)

Fig. 4. Transverse electric near-field distributions over an exampleRFID antenna (normalized amplitude), (a) x-component; (b) y­component.

Fig.5. Electric-field phase distributions over the RFID antenna. (topview). (a) Phase of transverse, x component; (b) phase of transverse,y component.

30

The phase information from the RFID-antenna transverse­field components, which can provide valuable information onpropagation directions and radiated polarizations, is shown inFig. 5, in 2-D plots. Two predominant phase angles areobserved in this data, where the green and red componentsindicate a 1800 phase shift. In the two dipole gaps on the plotof the x component of electric field, the signals are seen to bein phase, and thus the polarization that will combine in the farfield is identified. The energy in each part of the y component,which is seen to be of high amplitude only at the edges of themicrostrip lines, is always out of phase with that ofneighboring parts of the y component, indicating that thispolarization will destructively combine in the far field.

.' "T""

-----5

-20

-25 (b)600 700 800 900 1000 (MHz)

-15

-10

return loss(cS)o

Fig. 3. (a) Photograph of capacitively-Ioaded RFID antenna, wherethe shaded trace is a tuning patch on the bottom of the substrate. (b)Return loss measured via network analyzer (-22 dB at 900 MHz).

The -48 dBm of peak signal level was obtained fromelectronic down-conversion through a microwave mixer. A900 MHz RF signal at -41 dBm, demodulated by thephotodiode and measured on a spectrum analyzer, isdownconverted to a I-MHz IF using 0 dBm ofa 901-MHz LO.Although the mixer has ",7dB conversion loss, it does notimpact the overall SNR and the absolute value of the signalcan be changed with the LO power level.

The essential benefit of the signal down-conversion is toobtain the amplitude and phase information of the RF signalsimultaneously. This is possible by utilizing a lock-in amplifierand an IF that falls within its bandwidth. One can also exploitthe narrow passband afforded by lock-in detection to isolatethe signal at the IF despite the wide bandwidth of thephotodiode and its amplifier, the EO sensor, and so on.

minimized. The RFID antenna adopted the T-match topologyto adjust the input impedance of a planar dipole antenna. Acapacitively-Ioaded transmission line added in the middle ofthe antenna reduces the resonant frequency by providing extraelectrical length [10]. In addition, the size and shape of aparasitic patch on the back of the antenna substrate wasmanipulated to tune the resonant frequency to 900 MHz, asseen if Fig. 3(b). The fiber-based EO probe, positioned on anx-Y, computer-controlled translation stage as shown in Fig. 1,scanned the DUT at a separation of ",300 Ilm, with a 200-llmstep resolution. The scanning area is 38x30 mm2

•

The RFID DUT consists of two major dipole antennas. Theprimary one is the folded dipole, at the top of the structure inFig. 3(a), and the secondary one is the capacitive loading patchin the middle of the substrate. With a 20-dBm input power,comparable peak signal levels of -48 dBm ± 3dB are observedon the lock-in amplifier at the two terminal ports for both the xand y components of the electric field (depending on thepolarization of the tangential component and slight changes ofthe probe height or the spectral bias). This is the electricalpower in the demodulated, amplified, and downconvertedsignal channel. An oblique view of the normalized electric­field amplitude is shown in Fig. 4 for both transverse fieldcomponents. The noise level is -85 dBm ± 5dB, depending onthe EMI environment around the experiment. The SNR is thusin excess of 35 dB, which is sufficient for both field mappingand device characterization, and it can be increased bylowering the noise floor with improved shielding methods.

978-1-4244-1780-3/08/$25.00 © 2008 IEEE 1591

Authorized licensed use limited to: University Town Library of Shenzhen. Downloaded on April 21,2010 at 05:49:03 UTC from IEEE Xplore. Restrictions apply.

The amplitude and phase analysis over the entire antennaalso provides a detailed characterization of the antenna design.For instance, the 22-dB return loss of the antenna based onnetwork analyzer measurements indicates the matching quality,but does not provide genuine field flows. When the antennaoperates at the resonance frequency of 900 MHz with 22-dBreturn loss, the strongest fields are expected on the twoterminal ports at the top and center of the OUT, as observed.The fields at the ports should also diminish rapidly as theypropagate away from the terminals, as also seen in Fig. 4. Thestrongest fields are at the end of the horizontal lines (x­direction), and the fields drop rapidly and significantly as theymove towards the feed side via the vertical lines (- y-direction).Therefore, the amplitude and phase analyses of the antennausing this near-field-mapping system are suitable forscrutinizing the matching and the design of the OUT.

A simulation based on the finite-element method has alsobeen used to provide two orthogonal radiation components foran ideal version of the RFID antenna at resonance.Concentrating on the most critical parts of the structure basedon the full-antenna scans, the computational amplitude of thetangential electric fields for the primary and capacitively­loaded terminals are presented in Fig. 6. Detailedexperimental comparisons follow in Figs. 7 and 8, respectively.

As one observes, there is excellent agreement between thesimulated and measured fields for both of the transversecomponents in these dipole regions. The strong x-directionfields at the two terminals yield evidence of efficient radiationdue to good impedance-matching on-resonance. To expand theutility of the near-field measurements, the directionality andpattern of the near-field radiation components may also beextracted. The experimental calculations of the near fields andtheir evolution to the far field will be discussed in future work.

Fig.7. Transverse electric-field distributions over the primary antennatenninal. (nonnalized so that 0 dB = - 48 dBm photodetector output).(a) Photograph; (b) x-amplitude; (c) y-amplitude.

[1] K. Yang, J.G. Yook, L.P.B. Katehi, and J.F. Whitaker, "Electrooptic Mappingand Finite-Element Modeling of the Near -Field Pattern of a Microstrip PatchAntenna," IEEE Trans. Microwave Theory Tech., vol.48, pp. 228-294, Feb. 2000.[2] H. Togo, N. Shimizu, and T. Nagatsuma, "Near-Field Mapping System UsingFiber-Based Electro-Optic Probe for Specific Absorption Rate Measurement," IEICETrans. Electron., vol.E90-e, pp. 436-442, Feb. 2007.[3] K. Yang, T. Marshall, M. Fonnan, J. Hubert, L. Mirth, Z. Popovic, L.P.B. Katehi,and J.F. Whitaker, "Active-amplifier-array diagnostics using high-resolutionelectrooptic field mapping, "IEEE Trans. Microwave Theory Tech., vol.49, pp.849­857, May. 2001.[4] K. Yang, L.P.B. Katehi, and J.F. Whitaker, "Electric-field mapping system usingan optical-fiber-based electro-optic probe," IEEE Microwave Wireless Compo Lett.,vol. 11, pp. 164-166, Apr. 2001.[5] W.K. Kuo, C.H. Pai, H.Y. Chou, and S.L. Huang, "Electro-optic mapping systemsof electric-field using CW laser diodes," Optics and Laser Tech., vol.38, pp. 111­116,2006.[6] D.1. Lee, M.H. Crites, and J.F. Whitaker, "Reflection-Mode Electro-OpticSensing of Microwave Fields with a Wavelength-Tunable Modulation Depth,"Journal ofLightwave Technology, submitted, Jan. 2008.[7] D.1. Lee, and J.F. Whitaker, "A Simplified Fabry-Perot Electro-Optic-ModulationSensor." IEEE Photo Tech. Lett. accepted for publication, Feb. 2008.[8] A.Yariv, and P.Yeh, Optical Waves in Crystals. Ch.8. John Wiley andSons.1984.[9] A.1. Vickers, R. Tesser, R. Dudley, and M.A. Hassan, "Fabry-Perot enhancementelectro-optic sampling," Opt. Quantum Electron. vol.29, pp. 661-669, Mar.l997.[10] J.1. Kang, D.1. Lee, C.C. Chen J.F. Whitaker, and E.1. Rothwell, "CompactMobile RFID Antenna Design and Analysis Using Photonic-assisted Vector Near­field Characterization," to be published in the digest of the IEEE InternationalConference on RFID, Apr. 2008.

IV. CONCLUSION

In conclusion, a simplified electro-optic sensing system thatis suitable for scrutinizing near fields to evaluate theperformance of microwave devices has been developed. Usinga low-cost, cw, fiber-pigtailed laser diode and an all fiber­guided embodiment, a resonant-microcavity-based probeprovides a reliable probe, eliminating the need for multiplepolarization components. Additional microwave down­conversion provides full vector near-field mapping capabilities.

ACKNOWLEGMENT

This work was supported through a subcontract from Opteos,Inc., under the AFOSR STTR program, contract numberFA9550-04-C-0079.

REFERENCES

- -60

- -30

- -40

- -SO

- -20

- -10

(c)

(d)

(b)

(c)

(a)

(b)

(a)

(c)

(b)

(a)

Fig.8. Transverse electric-field distributions over the capacitively­loaded tenninal (normalized 0 dB = - 48 dBm photodetector output).(a) Photograph; (b) x-amplitude; (c) y-amplitude.

Fig. 6. Simulations of the transverse electric-field distributions overthe antenna terminals (arbitrary scale, peak: red; baseline: blue). (a)Primary port, x-amplitude; (b) primary port, y-amplitude; (c)secondary port, x-amplitude; (d) secondary port, y-amplitude.

(dB)o

978-1-4244-1780-3/08/$25.00 © 2008 IEEE 1592

Authorized licensed use limited to: University Town Library of Shenzhen. Downloaded on April 21,2010 at 05:49:03 UTC from IEEE Xplore. Restrictions apply.