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CRLH Metamaterial Leaky-Wave andResonant Antennas

Christophe Calm'1, Tatsuo Itoh2, and Andre RenningS3

1tcole Polytechnique de Montrealch. de Polytechnique, H3T IJ4, QC, Montr~al

E-mail: chdstophe.caloz~polymtl.ca2 University of California Los Angeles

405 Hilgard Aye, Los Angeles, CA 90095, USA3 General and Theoretical Electrical Engineering (ATE), Faculty of Engineering

University of Duisburg-Essen, 47048 Duisburg, Germany

Abstract

Composite right-/left-handed (CRLH) transmission-line (TL) metamnaterials, with their rich dispersion and fundamental right-/left-hand duality, represent a paradigm shift in electromagnetics engineering and, in particular, for antennas. This paperpresents an overview of the most practical leaky-wave and resonant CRLH antennas, which all exhibit functionalities or/andperformance superior to prior state of the art. The leaky-wave antennas provide full-space dynamic scanning capability, withfan beams, conical beams in uni-planar configurations, pencil beams without any complex feeding network, and activelyshaped beams based on the concept of aperture digitization. The resonant antennas offer alternative properties and asolution to beam-squinting when no scanning is required, including multi-band (dual/tri-band) operation, zeroth-order highefficiency, high directivity, and planar electric and magnetic monopole radiators.

Keywords: Metamaterials; composite right-/left-handed; leaky wave antennas; beam steering; shaped beam antennas;multifrequency antennas; directive antennas; monopole antennas

1. Introduction

M etamaterials are broadly defined as effectively homogeneousartificial structures exhibiting unusual properties, such as,

for instance, an index of refraction that may be negative (left-handedness), less than one, or modulated in a graded manner. Suchmaterials have spurred considerable interest and led to numerousapplications over the past decade [ 1-3 ].

Metamnaterials may be equivalently described in terms ofmedia parameters (electric/magnetic dipole moments, electric/magnetic susceptibilities, permittivity, permeability), or in terms oftransmission-line (TL) parameters (inductance/capacitance, imped-ance/admittance, propagation constant/characteristic impedance).The latter approach, introduced in [4-6], has led to low-loss andbroadband metamnaterials, due to the non-resonant nature of thestructural elements. This has been the foundation for the vastmajority of the practical applications reported to date. Moreparticularly, the concept of composite right-/left-handed (CRLH)transmission-line metamnaterials (introduced in [7] and theorized in[I1]), which describes in a simple and insightful manner thefundamentally dual right-handed (RH)/left-handed (LH) nature ofmetamaterials, has been widely recognized as a powerful paradigmfor the understanding of metamaterial phenomena and the designof metamaterial devices.

IEEE Antennas and Propagation Magazine, Vol. 50, No. 5, October 2008

The applications of metamaterials may be classified in threecategories: i) guided-wave components (multi-band, enhanced-bandwidth, and miniaturized components; tight broadbandcouplers; compact resonators; uniform power combiners andsplitters; UWB filters; agile distributed amplifiers; impulse delaylines and circuits); ii) refracted-wave systems (focusing slabs,super-resolution imagers, reflection-less curved refractors,coordinate-transformation-based graded-index structures forelectromagnetic manipulations); and iii) radiated-wave devices(mono/multi-band passive/active one-dimensional/two-dimensional leaky-wave/resonant antennas and reflectors). Thisreport is concerned with the third category. It presents a selectednumber of the most practical CRLH metamaterial leaky-wave andresonant antennas.

The paper is organized as follows. Section 2 recalls thefundamentals of CRLH transmission-line metamaterials [1).Section 3 first establishes the fundamental leaky-wave propertiesof CRLH structures (Section 3. 1). It then presents dynamically-scanning leaky-wave antennas, which radiate a fan beam(Section 3.2), a conical beam (Section 3.3), a pencil-beam(Section 3.4), and an actively-shaped beam (Section 3.5). Section 4first establishes the funadamental resonant properties of CRLHstructures (Section 4.1). It then presents resonant antennasexhibiting multiple-band operation (Section 4.2), efficient zeroth-order resonance (Section 4.3), low-cost high directivity with high

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efficiency (Section 4.4), and electric/magnetic planar-monopoleradiation (Section 4.5). Finally, Section 5 concludes the paper.

2. Fundamentals of CRLH Metamaterials

Figure 1 shows the equivalent circuit and three implementa-tions (one-dimensional, two-dimensional, and three-dimensional)of periodic CRLH metamnaterials. It should be noted thatperiodicity is here a convenience but not a necessity, as long as thelargest cell is much smaller than the guided wavelength (p << Ag)

for electromagnetic homogeneity. Another important note is that aslong as the effective-medium condition, p << A,~ is satisfied, there

is no constraint on the minimum number of unit cells required formetamaterial operation. Even one single cell, when perfectlymatched to the external world (i.e., presenting a Bloch impedanceequal to that of the external media or ports), behaves in a mannerthat cannot be distinguished from the behavior of a perfectlycontinuous medium of the same electrical size for the wavecrossing it [1, Section 3.2.8].

A CRLH transmission-line unit cell, as shown in Figure Ia, ischaracterized by the immittances

interdigital capacitor (C'L)

microstrip trace

,via

stub inductor (LL)

ground plane

Figure lb.metamaterial.

A one-dimensional

Z=R+j(wLR- I )L

(w/ws0e)2 _I pIA-*O Zý'p(DCL

Y=G+j COCR-j1

(lb)

___IC _ pA+O ho0,)21LLA

Figure Ic.metamaterial.

A two-dimensional CRLH (microstrip)

Z/2 Z/2

... .. ... .....Y~ ........ .......

R/2 LR/2 2 0 L 2 0 L LR,'2 R/2

Z' Z/P L'1 LR/P

Y'Y/P G~ kR CRJ CR/PR'= R/p L' L

L L

Fignre Ia. A CRLH transmission-line metamaterial(symmetric) unit cell (one-dimensional, extensible to twodimensions and three dimensions). The primed variablesrepresent per-unit-length and times-unit-length quantities.

26

Figure 1d. A three-dimensional CRLH (rotated transmission-line-matrix-based) metamnaterial [8].


CRLH (microstrip)

(Ia)


Here,

Cons=l/ JR CL =l/ JR7CT

and

a'sh =1/ VLL CR = 1/ Lý Cý

are the series and shunt resonances, respectively.p/2g = flp/27r -*ý0 (with p being the period and 2g being theguided wavelength) represents the infinitesimal limit of a perfectlyuniform transmission line (or homogenous metamaterial).

The general transmission matrix for a two-port-network unitcell of the type shown in Figure 1 a is obtained by multiplying theindividual element matrixes, i.e.,

[1, Z/2; 0, 1][[1, 0; Y, 1] [l, Z/2; 0, 1].

This yields

[ABCD] =[Il+ZY/2, Z(Il+ZY/4); Y,I±+ZY/2],

where A =D =1 +ZY12 due to symmetry, and AD -BC=lI due toreciprocity.

The Bloch-Floquet theorem is next applied to describe theperiodic structure obtained by cascading the unit cell. Thus,V i(x +p) e CBPV y(x), where rs = aB + jIB is the Bloch (peri-

odic) complex propagation constant, and V/' represents either thevoltage or current. This leads to the nullified-determinant equation

e +rap+ (AD -BC) e rBP -( )=0

which simplifies here to cosh(rBp) A by symmetry andreciprocity. Similarly, the Bloch impedance is obtained by the ratioof the periodic voltage and current at either port of the unit cell,and reads as follows:

ZBU(co) =2B/[D -A± V(A +D)2 -4(AD-BC)j

(±refers to positively/negatively traveling waves). This simplifies

here to ZB (co) = B!Vý 1- (positively traveling wave case) bysymmetry and reciprocity. Summarizing,

Y(~ T ,+ZY14 YBP+o ~~a) - (b

Equations (2a) and (2b), respectively. In the low-loss first-orderapproximation, the ZY and Z/Y parameters for the CRLH caseread


-2 2 2Ky+4 + 8

IA-*0O Z ZY

Z2(a'as)lC..) 2 _ l+n3m PIAg--O Z' Z(a'/a'sIh) 2 _I+.5den YY

where

'R =! 'ILRCR,

a)L =1/ LCL,

(3a)

(3b)

ZL = 4/ CL

and where = J (R, G) is a loss term that is much smaller than

each of the R, G -independent terms.

In the so-called balanced resonance case, w'se = a.'h (equalseries and shunt resonances), the above relations reduce to

Zy = _(0)10R--a'LIC) 2 +,5

and

Z/Y -Z 2 =L IC --

respectively. Equations (2) take then the simple form

rB(CO)=-coshl I---{ a' - +

y~gp--*O Z a)- CL=R

(4a)

(4b)

where it appears that in the infinitesimal limit, the dispersionrelation, 8 (w'), is purely real and therefore gapless. The dispersion

relation splits into a negative hyperbolic left-handed contribution,dominating at lower frequencies, and a positive linear right-handedcontribution, dominating at higher frequencies. The characteristicimpedance is independent of frequency, and therefore favorable forbroadband matching to constant impedance (generally, 50 fl) ports.A typical CRLH Bloch dispersion relation, j8B (a'), and a Bloch

impedance, Zs (a'), are plotted in Figure 2. The gap existing in

general between the left-handed and right-handed bands due to thedifferent series and shunt resonances closes up when the line isbalanced as a result of mutual cancellation of these resonances,

27


which leads to a continuous transition between the two bands. Thisproduces a nonzero group velocity (vg = •wag#0) and an

infinite guided-wavelength (Ag = 27r//J = oo) wave at the transition

frequency, WO = C~se = 0),h (8. = 0), and a fairly constant Bloch

impedance around ak), allowing broadband matching. Exact

expressions for the left-handed high-pass cutoff frequency, WcL,

and the right-handed low-pass cutoff frequency, WcR, were givenin [l1).

The CRLH transmission-line metarnaterials presented in Fig-ure 1 are double-Drude, and hence dual-band, media. They haveequivalent constitutive parameters

ii = -iZ'/a) =L 4 IaW2)

and

= = y'c (= -ý hI 2),

characterized by one pole at w = 0 and one zero at the plasma fre-quency (awewh). These parameters may be positive, negative, or

less than one. If both are simultaneously zero/negative (balancedtransition frequency wo), the index of refraction is zero/negative.Higher-order transmission-line metamaterials are possible, forinstance such as the double-Lorentz (one pole and one zero in V~and p) tri-band and the extended CRLH quad-band metamnaterialsintroduced in [9] and [10], respectively.

3. Leaky-Wave Antennas

Leaky-wave antennas, either in uniforn or periodicconfigurations, have been abundantly studied for over half acentury [11, 12]. They essentially provide the benefit of highdirectivity without requiring a complex feeding network, as doantenna arrays. However, they suffer from major limitations intheir scanning capabilities, which have limited their applications todate. CRLH meta-structures have essentially suppressed theselimitations, and thereby paved the way for novel perspectives forleaky-wave antennas.

A leaky-wave antenna is a traveling-wave structure with acomplex propagation constant y (w) = aw(w) +Ijfl(w).- a,. (w)

is the leakage factor (typically, al/ko < 0.02, i.e., a length of at

least 1bA0 is required to radiate 90% of the power [12]). /1(w) is

the dispersion relation. When the wave velocity is faster than thevelocity of light (or P (w)) < ko) the main beam of such an antenna

radiates in the direction

0 (w) = sin- [6 (a))/ko] sin-' [c,6 (w)/w],

where 9 is the elevation angle from the direction normal to thestructure [11, 12]. This formula shows that the main beam may bescanned with frequency, if the structure is dispersive. Conventionalleaky-wave antennas are restricted to strictly positive 9 (due tostrictly positive 61) for uniform configurations, or to adiscontinuous range of negative or positive 9 excluding broadside

28

(9 =0, due to the standing-wave nature of the wave at /3= 0) forperiodic configurations (using space harmonics).

3.1 Leaky-Wave Propertiesof a CRLH Structure

A CRLH metamaterial exhibits a unique dispersion curvethat extends across the dispersion diagram all the way from theregion 6 < -k0 to the region /3> +k0 . When open to free space,this gives rise to the four distinct regions shown in Figure 2a (1:left-handed, guiding; HI: left-handed, radiating; III: right-handed,radiating; IV: right-handed, guiding). Moreover, as pointed out inSection 2, in the balanced case (Ws, = sh), the /3= 0 transition

from the left-handed to the right-handed bands is seamless. Thistransition is characterized by the frequency a)0 where 61= 0 withvg•# 0, allowing an infinite-wavelength (or infinite-phase-

velocity) traveling wave. Finally, the (periodic) CRLH mode isessentially a fundamental (n = 0) space-harmonic mode, since thehigher space harmonics are negligibly excited in the metamaterialfrequency range (p P ,Kg) .

3.2 Fan Beam

As a consequence of these properties, a CRLH leaky-waveantenna scans the entire space from 9-= -90' to 9-= +90',including 09 0%, as frequency is varied from co = -/)c toco = +/8c (CRLH dispersion regions RI and MI). Moreover, it maybe excited by an elementary and efficient (simple transmission-line) feeding mechanism, due to the fact that it operates in afundamental mode. This backfire-to-endfire antenna wasdiscovered in [13], and later extensively studied and applied (e.g.[1]).

It should be noted that backward radiation using a one-dimensional left-handed transmission line (first demonstrated in[4]) was verified before [13] in [14]. However, the antenna in [14]was capable only of backward radiation, a capability alreadyavailable in conventional periodic-type leaky-wave antennas [11,12]. Exhibiting a gap at /l = 0, this antenna was also incapable ofbroadside radiation, which was achieved for the first time in [13].It is really the concept of CRLH transmission-line metamnaterials[7], observed in [ 13] but fully understood in [7, 11 ], which enabledthe most practical backfire-to-endfire leaky-wave antennas, andmany subsequent antenna and component applications [1].

Figure 3a shows a CRLH leaky-wave antenna and illustratesits full-space scanning capability, while Figure 3b presents typicalscanning capabilities for this type of antenna. Instead of beingfrequency scanned, the antenna may also be electronically scannedat a fixed frequency, as required in many applications, usingcapacitors or inductors controlled by a bias field. In this case, it isalso possible to equalize the beam by using a nonuniform biasingdistribution along the structure [15]. The antenna considered ishere a one-dimensional structure. Therefore, it provides scanningonly in one plane. While the beam is very directive in this plane (xzin Figure 3a), it is fat in the perpendicular direction (y inFigure 3a): this is afan beam.



W = Wo

broadside

WBF < W < WO

backwardz WO < W < WEk'

CX

Figure 3a. A backfire-to-end-frie fan-beam CRLH leaky-waveantenna (e.g., a structure of the tye shown in Figure 1b).

bias voltage (V)

10T

4.0 4.5 5.0 5.5frequency (GHz)

Figure 3b. The angle-scanning law for both a frequency-scanned leaky-wave antenna and a fixed-frequency (3.3 GHz)electronically scanned CRLH leaky-wave antenna. The insetshows a prototype unit cell of the latter, including both (forBloch-impedance equalization) series and shunt varactors, thebias voltages of which control the beam scanning. The broad-side gain of this antenna was 8 dB for N = 25 and 12 dB forN = 50 unit cells, with a cell size of p = 6 mm.

Figure 4a. A conical-beam two-dimensional CRLH structureand radiation principle, and three different implementations:1. A patch mushroom structure; 2. An inter-digital mushroomstructure; 3. A stepped-impedance structure.

'-b70 '

0 80 GAP ' Pi

~2010

frequency (GHz)

Figure 4b. The conical-beam two-dimensional CRLHdispersion and radiation.

IEEE Antennas and Propagation Magazine, Vol. 50, No. 5, October 2008 2

Z O(W)\_ý' W I

29


Due to its exceptional flexibility, this leaky-wave antennawas also applied successfully to various passive and active smartreflectors, a few of which were described in [I].

3.3 Conical Beam

Because it is essentially "seen" by electromagnetic waves as auniform medium, a one-dimensional CRLH structure may bestraightforwardly extended to two dimensions Ojust like a narrowstrip maybe extended to a rectangular patch), as shown inFigure 1 c. When excited in its center -for instance, by a coaxialprobe -such a two-dimensional GRIM structure supports in itsmetamaterial regime a perfectly circular wave. When this wave hasa phase velocity faster than the speed of light (CRLH dispersionregions II and RII, it radiates in a leaky-wave manner. This resultsin a conical-beam antenna (i.e., maximum radiated power on a

p=0 - 27r circle under an elevation 0 around the normal axis), asillustrated in Figure 4a and demonstrated in Figure 4b [16]. As thefrequency varies, the opening angle of the conical pattern varies,following the dispersion relation fl(ro), which leads to beam

scanning. Due to azimuthal symmetry, the left-handed and right-handed ranges provide the same fuinctionality, but with oppositescanning slopes (d9/dw).

A fundamental difference between this antenna and conven-tional two-dimensional leaky-wave antennas is that the latter use apartially reflecting sheet array [17] or a dielectric superstrate layer[18]. These are typically very sensitive to fabrication tolerances,whereas the former is an easy-to-fabricate uni-planar structure.

3.4 Pencil Beam

In point-to-point applications, a pencil beam (i.e., maximumradiated power in a unique direction, 9, q), of space) is required.Such a beam cannot be produced efficiently with dynamic scanningcapability in a two-dimensional structure of the type discussed inSection 3.3, even with edge excitation. On the other hand, the con-ventional phased-array option requires a complex, cumbersome,lossy, and dispersive two-dimensional feeding network.

Several interesting pencil-beam scanning CRLH leaky-waveantennas have been reported [ 19-22]. The antennas of [ 19-21 ] con-sisted of arrays of leaky-wave elements, using a combination offrequency tuning and phase-shift tuning to achieve pencil-beamscanning. [19] used heterodyne mixers and delay lines with filtersfor scanning. [20] used a Butler matrix. [21] used varactor diodesand a Rotman lens. [22] used a two-dimensional surface, of thetype discussed in Section 3.3, but excited at two of its edges bytuned power levels. We present here a particularly economical andflexible two-dimensional radiating-aperture solution, involvingCRLH structures playing two distinct roles.

This proposed antenna is depicted in Figure 5a, and its full-space scanning capability is shown in Figure 5b [23]. It consists ofa phased array of frequency-scanned (the case depicted inFigure 5a, with subsequent LO mixer control) or electronically-scanned (LO control replaced by varactor control) CRLH leaky-

wave antennas. These are fed by a series uniform (Ag = ) boxed

(to suppress leaky-wave radiation) [24] CRLH series power divider

30

[25-27]. This solution does not require any corporate feedingnetwork. In addition, it allows an arbitrary number of antennaelements, which may be arbitrarily spaced, and the number ofwhich may even be dynamically controlled for real-time beamshaping [27].

3.5 Active Beam Shaping

The exceptional flexibility of the CRLH metamaterials hassuggested several ideas incorporating active elements [28]. Due tothe sub-wavelength nature of the unit cell, low-gain transistors maybe ideally integrated along the structure to manipulate themagnitude of the signal along it, in addition to the phase.

The beam width of a leaky-wave antenna is controlled by itsleakage factor. For a given passive structure, this factor is fixed.Beyond the length for which all of the power has been radiated, nofurther increase of effective aperture, and therefore of directivity,may be achieved. In order to suppress this limitation, an activeCRLH leaky-wave antenna, incorporating amplifiers as repeaters(or power regenerators), was proposed in [29]. This idea wasalready implemented in a conventional leaky-wave antenna in [30].This antenna is virtually capable of providing an arbitrarily highdirectivity - since the effective aperture is unlimited - with singleand simple transmission-line excitation. Massive gainimprovement (8.3 dB) compared to the case of a correspondingpassive antenna was easily achieved, both from increaseddirectivity and from increased efficiency achieved by thereflection-canceling unilateral nature of the amplifiers.

A beam-shaping generalization of this concept was discussedin [29] and demonstrated in [3 1]. It is well known that the radiationpattern of an antenna is essentially the Fourier transform of itsaperture field distribution [32]. Therefore, the shape of the radiatedbeam of a metamnaterial antenna - again, due to its sub-wavelengthstructure and feeding-network-less configuration -may be easilymanipulated by approximating desired aperture distributions usinggain-controlled amplifiers. This concept, taking into account theexact exponentially decay nature of the leaky sections in thedesign, is called active "digitized aperture" beam forming. It isillustrated in Figure 6a, verified experimentally in Figure 6b for thecase of a uniform distribution, and illustrated in Figure 6c for thecompared cases of a uniform distribution (maximum directivity)and a binomial distribution (minimum sidelobe level).

LO (3 GHz

Figure 5a. A pencil-beam CRILH leaky-wave antenna.


IF(O-5-1.3 GHz)

Y


N

0

C)

0.)

3.5G~z3.9 GHz 4.3 GHz

Phaseshift

400

-do

#2

#5

#8

#3

#6

- #9

#4

#7

-yp = (a + jf3)pFigure 2a. The CRLH Bloch dispersion diagram computed byEquation (4a) for both the balanced (W,,s = Wh= wo) and the

unbalanced resonance (Ws, # Wh) designs. Balanced parame-

ters: LR =LL =2.5 nH, CR =CL =lpF, fcL= 1.32 GHz,

f~h fs =f0 31Gz fcR = 7.64 GHz. Unbalancedparameters: LR = 2 nH, LL = 2.5 nil, CR = 1 pF, CL = 0.75 pF,

fcL = 1.51 GHz, fsh =3.06 GHz, fo = 3.62 GHz, &s= 4.27 GIz,fcR = 8.64 GHZ.

10

9 -

0 ~-Re(ZB)6 ~ bal --- IM(ZB)

un- - mg un.

ZB ,norFiue5-TeCLHnraie t OD lc meac

ure Zn

Figure 5b. The full-space scanning capability of the antennashown in Figure 5n (same parameters as in Figure 3b).

Figure 6a. An active "digitized aperture" beam-forming con-cept and active CRLH antenna prototype (a microstrip groundplane with via transition slots between the top and the bottom).

Figure 6b. The radiation pattern at 3.7 GHz for the activeleaky-wave antenna shown in Figure 6a with a uniformaperture distribution. The "passive" case represents the sameantenna without any amplifier.


fRF= 3.5 GHz

31


Uniform Elhnomial 0

**-l0dB . tB300 3100

600D . 3VCsiSC 600TC2dBIFHPBIVW=54~SLL- -12.7d9, ......................... L -15.4

Figure 6c. The radiation patterns (array-factor results) corre-sponding to the aperture distributions shown in Figure 6a (uni-form: maximally directive, binomial: minimal sidelobe level)for N =48 CRLH unit cells with one amplifier every third cell.

CRLH leaky-wave structures incorporating active elementsmay lead to a quasi- "universal" type of antenna. This wouldsimultaneously provide beam-scanning and -shaping functionality,controlled in real time by a DSP (digital signal processor). Forinstance, the idea of dynamic radiation pattern diversity (DRPD)MIMO systems was recently introduced [33). In these, efficient andlow-cost CRLH antennas perform a real-time scanning calibrationto the scattering environmnent for channel maximization.

4. Resonant Antennas

Leaky-wave antennas offer the advantage of high directivitywithout requiring any complex feeding network. CRLHimplementations render leaky-wave antennas particularly flexiblein terms of dynamic scanning capability. However, all leaky-waveantennas suffer from the drawback of limited bandwidth related tobeam squinting aelac, in fixed frequency applications (a problemthat may find a solution using enhanced group velocity, possiblyeven superluminal with relatively high dispersion, and lossescompensable by active elements [34]). In this case, resonant CRLHantennas exhibit complementary properties and offer otherbenefits, compared to their leaky-wave counterparts.

4.1 Resonant Propertiesof a CRLH Structure

A CRLH resonant antenna is obtained by reactively terminat-ing a CRLH transmission-line structure that is open to free spaceby a short or an open circuit. Such an antenna may then bedesigned in the same way as a conventional uniform-metallizationantenna (e.g. a patch), but with the effective wavelength andfrequency response of a CRLH metamaterial.

The resonant modes of a CRLH structure of length I aregiven by A = mj~g/2 (m an integer) or, equivalently, by

&tm = m~r. In these, in can be both positive (right-handed band) ornegative (left-handed band), and even zero (transition frequency),following O3(w). More specifically, each positive (m > 0) resonancemode has a twin negative (m < 0) resonance mode, and a zeroth-order (m = 0) mode exists at the transition frequency, WOO. Due toits discrete nature and subsequent finite bandwidth, a CRLHstructure has a finite number of 2N (2N - 1 in the balanced case)resonances. These correspond to 6fip =,OPm (f/N) = m~r/N, where

Nis the number of unit cells of the resonator. These resonances are

32

found by calculating the frequencies, co., mapping the abscissas

Pm = m 7rl(Np) of the dispersion diagram (Equation (4a)), viz.

W2 2 My)] I C/'+ S-+W 21-cos 2ITIO e+Wh),eDR CM N L

(5)

where n=0,±l,...,±_(N-l). The discrete spectrum of a CRLH

resonator is shown in Figure 7. Whbile the resonance frequenciesdepend only on the dispersion relation, their coupling factor toexternal sources naturally depends on the Bloch impedance(Figure 2b), and on the excitation mechanism.

4.2 Multi-band

Figure 8a shows a dual-band half-wavelength (m = ±1) open-ended resonant antenna, the dual-band property of which is animmediate consequence of the positive and negative resonancepairs (Equation (5)) of a CRLH structure [35]. This antenna isback-fed by a coaxial line at an off-center location for 50 12matching at the frequencies f±,. In principle, all of the 2N -1

resonances may be excited (with the exception of w,,e, due to the

absence of series currents [1) for open-end termination) andmatched to the source with proper excitation. The modes of each±m pair have the same guided wavelength and field distribution,as illustrated for m = ±1 in Figure 8b. They therefore present verysimilar input impedances. This allows efficient dual-bandoperation from a single resonator. The return loss of the dual-bandhalf-wavelength antenna is plotted in Figure 8b (where the weaklyexcited mode o)0sh is also visible, whereas, as expected, no wseresonance exists).

This dual-band antenna has its polarization (E field) along theaxis of the structure (x in Figure 8a). It exhibits a half-wavelength-patch type of radiation pattern, with gains of 4.2 dB and 4.5 dB forf- and f,-,, respectively, and cross-polarization lower than-20 dB.

0 1 2 3 4 5 6 7 N-i1NN NN N NN N

Figure 7. The resonance spectrum of a CRLH resonant struc-ture comprised of N unit cells (here, the unbalanced situation,with two distinct # = 0 resonances, o),, # w~h).



7.06.56.05.5

S4.52.4.0

3.5,3.0~2.5'U_2.0

1.51.00.50.0

RH region

p +1 1

"-+4

banda~Y1

D f..-

f-i

Pah/i

Figure 8a. A half-wavelength (t = g /2, mn = ±1) open-ended

seven-cell CRLH microstrip resonant antenna (with symmetricLL stubs for low cross-polarization), and the correspondingdispersion relation including all possible resonance modes.

2.4 2.6frequency (GHz)

Figure 9a. A zeroth-order (t/2g = 0, mn = 0) short-ended (ws,)

nine-cell CRLH microstrip, resonant antenna, and correspond-ing return losses.


fre. (GHz) 2.42 2.23 2.42t (n*n 102 118 33

t40.83 0.88 0.27D (dB) 8.8 9.6 6.317,.d N% 72 1 39 176

Gacc,(dB) 7.4 1 5.5 15.1

33

Tni-band [9]1, quad-band [10] (and even potentially largernumbers of bands) operation is also possible in principle withhigher-order CRLH transmission lines. However, control of theparasitic contributions becomes increasingly challenging as thenumber of bands increases. Since no balancing is required forresonant antennas, a simple CRLH structure may be designed as atni-band antenna by exploiting the relaxation of this constraint [36].

4.3 Zeroth-Order Resonance

The zeroth-order CRLH resonance (ti/Ag = 0, m = 0) is

particularly unique and interesting [37]. Figure 9a shows a zeroth-order CRLH antenna, shorted at its output by via holes, and usingat its input an inter-digital capacitance with a high-imnpedancetransmission line to transform the impedance to a quasi-short of afew ohms. Being short-circuited at both ends, this antenna operatesin the w,, mode (no O~h mode exists, due to the absence of shuntcurrents) [38]. The corresponding return loss is plotted in the samefigure, while Figure 9b shows the fuill-wave-simulated uniformcurrent distribution and radiation pattern of this coe mode. Thepolarization is similar to that of the half-wavelength antennapresented in Section 4.2.

Figure 10 compares the CRLH zeroth-order antenna of Fig-ure 9 with a half-wavelength CRLH antenna (Section 4.2) and witha classical patch antenna. Figure 10a shows that while theresonance frequency varies with the size of the structure for thehalf-wavelength and patch antennas, it remains constant for thezeroth-order antenna, where w,,s depends only on the lumpedvalues LR and CL (or wh on LL and CR ). This property may beexploited to design electrically small or electrically large antennas.Table 1 shows that the zeroth-order antenna may reach an excellentefficiency, comparable to that of a conventional patch antenna,probably due to the more-uniform aperture-field distribution.

The properties of CRLH zeroth-order antennas might lead tooptimal electrically small antennas [39]. For these, the fundamentalreduction of efficiency may be mitigated by the perfectly uniformdistribution of energy along the structure. Furthermore, the funda-mental reduction of directivity may be mitigated by the correspond-ing maximized effective aperture.

Table 1. A performance comparison for three of the antennasof Figure 10a. (Note: The larger directivity of the n = I CRLH

antenna compared to the n = 0 CRLH antenna is due to thelarger physical size of the former).

bandgý

n --2LH relaion 3

CRLHn=O CRLRn = +1 ý Patch


4.4 High Directivity

A resonant CRLH antenna exhibits unique properties interms of directivity. Compared to a conventional array, it has asingle element, and does not require any corporate feeding network[32]. Compared to a leaky-wave antenna, it has a higher apertureefficiency, because it does not have an exponential decay of poweralong the aperture [38, 40].

Because its operating frequency is solely determined by itsLC unit-cell elements (Figure l~a), the size of a resonant CRLHantenna may be enlarged considerably at a fixed frequency. It mayhence provide a super-high directivity, resulting from a very largeeffective aperture. This fact is illustrated in Figure 10b, where theCRLH antennas, operating at around 2.4 GHz, are more than threetimes larger than a patch antenna operating at the same frequency.They therefore exhibit a strongly enhanced directivity (of over2.5 dB), which could be even much further increased with furtherincreased size. Figure I1I presents a CRLH zeroth-order antenna inthree different sizes. Its increasing gain with increasing size (withfixed frequency) corroborates the previous statements.

Figure 12a. A CRLH loop resonant microstrip antenna.A CRLH resonant antenna may thus virtually attain the

directivity of any large-scale array, without requiring the array'slarge, cumbersome, and lossy feeding network. To attain such ahigh directivity, a leaky-wave antenna may require active elements radial slot. w,,for power regeneration (Section 3.5).

4.5 Electric/Magnetic Planar Monopoles

CRLH resonant structures in loop configurations have severalunique functions. For instance, they may be used as versatile multi-band and multi-polarization (linear/circular) antennas [41], or aselectric/magnetic planar monopole (i.e., with azimuthallysymmetric radiation patterns) antennas, which will be presentedhere [42].

azimutha Slot, W~h

Figure 12b. The two-port feeding structure for thesimultaneous excitation of Wse, and 0coh modes by backplanemicrostrip-fed slots, with a slot parallel to one stub and with aslot perpendicular to one stub for ws, and wh, respectively.

A CRLH loop structure is obtained by folding a rectilinearCRJLH structure so as to form a closed circular loop, as shown in

-FT-FT-the prototype of Figure 12a. There, the stubs are compactly placed-w 7ý%, G .8ý (18(nosi(!Iof,)in a radial manner within the loop area, with a unique shorting via

at the center of the structure. Possible backplane slot excitationsare shown in Figure 1 2b. Due to the additional (compared to therectilinear case) azimuthal boundary condition, such a CRLHstructure supports only modes of even (in) order, i.e.,

I f I I II rI Icorresponding to loop circumferences that are multiples of the71 G 9 d~. ýS 14 B _j guided wavelength.

Figure 11. Three CRLH zeroth-order (w,,; shorting vias at the Monopole-type radiation may by achieved in the zeroth-order

output and quarter-wave transformer to low impedance at the (mn = 0) mode. The operation of this mode is described ininput) microstrip resonant antennas of different sizes, Figure 13 with the help of fuill-wave computed surface-currentoperating at the frequency of 2.4 Gil; with respective distributions. In the wse mode, Z -0 (Equation (la)), andefficiencies (qi), gains (G), and sidelobe levels (SSL). therefore the series paths are seen as open circuits. This leads to

34 IEEE Antennas and Propagation Magazine, Vol. 50, No. 5, October 200834


10-

-20F

-25!-

-3%,2.25 2.5 2.75 3 3.25 3.5

Frequency [GHz]

N

n= -1( -2

.04

0 50 100 150 200

S= Np (mm)

Figure 10a. A resonance-frequency comparison of a zeroth-order CRLH antenna and a half-wavelength CRLH antenna,for the structure of Figure 9a, with a classical patch antenna.

Figure 8b. The field distributions for the modes m = ±1 in Fig-ure 8a, and the corresponding return losses.

Figure 9b. The Current distribution along the CRLH zeroth-order mode ('e)antenna of Figure 9a.

Figure 13. The zeroth-order series and shunt resonances of aCRLH loop microstrip structure, leading to magnetic and elec-tric dipole radiators, respectively.


M

A, h

f- i

-i 81 Measurementsill MoM Simulation

I

I

35


constant nonzero current associated with zero voltage along theloop: a magnetic-monopole radiator is thus produced. In the W),h

mode, Y = 0 (Equation (lb)). The shunt paths are therefore seen asopen circuits (the radial currents in the stubs represent only the LL4excited currents, but the overall shunt resonator current is zero),leading to zero loop current associated with constant nonzerovoltage along the loop: an electric-monopole radiator is thusproduced. These two electric and magnetic monopole modes areindependent (uncoupled) from each other. It should be noted thatthey may be excited simultaneously using two different slots(radial for wse and azimuthal for oh ), as shown in Figure 12b.

Monopole radiators may also be obtained in the zeroth-orderCRLH resonance in a "patch" configuration, providing acomplement to the magnetic dipole of a conventional patch [32]. Amagnetic-monopole patch radiator, operating in the CRLH zeroth-order mode w0 h' is shown in Figure 14a, along with its typicalradiation patterns in Figure 14b [43]. This monopole behaviorresults from the magnetic current loop M, = -2n x E, createdaround the mushroom patch due to the uniformity of the verticalelectric field. An electric-monopole patch radiator operating in theCRLH zeroth-order mode w, may also be conceived.

A

y

COIIv.

iSY

z 1ýx

I ICRLH

z

Figure 14a. A magnetic monopole microstrip patch antennausing the CRLH zeroth-order mode o)1sh in a mushroom struc-ture. The case of a conventional patch is also shown, for com-parison.

0-

-10.

-20.

-40*

-30

-20-

-10.

0.

Z30

2?0 12X

240 , , -nmm

is$ - -- exp.

0.

.a0.

.30.

-40

0-M

-20

-10

0'

z

330 30

3W so

240 120

210 1SO0 num.100 - W-

Figure 141b. Typical radiation patterns for the magnetic mono-pole antenna of Figure 14a (CRLH patch of 24l cells, size of15 x 7.5 cm2, measured at 3.5 GHz).

36

5. Conclusions

CRLH transmission-line metamaterials represent a paradigmshift in electromagnetics engineering and, in particular, forantennas. They exhibit exceptional properties, resulting from theirrich dispersion and their fundamental left/right-hand duality. Theyoffer simple and deep insight into metamaterial phenomena, andprovide efficient tools for the practical design of components andantennas.

This paper has presented an overview of selected leaky-waveand resonant CRLH antennas, which all exhibit functions or/andperformance superior to prior state of the art. Conventional leaky-wave antennas have the advantage of providing high directivitywithout a complex feeding network, but suffer incompletescanning-range capability. CRLH leaky-wave antennas offer, forthe first time, full-space scanning capability, including truebroadside radiation (at the transition frequency, where the guidedwavelength is infinite). They operate in the fundamental mode ofthe structure, which is excited in a simple and efficient manner.This capability has been used in the design of novel antennasradiating an efficient fan beam, a conical beam in a convenient uni-planar configuration, and a pencil-beam avoiding a complexfeeding network from a novel CRLH uniform series feeding line.Moreover, the sub-wavelength nature of the CRLH metamnaterialcell has been exploited for the integration of variable-gainamplifiers, providing active "aperture digitization" for versatilebeam shaping and beam scanning.

CRLH leaky-wave antennas, like their conventional counter-parts, suffer from limited bandwidth associated with beamsquinting in fixed-frequency point-to-point applications. In thiscase, if no dynamic scanning is required, resonant CRLH antennasprovide complementary solutions, in addition to other benefits.CRLH metamnaterials are inherently dual-band, a property that hasbeen applied in dual-band antennas, and even in tri-band antennasprofiting from the CRLH balance-condition relaxation in theresonant regime. The zeroth-order resonances (corresponding toinfinite guided wavelength) have been demonstrated to beparticularly beneficial for high-efficiency and high-directivity (andhence, high-gain) antennas, still without a complex feedingnetwork and also without beam squinting. In particular, thesezeroth-order structures have been shown to provide originalsolutions for planar electric- and magnetic-monopole antennas.

6. References

1. C. Caloz and T. Itoh, Electromagnetic Metamaterials, Transmis-sion Line Theory and Microwave Applications, Piscataway, JohnWiley/IEEE Press, 2005.

2. G. V. Elefiheriades and K. G. Balmain (eds.), Negative-Refrac-tion Metamaterials, Piscataway, John Wiley/IEEE Press, 2005.

3. N. Engheta and R. W. Ziolkowski (eds.), Electromagnetic Meta-materials: Physics and Engineering Explorations, Piscataway,John Wiley/IEEE Press, 2006.

4. C. Caloz and T. Itoh, "Application of the Transmission LineTheory of Left-Handed (LH1) Materials to the Realization of aMicrostrip LII Transmission Line," IEEE International Symposiumon Antennas and Propagation Digest, San Antonio, USA, June2002, pp. 412-415.



5. A. A. Oliner, "A Planar Negative-Refractive-Index MediumWithout Resonant Elements," CNCIUSNC URSI National RadioScience Meeting, San Antonio, USA, June 2002, p. 41.

6. A. K. Iyer and G. V. Eleftheriades, "Negative Refractive IndexMetamaterials Supporting 2-D Waves," IEEE International Sympo-sium on Microwave Theory and Techniques Digest, Seattle, USA,June 2002, pp. 1067-1070.

7. C. Caloz and T. Itoh, "Novel Microwave Devices and StructuresBased on the Transmission Line Approach of Meta-Materials,"IEEE International -Symposium on Microwave Theory andTechniques Digest, Philadelphia, USA, June 2003, pp. 195-198.

8. M. Zedler, C. Caloz, and P. Russer, "On a 3D Isotropic Left-Handed Metamnaterial Based on the Rotated TLM Scheme: Sche-matic Analysis, Experimental Verification, and Planarized Inmple-mentation," IEEE Transactions on Microwave Theory and Tech-niques, MTT-55, 12, December 2007, pp. 2930-2941.

9. A. Rennings, T. Liebig, C. Caloz, and I. Wolff, "Double LorentzTransmission Line Metaniaterials and their Applications to TribandDevices," IEEE International Symposium on Microwave Theoryand Techniques Digest, June 2007, pp. 1427-1430.

10. A. Rennings, S. Otto, J. Mosig, C. Caloz, and 1. Wolff,"Extended Composite Right/Left-Handed (E-CRLH) Metamnaterialand its Application as Quadband Quarter-WavelengthTransmission Line," IEEE Asia-Pacific Microwave Conference,Yokohama, Japan, December 2006.

11. R. E. Collin and F. J. Zucker (eds.), Antenna Theory, Part HI,New York, McGraw Hill, 1969, Chapters 19 and 20.

12. R. C. Johnson (ed.), Antenna Engineering Handbook, ThirdEdition, New York, McGraw Hill, 1992, Chapter 10.

13. L. Liu, C. Caloz, and T. Itoh, "Dominant Mode (DM) Leaky-Wave Antenna with Backfire-to-Endfire Scanning Capability,"Electronics Letters, 38, 23, November 2002, pp. 1414-1416.

14. A. Grbic and G. V. Eleftheriades, "Experimental Verificationof Backward-Wave Radiation from a Negative Refractive IndexMetamnaterial," J. App. Phys., 92, 10, November 2002, pp. 5930-5935.

15. S. Lim, C. Caloz, and T. Itoh, "Metamaterial-Based Electroni-cally-Controlled Transmission Line Structure as a Novel Leaky-Wave Antenna with Tunable Angle and Beamwidth," IEEE Trans-actions on Microwave Theory and Techniques, MTT-53, 1, Janu-ary 2005, pp. 161-173.

16. C. A. Allen, C. Caloz, and T. Itoh, "A Novel Metamaterial-Based Two-Dimensional Conical-Beam Antenna," IEEEinternational Symposium on Microwave Theory and TechniquesDigest, Fort Worth, TX, USA, June 2004, pp. 305-308.

17. G. von Trentini, "Partially Reflecting Sheet Arrays," IRETransactions on Antennas and Propagation, AP-4, October 1956,pp. 666-671.

18. N. G. Alexopoulos and D. R. Jackson, "FundamentalSuperstrate (Cover) Effects on Printed Circuit Antennas," IEEETransactions on Antennas and Propagation, AP-32, 8, August1984, pp. 807-8 16.


19. C. A. Allen, K. M. K. H. Leong, and T. Itoh, "2-D Frequency-Controlled Beam-Steering by a Leaky/Guided-Wave TransmissionLine Array," IEEE International Symposium on Microwave Theoryand Techniques Digest, San Francisco, USA, June 2006, pp. 457-460.

20. T. Kaneda, A. Sanada, and H. Kubo, "2D) Beam ScanningPlanar Antenna Array Using Composite Right/Left-Handed LeakyWave Antennas," MEICE Trans. on Electronics, 89, 12, December2006, pp. 1904-1911.

21. D. Lee, S. Lee, C. Cheon, and Y. Kwon, "A Two-DimensionalBeam Scanning Antenna Array Using Composite Right/LeftHanded Microstrip Leaky-Wave Antennas," Proc. Int. MicrowaveSymposium (IMS), Honolulu, USA, June 2007, pp. 1883-1886.

22. A. Lai, K. M. K. H. Leong, and T. Itoh, "Leaky-Wave Steeringin a Two-Dimensional Metamnaterial Structure Using Wave Interac-tion Excitation," IEEE International Symposium on MicrowaveTheory and Techniques Digest, San Francisco, USA, June 2006,pp. 1643-1646.

23. H. V. Nguyen, S. Abielmona, A. Rennings, and C. Caloz,"Pencil-Beam, 2D Scanning Leaky-Wave Antenna Array," Interna-tional Symposium on Signals, Systems and Electronics (ISSSE)Digest, Montrial, Canada, July-Aug. 2007, pp. 139-142.

24. N. Yang, C. Caloz, H. V. Nguyen, S. Abielmona, and K. Wu,"Non-Radiative CRLH Boxed Stripline Structure with High Q Per-formances," International Symposium on Electromagnetic Theory(EMTS) Digest, Ottawa, Canada, July 2007.

25. A. Lai, K. M. K. H. Leong, T. Itoh, "A Novel N-port SeriesDivider Using Infinite Wavelength Phenomena," IEEEInternational Symposium on Microwave Theory and TechniquesDigest, San Francisco, USA, June 2005, pp. 100 1- 1004.

26. M. A. Antoniades and G. V. Elefiheriades, "A MetamaterialSeries-Fed Linear Dipole Array with Reduced Beamn Squinting,"IEEE International Symposiuim on Antennas and PropagationDigest, Albuquerque, USA, July 2006, pp. 4125-4128.

27. H. V. Nguyen and C. Caloz, "Arbitrary N-port CRLH Infinite-Wavelength Series Power Divider," Electronics Letters, 43, 23,November 2007.

28. C. Caloz, F. P. Casares-Miranda, and C. Camacho-Pefialosa,"Active Metamnaterial Structures and Antennas," MediterraneanElectrotechnical Conference (MELECON) Digest, Benalmidena,Spain, May 2006, pp. 268-271.

29. F. P. Casares-Miranda, C. Camacho-Pefialosa, and C. Caloz,"High-Gain Active Composite Right/Left-Handed Leaky-WaveAntenna," IEEE Transactions on Antennas and Propagation, AP-54, 8, August 2006, pp. 2292-2300.

30. M. Chen, H. Z. Chan, B. Houshmand, and T. Itoh, "Characteri-zation of Leaky Wave Antenna and Active Gain Enhancement,"26th European Microwave Conference, September 1996, Prague,Czech, pp. 579-5 82.

3 1. S. Abielniona, H. V. Nguyen, F. Casares-Miranda, C.Camnacho-Pefialosa, and C. Caloz, "Real-Time Digital Beam-Forming Active Leaky-Wave Antenna," IEEE Symposium onAntennas and Propagation Digest, Honolulu, USA, June 2007, pp.5593-5596.

37


32. J. D. Kraus and R. J. Marhefka, Antennas, Third Edition, NewYork, McGraw Hill, 1991.

33. J.-F. Frigon, C. Caloz, and Y. Zhao, "Dynamic RadiationPattern Diversity (DRPD) MIMO Using CRLH Leaky-WaveAntennas," IEEE Radio Wireless Symposium (RWS) Digest,Orlando, USA, January 2008 (accepted for publication).

34. C. Caloz, S. Abielmona, H. V. Nguyen, and A. Rennings,"Dual Composite Right/Left-Handed (D-CRLH) Leaky-WaveAntenna with Low Beam Squinting and Tunable Group Velocity,"Phys. Stat. Solidi (b), 244, 4, April 2007, pp. 1219-1226.

35. S. Otto, C. Caloz, A. Sanada, and T. Itoh, "A Dual-FrequencyComposite Right/Left-Handed Half-Wavelength ResonatorAntenna," IEEE Asia Pacific Microwave Conference Digest, NewDelhi, India, December 2004.

36. A. Rennings, T. Liebig, S. Abielmona, C. Caloz, and P.Waldow, "Tni-Band and Dual-Polarized Antenna Based on CRLHTransmission Line," 37th European Microwave Conference(EuMC) Digest, Munich, Germany, October 2007, pp. 720-723.

37. A. Sanada, C. Caloz, and T. Itoh, "Zeroth Order Resonance inComposite Right/Left-Handed Transmission Line Resonators,"Asia Pacific Microwave Conference (APMC) Digest, Seoul,Korea, November 2003, pp. 1588-1592.

38. A. Rennings, T. Liebig, S. Otto, C. Caloz, and I. Wolff,"Highly Directive Resonator Antennas Based on CompositeRight/Left-Handed (CRLH) Transmission Lines," 2nd hInt. ITGConference on Antennas (MNCA) Digest, Munich, Germany,March 2007.

39. R. C. Hansen, Electrically Small, Superdirective, and Super-conducting Antennas, New York, Wiley Interscience, 2006.

40. A. Rennings, S. Otto, C. Caloz, and P. Waldow, "EnlargedHalf-Wavelength Resonator Antenna with Enhanced Gain," IEEEInternational Symposium on Antennas and Propagation Digest,Washington, USA, June 2005.

41. A. Rennings, S. Otto, T. Liebig, C. Caloz, and 1. Wolff, "Dual-Band CRLH Ring Antenna with Linear/Circular-PolarizationCapability," European Conference Antennas and Propagation(EuCAP) Digest, Nice, France, November 2006.

42. S. Otto, A. Rennings, C. Caloz, and P. Waldow, "Dual-ModeZeroth Order Ring Resonator with Tuning Capability and SelectiveMode Excitation," IEEE 35th European Microwave Conference(EuMC) Digest, Paris, France, October 2005, pp. 149-152.

43. A. Lai, K. M. K. H. Leong, and T. Itoh, "Infinite WavelengthResonant Antennas with Monopolar Radiation Pattern Based onPeriodic Structures," IEEE Transactions on Antennas and Propa-gation, AP-55, 3, March 2007, pp. 868-876.

Introducing the Feature Article Authors

Christophe Calloz received the Dipl6me d'Ing~nieur en tlec-tricit6 and the PhD degrees from the Lausanne Swiss FederalInstitute of Technology (EPFL), Switzerland, in 1995 and 2000,respectively. From 2001 to 2004, he was a Postdoctoral ResearchEngineer at the Microwave Electronics Laboratory of University ofCalifornia at Los Angeles (UCLA). In June 2004, he joined EcolePolytechnique of Montreal, where he is now an AssociateProfessor, a member of the Microwave Research Center Poly-Grames, and a Canada Research Chair (CRC).

He has authored and coauthored over 250 technicalconference, letter, and journal papers, among which 35% wereinvited papers (around 50% of conference papers), and he holdsseveral patents. He has authored the first unified textbook onmetamaterials, Electromagnetic Metamaterials, Transmission LineTheory and Microwave Applications (John Wiley/IEEE Press). hInaddition, he has authored six book chapters. He has participated in25 courses, tutorials, and workshops around the world over the pastthree years, and he has organized several focused sessions andworkshops at international conferences.

Dr. Caloz is a Senior Member of the IEEE. He is a Memberof the IEEE Microwave Theory and Techniques Society, of theAntennas and Propagation Society, and of the TechnicalCoordinating Committee MUT-IS. He is also a speaker of theMTT-15 Speaker Bureau, and the Chair of Commission D(Electronics and Photonics) of the Canadian Union de RadioScience Internationale (URSI). He is a member of the EditorialBoard of the UNM, of the International Journal of RF andMicrowave Computer-Aided Engineering, and of the journalMetamaterials of the Metamorphose Network of Excellence. Hewas the guest editor of the 2006 special issue on metamaterials inthe International Journal for Numerical Methods (IINM), and theTPC Chair of the 2007 International Symposium of Signals,Systems and Electronics (ISSSE). He is the guest editor of the2009 special issue "Functional Nanophotonics and Nanoelectro-magnetics" of the Journal of Computational and TheoreticalNanoscience.

Dr. Caloz has received a number of awards, including theUCLA Chancellor's Award for Postdoctoral Research "for excep-tional accomplishments in research" in March 2004, and the IEEEMUT-S Outstanding Young Engineer Award 2007 with the citation

38 IEEE Antennas and Propagation Magazine, Vol. 50, No. 5, October 200838


"for pioneering contributions to the development of novel conceptsand applications of electromagnetic metamaterials."

His research interests include all fields of theoretical,computational, and technological electromagnetics engineering,with strong emphasis on emergent and multidisciplinary topics.Specifically, he is currently active in the following fields:electromagnetic metamnaterials, ferromagnetic nanostructures,resonant and leaky-wave passive and active smart antennas andreflectors, nonlinear and active devices, ultra-widebandtechnology, MIMO, FDTD for highly dispersive media, and time-domain integral techniques.

Tatsuo Itoh received the PhD degree in ElectricalEngineering from the University of Illinois, Urbana in 1969. FromSeptember 1966 to April 1976, he was with the ElectricalEngineering Department, University of Illinois. From April 1976 toAugust 1977, he was a Senior Research Engineer in the RadioPhysics Laboratory, SRI International, Menlo Park, CA. FromAugust 1977 to June 1978, he was an Associate Professor at theUniversity of Kentucky, Lexington. In July 1978, he joined thefaculty at the University of Texas at Austin, where he became aProfessor of Electrical Engineering in 1981 and Director of theElectrical Engineering Research Laboratory in 1984. During thesummer of 1979, he was a guest researcher at AEG-Telefunken,Ulm, West Germany. In September 1983, he was selected to holdthe Hayden Head Centennial Professorship of Engineering at theUniversity of Texas. In September 1984, he was appointedAssociate Chair for Research and Planning of the Electrical andComputer Engineering Department at the University of Texas. InJanuary 1991, he joined the University of California, Los Angeles,as Professor of Electrical Engineering and holder of the TRWEndowed Chair in Microwave and Millimeter Wave Electronics(currently Northrop Grumman Endowed Chair). He was anHonorary Visiting Professor at Nanjing Institute of Technology,China, and at Japan Defense Academy. In April 1994, he wasappointed as Adjunct Research Officer for the CommunicationsResearch Laboratory, Ministry of Post and Telecommunication,Japan. He currently holds a visiting professorship at the Universityof Leeds, UK. He has received a number of awards, including theShida Award from Japanese Ministry of Post andTelecommunications in 1998, Japan Microwave Prize in 1998,IEEE Third Millennium Medal in 2000, and IEEE MTT

Distinguished Educator Award in 2000. He was elected a memberof the US National Academy of Engineering in 2003.

Dr. Itoh is a Fellow of the IEEE, a member of the Institute ofElectronics and Communication Engineers of Japan, and a memberof Commissions B and D of USNC/URSI. He served as the Editorof the IEEE Transactions on Microwave Theory and Techniquesfor 1983-1985. He serves on the Administrative Committee ofIEEE Microwave Theory and Techniques Society. He was VicePresident of the Microwave Theory and Techniques Society in1989 and President in 1990. He was the Editor-in-Chief of IEEEMicrowave and Guided Wave Letters from 1991 through 1994. Hewas elected as an Honorary Life Member of the MUT Society in1994. He was the Chair of USNC/URSI Commission D from 1988to 1990, and Chair of Commission D of URSI for 1993-1996. Hewas Chair of the Long Range Planning Committee of URSI. Heserves on advisory boards and committees of a number oforganizations. He served as Distinguished Microwave Lecturer onMicrowave Applications of Metamaterial Structures of IEEEMTT-S for 2004-2006.

He has 370 journal publications, 700 refereed conferencepresentations, and has written 40 books/book chapters in the areaof microwaves, millimeter-waves, antennas, and numericalelectromagnetics. He has generated 68 PhD students.

Andre Rennings studied electrical engineering at Duisburg-Essen University, Germany. From 1999 to 2000, he was a visitingstudent at the Microwave Electronics Laboratory of the Universityof California at Los Angeles (UCLA). He received the Dipl.-Ing.and the Dr.-Ing. degree from the University of Duisburg-Essen in2000 and 2008, respectively. Since 2006, he has been with IMSTGmbH in Kamp-Lintfort, Germany, where he works mainly onmetamaterial antennas, Finite-Difference Time-Domain (FDTD)software, and RF components for magnetic-resonance imaging(MRI) systems. His general research interests include all fields oftheoretical, computational, and technological electromagneticengineering. He has authored 45 papers, one book chapter, andfiled three patents. He received the VDE prize for his diplomathesis, and was the recipient of the student paper competitionaward (second prize) presented at the 2005 IEEE Antennas andPropagation Society International Symposium, Washington, DC. 'Ii



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