linear technologlinear technology - analog devices · 2018. 3. 20. · linear technology magazine...

DECEMBER 1995 VOLUME V NUMBER 4

by Jaime Tseng

The LT1160 and LT1162:Medium Voltage,N-Channel BridgeDesign Made Easy

IN THIS ISSUE . . .

COVER ARTICLE

The LT®1160 and LT1162:Medium Voltage, N-ChannelBridge Design Made Easy........................................ 1Jaime Tseng

Editor’s Page ................... 2Richard Markell

LTC in the News .............. 2

DESIGN FEATURES

The LT1166: Power OutputStage Automatic BiasSystem Control IC ............ 3Dale Eagar

The LT1186: A SimpleBits-to-Light Converter forLCD Displays ................... 6Anthony Bonte

Power Factor Correction,Part Four: The BrainsBehind the Brawn ......... 10Dale Eagar

How Do You Slew 200V/µswith a 250µA Op Amp?...................................... 12George Feliz

DESIGN IDEAS.................................17–26(complete list on page 17)

New Device Cameos ....... 27

Design Tools .................. 28

Sales Offices ................. 28

any input condition. When bothinputs are high, both outputs areactively held low.

Bipolar versus CMOS DriversBipolar technology was chosen for

operation to 60V (absolute maximum)due to its inherent greater-than-60Vjunction-breakdown voltage. This isparticularly important in an N-chan-nel topside driver, which must operateabove the supply rail. In addition,bipolar technology provides consis-tent drive capability, resulting intransition times that increase moder-ately with large increases in capacitiveload. The LT1160 switches 1,000pFloads in 75ns, but for 10,000pF loads,this only increases to 180ns, makingoperation to 100kHz possible withthe largest MOSFETs. Figures 2ashows a simplified schematic of theLT1160 output stage; Figure 2b showshow it delivers progressively morecurrent as the capacitive load in-creases. On the other hand, CMOSdrivers’ switching times tend to befast at light loads but increase rapidlyat higher capacitive loads, as shownin Figures 3a and 3b.

IntroductionThe new LT1160 is a second-gen-

eration, half-bridge, N-Channel powerMOSFET driver. The LT1160, whichis derived from the LT1158 half-bridgedriver, has an increased voltage han-dling capability and a higher outputdrive current. The LT1162 is a full-bridge version of the LT1160

Like its predecessor, the LT1160effectively deals with the many prob-lems and pitfalls encountered in thedesign of high efficiency motor-con-trol and switching regulator circuits.This article will discuss these prob-lems, along with the solutions theLT1160 offers the bridge designer.

LT1160 OverviewFigure 1 is a block diagram of the

LT1160 (the LT1162 has two circuitsidentical to that in Figure 1). New inthe LT1160 are two independent un-dervoltage-detect circuit blocks thatdisable the drivers when either thesupply voltage or the voltage acrossthe bootstrap capacitor drops belowits undervoltage trip point. Two inputpins control the switching of bothN-channel MOSFETs; both channelsare noninverting drivers. The inter-nal logic prevents both outputs fromturning “ON” simultaneously under continued on page 14

, LTC and LT are registered trademarks of Linear Technology Corporation.Burst Mode, Bits-to-Nits and C-Load are trademarks of Linear Technology Corporation.

LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGY

2 Linear Technology Magazine • December 1995

EDITOR’S PAGE

by Richard Markell

On Being Analog in Silicon Valley

Authors can be contactedat (408) 432-1900

LTC in the News...Working in Silicon Valley is inter-

esting. The hottest technologies blowthrough, virtually overnight. Some-times the technologies take off, andsometimes they self-destruct. But weat LTC need to know what’s coming sothat we can design analog circuitrythat allows the fast-paced digital worldto talk to our linear world. The micro-processor gurus might dispute it, butthe expertise required to optimize thetransient response of a DC/DC con-verter is nontrivial. The IC designermust know a multiplicity of disci-plines, from transistor-level integratedcircuit simulation to the system-levelESR requirements of capacitors. Weat LTC pride ourselves on being thebest in the industry at what we do,whether it’s building applications cir-cuits for customers, designing IC’s ortaking an order in customer service.The design of ASICs, multimedia andnetworks may be hot today, cold to-morrow, but analog will always bewith us. We press on, growing at agood clip, supplying the analog coreto the world of digital designers. Weare always looking for people who arein love with analog circuit design.

Our feature article highlights theLT1160 and LT1162 N-channelMOSFET half- and full-bridge drivers,with increased maximum operatingvoltage (60V abs. max.) and higheroutput drive current. Also new tothese topologies is undervoltage lock-out, which disables the drivers wheneither the supply voltage or the volt-age across the bootstrap capacitordrops below its trip point. Cross-con-duction or “shoot-through” has beencompletely eliminated. The LT1160and LT1162 will see use as motor-control circuits, high current,high-efficiency switching regulatorsand other half- and full-bridge drivercircuitry.

Another “control” device is the newLT1186 Bits-to-Light converter forLCD displays. The LT1186 is a com-panion part to LTC’s LT1182/LT1183/LT1184 CCFL/LCD contrastswitching regulator family of devices.The LT1186 adds simple digital con-trol for CCFL to a fixed frequency,current mode switching regulator. Thedevice incorporates a micropower 8-bit, 50µA full-scale, current outputDAC that provides the “bits-to-lamp-current” control. The devicecommunicates in two modes: stan-dard SPI and pulse mode.

The LT1352 is a new micropoweroperational amplifier that can slew200V/µs and settle to 0.1% in 700nsfor a 10V step. The LT1352 is stablewith any capacitive load and can driveheavy loads (≤40mA) with low distor-tion. These specifications are achievedwithout compromising other specifi-cations over the part’s ±2.5V to ±15Vsupply range. Another Design Fea-ture spotlights the LT1166, a poweroutput stage automatic bias systemcontrol IC. The part is designed tooptimize amplifier output stage bias-ing (and quiescent current) and toeliminate the need for temperaturetracking, matching transistors or trimpots. The device allows automatic bi-asing of an amplifier’s output stage inthe Class AB operational region.

In this issue we present the fourthand concluding part of the “PowerFactor Correction” series. Also pro-vided with this issue is an index ofcircuits presented in the first fiveyears of the Linear Technology Magazine.

This issue also has the usual selec-tion of Design Ideas, carefully chosenand tested in our Applications Lab.Here we have circuits ranging from GTLterminators to rail-to-rail filters.

Once again Linear Technologyreports another quarter of recordsales and profits. Net salesfor LTC’s first quarter endingOctober 1, 1995 were a record$87,005,000, an increase of 50%over net sales of $58,082,000 forthe previous year. The companyalso reported record net incomefor the quarter of $30,520,000 or$0.39 per share, an increase of71% over $17,830,000 or $0.24per share reported for the firstquarter of the last year.

More good news was reported inInvestor’s Business DailySeptember 29th issue, whichfeatured a corporate profile undertheir New America section. BobSwanson talked about the con-tinuing need for analog circuitry.“We cheer the growth of digital, butwhen the rubber hits the road, welive in an analog world and wedon’t see this ever changing.”

In Financial World October 10,1995 issue Linear TechnologyCorporation was included on itslist of “America’s Best 100 GrowthCompanies.” LTC was ranked#22 out of all the companiesconsidered: #3 for semi-conductors and # l for analog!

In the “What’s Hot for Product”category, Electronic DesignOctober 13, 1995 issue high-lighted the LTC1410 in a featurearticle titled “12-Bit IC ADCsRelieve Error Budgets.” TheLTC1410 is the fastest 12-bitSuccessive Approximation (SAR)based ADC to date. The part usesa novel design technique toachieve excellent power and costperformance. The LTC1410 joinsLTC’s fast growing line of A to Dand D to A products for highperformance applications.

Linear Technology Magazine • December 1995 3

The LT1166: Power Output StageAutomatic Bias System Control IC

1166_1.eps

>4mA

>4mA

GATE

GATE

IN OUTINPUT OUTPUT

R2

R1 V1

V2

ISENSE

ISENSE

LT1166

V+

V–

Q2

Q1

1166_2.eps

V1

V2

V3

V+

V–

V3 IS ADJUSTED SO THAT V1 * V2 = 0.0004

WHEN THE OUTPUT CURRENT IS ZERO V1 = V2 = 20mV

Figure 1. Basic LT1166 circuit configuration Figure 3. LT1166 Voltage-control loop Figure 5. LT1166’s can be easily paralleled

1166_4.eps

MAGICAL OPERATING

POINTCROSSOVER DISTORTION

QUIESCENT POWER

BIAS00

IntroductionClass AB amplifiers are popular

because of their “near Class A” per-formance and their ability to operateon considerably less quiescent powerthan Class A. Class AB amplifiers areeasy to construct, rugged and reli-able. However there is an aspect ofthese amplifiers that can cause per-plexity, consternation and finally hairloss—their bias scheme. The problemis that the very parameter that makesClass AB so good, namely low quies-cent current, is poorly controlled. TheLT1166 offers control over the quies-cent current directly, removing thenecessity of temperature tracking,matching transistors or trim pots.

Functional DescriptionThe LT1166 (Figure 1) controls the

Class AB output stage by incor-porating two control loops, thecurrent-control loop and the voltage-control loop. The current-control loop(Figure 2) operates independently ofthe voltage loop while keeping the

set bias and the nerves to take thecatastrophic results of misalignment.The LT1166 removes all excess cross-over distortion caused by improperlyset quiescent current, while signifi-cantly reducing the distortion causedby the effects of nonlinear transcon-ductance in the output transistors.Figure 4 shows how the magic pointfor a Class AB amplifier is visualized(keep in mind that the magic point istemperature sensitive in many de-signs, leading the designer totemperature tracking tricks, whichare not needed with the LT1166.)

Makes RandomTransistor Pairs intoParallelable Modules

The LT1166, combined withexternal transistors, implements aunity-gain buffer with an offset volt-age of ±50mV. This buffer in turnbecomes a component that can easilybe paralleled to increase the outputpower (see Figure 5).

Terra Firma!California, a place where the

ground is anything but stable, is nowthe birthplace of “The Rock,” a virtualground with an attitude. The Rock(detailed in Figure 6) is a 5V, ±0.4%

1166_3.eps

V1

V2–

+V3 2

V3 2

OUT

by Dale Eagar

product of V1 and V2 constant. Thevoltage loop (Figure 3) maintains theoutput voltage at the input voltagelevel by driving both gates up or down.The two loops, although mutually in-dependent, act in harmony to providea component insensitive, tempera-ture insensitive, simple Class AB biasnetwork.

Crossover DistortionAdjusting the bias point of a Class

AB amplifier is quite like walking atight rope. Quiescent current needsto be set precisely; if it is adjusted toolow, the amplifier exhibits crossoverdistortion, whereas if it is adjustedtoo high, the amplifier becomes a veryeffective heater. Setting the bias of aClass AB amplifier is often reservedfor the “Medicine Man,” an individualwith both the smarts to know how to

DESIGN FEATURES

Figure 2. LT1166 Current-control loop

1166_5.eps

IN OUTPUT

LT1166

LT1166

Figure 4. “Magical” operating point ofClass AB amplifier


DESIGN FEATURES

low noise virtual ground. The Rockwill source or sink 1A of current withease. In this circuit, the current limitsare set to 1.5A for both sourcing andsinking currents. Changing the val-ues of R5 and R6 to 0.33Ω increasesthe output current by a factor of

three. This circuit, properly built, willeasily withstand an earthquake ofmagnitude 8, with unmeasurableoutput deviation.

Shaker-Table Driver/Brute Audio Amplifier

In the event that you don’t live inCalifornia, you might need the circuitshown in Figures 7a–7c, the 1.8kWshaker-table driver/audio amplifier.In Figure 7a, we develop a slice ofpower by connecting the LT1166 totwo power MOSFETs, two currentsense resistors and two current

1166_7a.eps

15VA*

12.5VA*

100V

1k

100Ω

1k

7.5k

–12.5VA*

–15VA*

–100V

100Ω

100Ω

CURRENT LIMIT ±6.8A

*FROM FIG. 7c

IRF9240

0.22

IRF230

0.22

OUT 300W RMS INTO 16Ω

1µF

1µF

IN2

1

8

9

3

6

5

4

2N3904

2N3906

GATE

GATE

LT1166

SENSE

ILIM

OUT

ILIM

SENSE

100Ω

1166_7B.eps

OUTPUT

0.22

0.22

0.22

0.22

TO FIG. 7c

0.22

0.22

–

+

1k

1k10k

9.1k

3

2

6

INPUT ±10Vp-p

1800W RMS INTO 2.6Ω

1250W RMS INTO 4Ω

625W RMS INTO 8Ω

EXTENDED POWER SUPPLY AV = 10 VOLTAGE AMP

POWER SLICES OF FIG. 7A

U1 LT1360

Figure 6. “The Rock,” a 5V, ±0.4% tolerance, ±1A low noise voltage source

Figure 7b. Paralleling power slices to increase output drive

sources. The power slice shown inFigure 7a will deliver 300W of sinewave power into 16Ω when poweredfrom ±100VDC. The MOSFETs and/or the sense resistors can be increasedor decreased to get output power inalmost any flavor.

Figure 7b details how six powerslices can be paralleled to put out acool, clean 1800W of Richter or 1250Wof rock and roll. The LT1360 amplifieroperates in the extended supply mode(see “Extending Op Amp Supplies toGet More Output Voltage,” LinearTechnology, IV:2, pp. 20–22).

Figure 7a. ±100V, 6A “power slice”

1166_6.eps

U1 LT1431

12V

12V

12V

R2 2k

R4 1k

R7 1k

100Ω

100Ω

R1 1k

Q1 IRFZ34

Q2 IRF9Z30

C3 270µF

5V OUTOUT

R5 1Ω 2W

R6 1Ω 2W

R3 100Ω

4

5 6

C1 1µF

C2 1µF

R8 100Ω

2.5V5k

5k

7 8 3 1

2IN

U2 LT1166

TOP DRIVE

TOP SENSE

TOP ILIM

BOTTOM ILIM

BOTTOM SENSE

BOTTOM DRIVE

1

8

7

3

6

5

4

+


1166_9.eps

TRIM R. POTENTIOMETER

1166_7c.eps

OUTPUT OF FIG. 7B

LT1004 – 2.5

LT1004 – 2.5

– 15VA*

*TO FIG. 7A

15VA*

15V

15V

1k

R1 390Ω

R2 390Ω

LT1360

4

7

C2 3300pF

C10.1µF

12.5VA*

– 12.5VA*

U1 OF FIG. 7B

R3 1k

DESIGN FEATURES

Figure 7c. Connection diagram for the ±15V floating supply

Figure 8a. Ring-tone generator schematic diagram

1166_8A.eps

0.33µF

11k 10k 10k7

0.22

0.22 RING-TONE OUTPUT

300W/SLICE

POWER SLICES OF FIGURE 7a

2.5VA*

51k 51k

–2.5VA*

VSS

R VCC

CMOS 555

0.01µF6

2

3

*TO FIG. 7c

1 0.15µF

0.047

120Hz

84

0.01µF360k

6

5

OUT3

2

360k

–

+LT1078

–

+LT1078

TO FIG. 8b

1166_8B.eps

LT1004 – 2.5

LT1004 – 2.5

15VA

–15VA

15V

FROM FIGURE 8a

15V

1k

1k

4

12.5VA

–12.5VA

LT10780.1µF

8

LT1004 – 2.5

LT1004 – 2.5

7.5k

7.5k

3.3k

0.1 100V

– 2.5VA

2.5VA

Figure 9. Trim pots are no longer required

ConclusionWith the advent of the LT1166,

biasing Class AB amplifiers becomestrivial. Dependencies on temperaturetracking, component matching andalignment pots have succumbed tocool, clean, closed-loop control ofoperating points. Suddenly, MOSFETslook a lot friendlier to the power am-plifier designer. Suddenly, pots havebitten the dust.

Finally, Figure 7c shows how toconnect the ±15V floating supply andalso details the PSRR snubber for theLT1360 (R1, R2, R3 and C2).

Ring-Tone GenerationBuilding upon an earlier Linear

Technology Magazine article (seeabove) we show a power ring-tonegenerator. This generator is detailed

in Figures 7a, 8a and 8b. This circuitstarts at 300W (or less if you usesmaller FETs and bigger resistors inFigure 7a) and can be stacked up toany desired power level. With thiscircuit, you could easily build a ringercapable of ringing the phones of everyU.S. politician simultaneously (notthat this would do any good).

Figure 8b. Power supply circuitry forFigure 8a


The LT1186: A Simple Bits-to-LightConverter for LCD Displays by Anthony Bonte

IntroductionMany current-generation portable

and handheld instruments use back-lit liquid crystal displays (LCDs).Cold-cathode fluorescent lamps(CCFLs) provide the highest availableefficiency for backlighting the dis-play. These lamps require high voltageAC to operate, mandating an efficienthigh voltage DC/AC converter. In ad-dition to good efficiency, the convertershould deliver the lamp drive in theform of a sine wave. The sine waveexcitation helps to minimize RF emis-sions and also provides optimalcurrent-to-light conversion in thelamp. The circuit should also permitlamp intensity control from zero tofull brightness with no hysteresis or“pop-on.”

Manufacturers offer an array ofmonochrome and color displays.These displays vary in size, operatinglamp current, lamp-voltage range andpower consumption. The small sizeand battery-powered operation asso-ciated with LCD-equipped apparatus

dictate low component count and highefficiency. Size constraints place limi-tations on circuit architecture, andlong battery life is a priority. All com-ponents, including pc board andhardware, usually must fit within theLCD enclosure, with a height restric-tion of 5mm–10mm.

Handheld instruments are gener-ally characterized by dedicatedfunctionality and reduced processingpower in comparison with notebookor laptop computers. These portablesystems often use general-purpose,low cost microcontrollers such as theIntel 80C51 or Motorola 68HC11.These microcontrollers are popularbecause of their versatile I/O capabil-ity. A simple interface that allows themicrocontroller to easily control theoperating lamp current and displaybrightness is required. To addressthe requirements of these portableinstruments, Linear Technologyintroduces the LT1186 DAC program-mable CCFL switching regulator.

BAT 8V TO 28V

16

15

14

13

12

11

10

9

ICCFL

DIO

CCFL VC

AGND

SHDN

CLK

CCFL VSW

BULB

BAT

ROYER

VCC

IOUT

DOUT

CCFL PGND

CS

LT1186

LAMP

UP TO 6mA

10 6

L1

R1 750Ω

L2 100µH

3 2 1 54

+

+

C7, 1µF

C1* 0.068µF

C5 1000pF

R2 220k

R3 100k

C3A 2.2µF 35V

C4 2.2µF

TO MPU

FROM MPU

VIN 3.3V OR 5V

1186_1.eps

C3B 2.2µF 35V

C2 27pF 3kV

+

Q2* Q1*SHUTDOWN

ALUMINUM ELECTROLYTIC IS RECOMMENDED FOR C3A AND C3B. MAKE C3B ESR ≥ 0.5Ω TO PREVENT DAMAGE TO THE LT1186 HIGH-SIDE SENSE RESISTOR DUE TO SURGE CURRENTS AT TURN-ON C1 MUST BE A LOW LOSS CAPACITOR, C1 = WIMA MKP-20

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

0µA TO 50µA ICCFL CURRENT GIVES 0mA TO 6mA LAMP CURRENT FOR A TYPICAL DISPLAY.

FOR ADDITIONAL CCFL/LCD CONTRAST APPLICATION CIRCUITS, REFER TO THE LT1182/83/84/84F DATA SHEET

D1 BAT85

DIN

FROM MPU

FROM MPU

1

2

3

4

5

6

7

8D1

1N5818

L1 = COILTRONICS CTX210605

L2 = COILTRONICS CTX100-4

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876

New LTC CCFL PartThe LT1186 is a fixed frequency,

current mode switching regulator thatprovides a simple digital control func-tion for cold-cathode fluorescentlighting. The regulator is a compan-ion part to Linear Technology’sLT1182/LT1183/LT1184 CCFL/LCDcontrast switching regulator family.The LT1186 supports grounded-lampand floating-lamp configurations. TheIC includes an efficient high currentswitch, an oscillator, output-drive logic, control circuitry and amicropower 8-bit 50µA full-scale cur-rent output DAC.

The DAC provides simple“bits-to-lamp-current control” andcommunicates in two interface modes:standard SPI mode and pulse mode.On power-up, the DAC counter resetsto half-scale and the DAC configuresitself to SPI or pulse mode, dependingon the CS (chip select) signal level. InSPI mode, the system microprocessorserially transfers the present 8-bitdata and reads back the previous

Figure 1. Up to 90% efficient floating CCFL with SPI control of lamp current

DESIGN FEATURES


returned to the user to verify the lastprogrammed current level. An inputof one corresponds to 1 LSB of DACoutput current (195.3nA) and an in-put of 255 corresponds to the full-scaleDAC output current of 50µA.

Listings 1 and 2 on pages 8–9 arethe complete C and assembly languagecode that controls the evaluation setupof Figure 2. My thanks and apprecia-tion go to Tommy Wu at LinearTechnology for developing this soft-ware. Any questions or commentsregarding the software should be ad-dressed to Tommy, in care of thispublication. The software is fairly ge-neric and is easily adaptable to othermicrocontrollers or microprocessors.

ConclusionThe LT1186 is a novel Bits-

to-NitsTM* converter for drivingcold-cathode fluorescent lamps inhandheld or portable instrumentswith LCD displays. Lamp current isdigitally programmed in either of twointerface modes: standard SPI modeand pulse mode, which providessimple push-button up/down con-trol. The LT1186 represents anotherstep in Linear Technology’s commit-ment to providing efficient, compactand economical backlight solutions.

connected between the BAT and Royerpins, provides feedback, and a singlecapacitor (C7) on the VC pin providesstable loop compensation.

Lamp current is programmed bydriving the ICCFL pin from the IOUT pin.The transfer function between lampcurrent and input programming cur-rent must be empirically determinedand is dependent on a myriad of fac-tors, including lamp characteristics,display construction, transformerturns ratio and the tuning of theRoyer oscillator. The LT1186 controlcircuitry is powered by either a 3.3Vor 5V logic supply, and the inputbattery voltage range for this exampleis 8V to 28V. Lower input batteryvoltage operation is accomplished byincreasing the turns ratio of trans-former L1 and retuning the Royer bytweaking the values of C1 and C2.Electrical efficiency with typical CCFLlamps ranges between 80%–90% atfull load with lamp current being vari-able from 200µA to 6mA for a typicaldisplay. The component values cho-sen are an interactive compromisein maximizing photometric outputversus input power.

Figure 2 is a block diagram for anIntel 80C31 microcontroller-basedevaluation setup for the LT1186. Thelamp current is controlled by a gen-eral PC keyboard interface. The userinputs a number between 0 and 255to set a new input programming cur-rent, and therefore, a new lampcurrent level. The previous code is

8-bit data from the DAC. In pulsemode, the upper six bits of the DACare configured for increment-only(1-wire interface) or increment/dec-rement (2-wire interface) operation,depending on the DIN signal level.

The LT1186 control circuitry oper-ates from a logic supply voltage of3.3V or 5V. The IC also has a batterysupply voltage pin that operates from4.5V to 30V. The LT1186 draws 6mAtypical quiescent current. An activelow shutdown pin reduces total sup-ply current to 35µA for standbyoperation and the DAC retains its lastsetting. A 200kHz switching frequencyminimizes magnetic component size,and the use of current mode switch-ing techniques gives high reliabilityand simple loop frequency compen-sation. The LT1186 is available in a16-pin, narrow-body SO.

Typical ApplicationFigure 1 is a complete floating CCFL

circuit using the LT1186 in standardSPI mode to control operating lampcurrent from a microprocessor. Afloating-lamp circuit allows thetransformer to provide symmetric, dif-ferential drive to the lamp. Balanceddrive eliminates the field imbalanceassociated with parasitic lamp-to-frame capacitance and reduces“thermometering” (uneven lamp in-tensity along the lamp length) at lowlamp currents.

Q1, Q2, L1 and C1 form the core ofa current-fed, Royer-class converter.The LT1186 power switch and L2create a switch-mode current sourceto drive the Royer. The 200kHz switch-ing frequency allows the use of only100µH for L2. Base drive for Q1 andQ2 is provided by L1’s tickler windingand R1. R1 is selected to guaranteesufficient base drive with minimumbetas for Q1 and Q2. R2, R3 and C5comprise an external divider networkfor the open-lamp protection circuit.The divider monitors the voltageacross the Royer converter and com-pares it with an internal 7V clampvoltage between the BAT and BULBpins. Monitoring the primary-sideconverter current through a high-side sense resistor and amplifier,

CS

CLK

DIN

DOUT

5 A0-A15

D0-D7

4

RS232 FROM PC

1

2

8

7

9

10

31

P1.4

P1.3

P1.0

P1.1

TX

RX

LT1186

ROYER CONVERTER

LAMP

ROM

80C31

1186_2.eps

Note:

* Bits-to-Nits is a trademark of Linear TechnologyCorporation. 1 Nit = 1 Candela/meter2. Nit isderived from the Latin word nitere meaning “toshine.”

DESIGN FEATURES

Figure 2. LT1186 Floating CCFL system using an Intel 80C31 microcontroller


Listing 1. LT1186.C

#include <stdio.h>#include <reg51.h>#include <absacc.h>

extern char lt1186(char); /* external assembly function in lt1186a.asm */sbit Clock = 0x93;

main()

int number=0;int LstCode;

Clock = 0;

TMOD = 0x20; /* Establish serial communication 1200 baud */TH1 = 0xE8;SCON = 0x52;TCON = 0x69;

while(1) /* Endless loop */

printf(“\nEnter any code from 0 - 255: “);scanf(“%d”,&number);if((0>number)|(number>255))

number = 0;printf(“The number exceeds its range. Try again!”);

else

LstCode = lt1186(number);printf(“Previous # %u”, (LstCode&0xFF)); /* AND the previous

number with 0xFF to turn off sign extension */

number = 0; /* Reinitialize number */

C and Assembly Language Program Code for LT1186 Evaluation

The LT1186.C program demon-strates the use of the LT1186DAC programmable CCFL switch-ing regulator.

The example circuit usesthe popular 80C51 family ofmicrocontrollers to interface betweenthe user and the LT1186.

The LT1186 DAC algorithm iswritten in assembly language,and is contained in a file namedLT1186A.ASM; it is called as a func-tion from the MAIN function ofLT1186.C. The user inputs an integerfrom 0 to 255 on a keyboard and theLT1186 adjusts the IOUT programmingcurrent to control the operating lamp

current and the brightness of theLCD display. The assembly languageprogram, LT1186A.ASM, receivesthe DIN word from the main C pro-gram function, lt1186( ). Assemblyto C interface headers, declarationsand memory allocations are listedbefore the actual assembly code.

DESIGN FEATURES


DESIGN FEATURES

Listing 2. LT1186A.ASM

; Port p1.4 = CS; Port p1.3 = CLK; Port p1.1 = Dout; Port p1.0 = Din;NAME LT1186_CCFLPUBLIC lt1186, ?lt1186?BYTE

?PR?ADC_INTERFACE?LT1186_CCFLSEGMENT CODE?DT?ADC_INTERFACE?LT1186_CCFLSEGMENT DATA

RSEG ?DT?ADC_INTERFACE?LT1186_CCFL?lt1186?BYTE: DS 2

RSEG ?PR?ADC_INTERFACE?LT1186_CCFL

CS EQU p1.4CLK EQU p1.3DOUT EQU p1.1DIN EQU P1.0

lt1186: setb CS ;set CS high to initialize the LT1186mov r7, ?lt1186?BYTE ;move input number(Din) from keyboard to R7mov p1, #01h ;setup port p1.0 becomes inputclr CS ;CS goes low, enable the DACmov a, r7 ;move the Din to accumulatormov r4, #08h ;load counter 8 countsclr c ;clear carry before rotatingrlc a ;rotate left Din bit(MSB) into carry

loop: mov DIN, c ;move carry bit to Din portsetb CLK ;Clk goes high for LT1186 to latch Din bitmov c, DOUT ;read Dout bit into carryrlc a ;rotate left Dout bit into accumulatorclr CLK ;clear Clk to shift the next Dout bitdjnz r4, loop ;next data bit loopmov r7, a ;move previous code to R7 as character returnsetb CS ;bring CS high to disable DACretEND

;NOTE: When CS goes low, the MSB of the previous code appears at Dout


Power Factor Correction, Part Four:The Brains Behind the BrawnThe Control Room

In the three previously publishedparts of our series on power factorcorrection, we investigated theworkings of the “power transfermechanism” as used in the ever sopopular “boost-mode power factorcorrection conditioner.” This powertransfer mechanism is shown in Fig-ure 1 as the “engine room.” Also shownis the “control room,” where the deci-sions are made and actions taken tomake the PFC power conditioner be-have like a resistor (the ultimatedisguise for a switching power sup-ply). Having had a brief tour of theengine room in an earlier article,we can summarize its workings in afew rules:1. The amount of power intercepted

from the input line is related tothe duty factor of the boostengine.

2. When the duty factor is at 0%, nopower is intercepted from theinput line.

3. Under all conditions, increasingthe duty factor will increase theamount of power interceptedfrom the input line.

4. The boost engine is essentiallylossless, so all power interceptedfrom the input line ends up inthe load.

5. The load voltage is to be heldconstant.

6. The load voltage shall be setabove the peak input voltage,where the peak input voltage is1.414 times the maximum RMSinput power-line voltage; this iswhere the power factor is to becorrected.If the peak input voltage exceeds

the output voltage at any time, theoutput diode (internal to the boostengine) takes it upon itself to forwardconduct, transferring power from theinput to the output without any

by Dale Eagar

EMI FILTER

AC BRIDGE

BOOST ENGINE LOAD

POWER PATH (ENGINE ROOM)

AC POWER

LINE

00

1020

3040 50 60

7080

90100

CONTROL ROOM

pfc4_1.eps

+

–

~

~

INPUT VOLTS

DUTY FACTOR

OUTPUT VOLTS

INPUT AMPS

0 0

Figure 1. Boost PFC block diagram

DESIGN FEATURES

controller to watch the input voltsand amps while adjusting the dutyfactor. The duty factor is constantlyadjusted to make the input ampsequal to some constant (K) times theinput volts. Presto resistor! Lookinginto the input terminals, one wouldsee a pure resistor. The input wouldlook resistive at a value of ZIN = 1/K.This impedance would be fairly con-stant for frequencies from DC to thefastest frequency at which the con-troller could accurately keep up withthings. In the real world there is thisguy named Nyquist who says, amongother things, that the controller wouldmake serious blunders if he were totry to control the duty factor fasterthan 1/2 of the switching frequency.With the switching frequency set tosomewhere around 100kHz, the in-put looks resistive from DC to manykiloHertz, which is good enough forthe 60Hz line and many of its nastyharmonics.

Careful inspection of the outputwould reveal an output impedance ofVOUT

2/K, a big number. The wholesystem would look like a constantpower source, with the power level setto K times VIN

2. These observationsare entered in the first row of Table 1.

authorization from the control room.Pure Insubordination! The only wayto prevent this disgraceful act bythe output diode is to keep the out-put voltage out of reach of the peakinput voltage.

At the ConsoleIn the control room is a control

console, a panel with meters and aduty factor control. Sitting in the con-trol seat is the controller, an intelligententity. The controller is given the taskof watching the three meters (Figure1) and controlling the duty factor tokeep things in check.

The controller is given two tasks toperform, namely:1. Make the input look resistive.2. Keep the output voltage constant

at 386V, regardless of the loadcurrent or input line voltage.Careful inspection of the above two

tasks may surprise you. Let’s look ateach task separately.

Task One: Making theInput Look Resistive

It would not be too difficult to teachthe controller to make the input lookresistive. One would need to teach the


Task-2: Keeping the OutputVoltage Constant at 386V

Looking at task-2 in the absence ofany task-1 constraints, we teach ourcontroller to adjust the duty factor tokeep the output voltage constant, re-gardless of the load current or inputline voltage. Having visualized such asystem, we can look at its character-istics. Looking into the input port atany constant load current, we see anegative input impedance. (As theinput line voltage goes up, the inputcurrent goes down.) Looking into theoutput port we see an impedance ofzero. (Any finite increase in outputcurrent causes no change in outputvoltage.) Finally, looking at the powerintercepted from the input line wehave a constant at VOUT × IOUTregardless of input voltage. These ob-servations occupy row two of Table 1.

Putting Them TogetherIt should be apparent that no de-

vice can exhibit positive impedanceand negative impedance at the sametime. However, we know how tocheat—we take the reciprocal of timeto get frequency.

This trick doesn’t need smoke andmirrors, unless, of course, you hookit up wrong, in which case you getseven years of bad luck. This is a trickin the frequency domain.

The first step is to loosen up theoutput specification a little. We cansee that the power delivered from asine wave into a resistive load is any-thing but constant (see Figure 2). We

at low input currents where inputripple is less critical.

The boost converter PFC providesno isolation from the line, has highinrush current, has a very high out-put voltage, has 120Hz ripple on theoutput and exhibits slow load-stepresponse. All of these difficulties ofthe boost converter are small com-pared to the problems in the design ofan input EMI filter if other technolo-gies were used.

Isolation, hold-up time and regula-tion are performed by the switchingpower supply that follows the PFCconditioner. Inrush protection is pro-vided by a negative tempco thermistoror other external circuitry.

It is advantageous to synchronizethe PFC conditioner and the followingswitching power supply 180 degreesout of phase to reduce the ripplecurrent on the output capacitor andreduce overall system noise. The Lin-ear Technology PFC family of parts allhave synchronizing ability, and theLT1508 and LT1509 controllers haveboth a PFC conditioner and a switch-ing power converter on one chip, fullysynchronized at 180 degrees.

The input impedance of a function-ing PFC will be negative from DC toabout 10Hz; from there the imped-ance goes reactive, then to positiveresistance at frequencies above 10Hz.The output impedance of the PFCconditioner is near zero ohms fromDC to about 10Hz; above 10Hz theoutput impedance looks like a vari-able resistor in parallel with the outputcapacitor’s reactance.

Table 1. Task-1 and task-2 characteristics

Input Output PowerTask impedance impedance transferred

Make InputResistive 1/K V2

OUT/K K VINConstant VOUT Negative Zero VOUT × IOUT

continued on page 25

DESIGN FEATURES

would have to have an infinitely largeoutput capacitor to realize zero out-put ripple when there is input ripplecurrent. Infinite capacitors went outof vogue back in the Cretaceous era.First we loosen up the output voltagespecification to allow a few volts ofripple, bringing the output capacitordown to a realistic size. Second, wefurther loosen the output specifica-tion to allow the output voltage toswing tens of volts during heavy loadsteps.

We do this by teaching the controllerto keep the output voltage at 386Vwhen averaged over any 3-secondinterval. This is in effect telling thecontroller to make the input imped-ance negative for frequencies from DCup to several Hertz. We are instructingthe controller to adjust the K value inthe task-1 operation slowly to keep theoutput voltage at an average of 386V.See Figure 3 for the input impedanceversus frequency characteristics.

Summary/HindsightPower factor correction is incorpo-

rated into a product design for one ofseveral reasons:A. Government standards, such as

IEC555, compliance with whichis required to sell product inseveral worldwide markets.

B. The marketing boys upstairs feelthat they can sell it as a feature.

C. Management needs to keep thepower supply design group out oftrouble until the next majordesign.The boost converter is the best way

to perform PFC at any power greaterthan 50W. The crowning advantage ofthe boost topology is its continuousinput current when running in thecontinuous mode. The loss ofcontinuity of input current in thediscontinuous mode is unimportant,as discontinuous mode only happens

VOLTS WATTS AMPS

INPUT POWER

INPUT VOLTS AND AMPS

TIME

Figure 2. Input power waveform

POSITIVE

RIN

NEGATIVENOTE: WHEN THE INPUT RESISTANCE IS AT ZERO OHMS THE INPUT STILL HAS IMPEDANCE BECAUSE AT THAT POINT THE INPUT LOOKS REACTIVE.

FREQUENCY

0

0

10Hz

pfc4_3.eps

Figure 3. Input impedance (resistivecomponent) versus frequency


FREQUENCY (Hz)

0.0001

0.001

0.010

0.1

THD+

N (%

)

20K

1352_2.eps

100 1K 10K

AV = 1

AV = –1

How Do You Slew 200V/µswith a 250µA Op Amp? by George Feliz

IntroductionThe LT1354/LT1365 family of low

power, high-slew-rate op ampschanged the relationship between slewrate and supply current found in tra-ditional voltage-feedback operationalamplifiers. These parts slew from400V/µs with 1mA supply current upto 1,000V/µs with 6mA supply cur-rent. The LT1352 dual op amp is aprecocious new sibling that extracts200V/µs from a mere 250µA of supplycurrent (see Figure 1).

The new design shares the familycircuit topology, but adds a frugaloutput stage that enthusiasticallydrives heavy loads with up to 40mA ofcurrent. A primary benefit, unheardof with other low power op amps, isthe ability to drive large signals intoheavy loads with low distortion. A plotof distortion for 20V peak-to-peaksignals with a 1kΩ load is shown inFigure 2. Table 1 shows that thisremarkable output performance isachieved without compromising otherspecifications over its 2.5V to 15Vsupply voltage range. The settling timefor a 10V step is only 700ns for 0.1%and 1250ns for 0.01%. In addition toperformance, the LT1352 is a user-friendly C-Load op amp, which isstable with any capacitive load.

output swing and/or create instabil-ity with a capacitive load.

The solution used on the LT1352was to create a pair of compositetransistors formed by transistorsQ19–Q21 and Q22–Q24. The currentmirrors attached to the collectors ofemitter followers Q19 and Q22 pro-vide additional current gain. The ratioof transistor geometries Q20 to Q21and Q23 to Q24, and resistor ratiosR2 to R3 and R4 to R5 set the gain tofive at low currents and to a maxi-mum of 23 at high currents. Thevariation in gain is required in orderto have low quiescent current yet beable to provide output drive on de-mand. There is no output-swing losswith this output stage, as the swing islimited at the high impedance node.The dynamics of a composite are notas benign as an emitter follower, socompensation is required and is pro-vided by R6, C1 and R7, C2.

The C-Load stability is provided bythe RC, CC network between the out-put stage and the high impedancenode. When the amplifier is driving alight or moderate load the output canfollow the high impedance node andthe network is bootstrapped and hasno effect. When driving a heavy loadsuch as a big capacitor or small-value

How We Did ItA simplified schematic of the cir-

cuit is shown in Figure 3. The circuitlooks similar to a current-feedbackamplifier, but both inputs are highimpedance, as in a traditional voltage-feedback amplifier. A complementarycascade of emitter followers (Q1–Q4)buffers the noninverting input anddrives one side of resistor R1. Theother side of the resistor is driven byQ5–Q8, which form a buffer for theinverting input. The input voltageappears across the resistor, generat-ing currents in Q3 and Q4, which aremirrored by Q9–Q11 and Q13–Q15into the high impedance node. Tran-sistors Q17–Q24 form the outputstage. Bandwidth is set by R1, thegm’s of Q3, Q4, Q7 and Q8, and thecompensation capacitor CT.

The current available to slew CT isproportional to the voltage that ap-pears across R1. This method of “slewboost” achieves low distortion due toits inherent linearity with input stepsize. Since large slew currents can begenerated without increasing quies-cent current, CT can be increased andR1 can be decreased. Lowering R1reduces the input noise voltage to14nV/√Hz and helps reduce inputoffset voltage and drift.

The output stage buffers the highimpedance node from the load byproviding current gain. The LT1352and its relatives are single-gain-stageamplifiers in order to remain stablewith capacitive loading and to achievetheir high slew rate with low quies-cent supply current. The simplestoutput stage would be two pairs ofcomplementary emitter-followers,which would provide a current gain ofBetaNPN × BetaPNP. Unfortunately, thisgain is insufficient for driving even a2kΩ load when running at the lowcurrent levels of the LT1352. Addinganother emitter-follower can reduce

DESIGN FEATURES

Figure 2. THD + noise, 20VP–P, RL = 1kΩFigure 1. Large-signal response, Av = −1


OUT+IN–IN

V+

V–

R1 1k

CCRC

CT

C1

C2

R6

R7

R2

R4 R5

R3

Q14Q23

Q1Q2

Q3

Q4

Q7

Q8

Q6Q5

Q10

Q9

Q22

Q20

Q12

Q21

Q17

Q13

Q18

Q19

Q16Q24

Q15

Q11

1352_3.eps

Table 1. Important specifications for the LT1352 at 25°C

Parameter Conditions Minimum Typical Maximum Units

VOS VS = 5V, ±15V 300 600 µVVS = ±2.5V 400 800 µV

IB VS = ±2.5V to ±15V 20 50 nAIOS VS = ±2.5V to ±15V 5 15 nAAVOL VS = ±15V, 1kΩ 20 60 V/mV

VS = ±5V, 1kΩ 20 60 V/mVVS = ±2.5V, 5kΩ 20 60 V/mV

GBW VS = ±15V 2.2 3 MHzVS = ±5V 2.0 2.7 MHz

VS = ±2.5V 2.5 MHzSR VS = ±15V 125 200 V/µs

VS = ±5V 40 50 V/µstSETTLE 10V step, 0.1% 700 ns

10V step, 0.01% 1250 nsCMRR Vs = ±15V 80 97 dB

VS = ±5V 78 84 dBVS = ±2.5V 68 75 dB

PSRR VS = ±2.5V to ±15V 90 106 dBNoise voltage VS = ±2.5V to ±15V,10kHz 14 nV/√HzNoise current VS = ±2.5V to ±15V,10kHz 0.5 pA/√HzOutput Swing VS = ±15V, 1kΩ 12.0 13.0 V

VS = ±5V, 500Ω 3.1 3.5 VVS = ±2.5V, 2kΩ 1.3 1.7 V

ISUPPLY VS = ±15V 250 325 µAVS = ±5V 220 290 µA

Figure 3. LT1352 simplified schematic

resistor, the network is incompletelybootstrapped and adds to the com-pensation at the high impedance node.The added capacitance provided byCC slows down the amplifier and thezero created by RC adds phase margin,ensuring stability. The small-signal

DESIGN FEATURES

Figure 4. Small-signal step response, Av = −1,CL = 500 pF

Figure 5. Large-signal step response, Av = 1,CL = 10,000 pF

response with a 500pF load is shownin Figure 4 and the large-signal re-sponse with 10,000pF is shown inFigure 5. Note that the slew rate inFigure 5 is limited to 4V/µs by theshort-circuit current limit of 40mA.

ConclusionThe LT1352 is a dual op amp with

an unmatched combination of lowsupply current, large-signal perfor-mance, stability and DC precision. Ifa figure of merit equal to the slew ratedivided by supply current is calcu-lated, the LT1352 stands alone amongvoltage-feedback op amps with800V/µs-mA.


1160_2.eps

+

VOUT

IOUT

CLOAD = 1000 - 9000pF

1160_1.eps

V+V+ BOOST

IN TOP DC-100kHz

IN BOTTOM DC-100kHz

UV OUT

BOTTOM UVLOCK

TOP GATE DR

BOTTOM GATE DR

CBOOTSTRAP

DRIVE

FEEDBACK

TO LOAD

SOURCE

DRIVE

FEEDBACK

HVTOP UVLOCK

–

+

–

+

CONTROL LOGIC

Figure 2a. Bipolar driver topology

The best solution to this problem isto use a floating topside N-channelgate driver operating from a boot-strap capacitor. Although widely used,the technique of bootstrapping mustbe carefully implemented, sincethe gate voltage applied to the topMOSFET derives directly from the

voltage on the bootstrap capacitor. Ifthis voltage becomes too low, it cancause problems for the MOSFET. Forexample, if at high duty cycles theoutput is not swinging low enough tofully recharge the capacitor, thetopside gate drive will be starved,leading to overdissipation.

Life Above the Supply RailThe first problem designers face in

N-channel half-bridge circuits is de-veloping the topside gate drive, whichmust swing at least 10V above thesupply rail for standard MOSFETs.This challenge alone is so demandingthat designers often resort to moreexpensive and less efficient P-chan-nel MOSFETs on the topside in lowvoltage bridge circuits. However, eventhis solution is no longer straight-forward when the supply voltageexceeds the maximum gate-to-sourcevoltage (VGS) rating for the MOSFET(typically 20V).

Figure 1. LT1160 block diagram with pins shown in actual package sequenceLT1160, continued from page 1

SWITCHING TIME (ns)0

0V

V OUT

15V

600 800 1000 1200 1400 1600400200

1000pF 9000pF1000pF

9000pF


–1.2A

I OUT

0A

1.2A

1160_2b.eps

600 800 1000 1200 1400 1600400200

1000pF1000pF

9000pF

9000pF

Figure 2b. Bipolar driver switching time increases only 150ns for an increase of 8,000pF inload capacitance

DESIGN FEATURES


In the LT1160, a floating under-voltage detection circuit monitors thevoltage across the bootstrap capaci-tor and a separate undervoltagedetection circuit monitors the supplyvoltage. If, as a result of operation athigh duty cycles, the voltage acrossthe bootstrap capacitor drops belowapproximately 8.7V, the top driver ispulled low to prevent overdissipatingthe top MOSFET, and also to allowthe capacitor to be recharged. Simi-larly, if the supply voltage drops belowabout 8.3V, both drivers are pulledlow to prevent overdissipating eitherMOSFET. The LT1162 has separateand independent undervoltage detec-tors for the two half-bridge drivers.

Shoot-Through UnwelcomeProbably the most frustrating ele-

ment to address in a synchronous

The most common techniques relyon fixed delay times to create a dead-time between conduction of the topand bottom MOSFETs. Although thiscan work, the delay time must takeinto account temperature changes,supply variations, and production tol-erances of the MOSFETs and other

MOSFET drive circuit is the timingbetween the top and bottom drives toprevent cross-conduction. Thepresence of cross-conduction or“shoot-through” currents always sapsefficiency and erodes precious ther-mal margins; in extreme cases it canlead to catastrophic failure.

Figure 4. 50W high efficiency switching regulator illustrates the design ease afforded by adaptive deadtime generation

1160_3a.eps

VINVOUT

IOUT CLOAD = 1000 - 9000pF+

Figure 3a. CMOS driver topology

1000pF


0V

V OUT

15V

600 800 1000 1200 1400 1600400200

9000pF

1000pF

9000pF


–1.2A

I OUT

0A

1.2A

1160_3b.eps

600 800 1000 1200 1400 1600400200

1000pF1000pF

9000pF

9000pF

Figure 3b. CMOS driver switching time increases 400ns for an increase of 8,000pF in loadcapacitance

1160_4.eps

1

2

3

4

5

6

7

1

2

3

4

5

6

78

16

15

14

13

12

11

10

9

14

1N4148

13

12

11

10

9

(2) IRFZ44

IRFZ34

L 47µH

RS 7mΩ

5400µF LOW ESR

BOTH GNDS JOINED AT ONE POINT

5V

MBR340

8

12V

60V (MAX)

L: HURRICANE LABS HL-KM147U 801 6352003 RS: DALE TYPE LVR-3 ULTRONIX RCS01

+

+

+

2.2µF

1µF

4700pF

f = 40kHz

+

18kΩ

0.1µF

0.1µF

2 x 2200µF LOW ESR

10µF

6800pF

100pF

25kΩ

1kΩ

500kΩ

2kΩ

10kΩ

18kΩ

10kΩ

1kΩ

2N2222

5kΩ

330kΩ

+ +

+

1N4148

1N4148

LT1846

BOOST

T GATE DR

T GATE FB

T SOURCE

P V+

B GATE DR

B GATE FB

S V+

IN TOP

IN BOT

UV OUT

S GND

P GND

NC

LT1160

DESIGN FEATURES


losses. Figure 5 shows the operatingefficiency for Figure 4’s circuit.

Two Halves Equal a WholeThe LT1162 is functionally and

electrically identical to two LT1160s.The LT1162 is ideal for motor-controlapplications driving an N-channelMOSFET H-bridge, as shown in Fig-ure 6. The DC motor shown can bedriven in both directions. By applyingthe appropriate control signals at theinputs, the motor can be made to runclockwise, counterclockwise, stop rap-idly (“plugging” braking) or free run(coast) to a stop. A very rapid stop canbe achieved by reversing the current,though this requires more carefuldesign to stop the motor dead. Inpractice, a closed-loop control sys-tem with tachometric feedback isusually necessary.

Designing In RuggednessThe output of a bridge circuit driv-

ing multiple-ampere inductive loadscan be a very ugly signal to coupleback into a hapless IC. AlthoughMOSFETs contain integral body di-odes which conduct during thedead-time, slow turn-on times andwiring inductances can result inspikes of several volts below ground

Figure 6. Full-bridge motor control with the LT1162

components. As a result, more dead-time than is normally needed must beused, increasing the conduction timein the lossy MOSFET body diodes.

The LT1160 and LT1162 (like theLT1158) use an adaptive system thatmaintains dead-time independent ofthe type, the size and even the num-ber of MOSFETs being driven. It doesthis by monitoring the MOSFET gateturn-off to see that it has fully dis-charged before allowing the oppositeMOSFET to turn on. In this way,cross-conduction is completely elimi-nated as a design constraint.

The high efficiency 10A stepdown(buck) switching regulator shown inFigure 4 illustrates how different sizedMOSFETs can be used without hav-ing to worry about shoot-throughcurrents. Since 40V is being droppeddown to 5V, the duty cycle for theswitch (top MOSFET) is only 5/40 or12.5%. This means that the bottomMOSFET will dominate the RDS(ON)efficiency losses, because it is turnedon seven times as long as the top.Therefore a smaller MOSFET is usedon the top, and the bottom MOSFETis doubled up, all without having toworry about dead-time. The noncriti-cal Schottky diode across the bottomMOSFETs reduces reverse-recovery

OUTPUT CURRENT (A)

75

80

85

90

95

EFFI

CIEN

CY (%

)

10

1160_5.eps

0 2 4 86

1160_6.eps

UV OUT A

BOOST

V+

60V MAX

12V

T GATE DR A

T GATE FB A

T SOURCE A

B GATE DR A

B GATE FB A

GND

IN TOP A

IN BOT A

LT1162

UV OUT B

BOOST

T GATE DR B

T GATE FB B

T SOURCE B

B GATE DR B

B GATE FB B

GND

IN TOP B

IN BOT B

+

+

10µF

IRFZ44

IRFZ44

1N4148

1000µF

PWM 0-100kHz

25kΩ

CBOOTSTRAP 0.1µF

+IRFZ44

IRFZ44

1N4148

1000µF

PWM 0-100kHz

25kΩ

CBOOTSTRAP 0.1µF

V+ V+ V+

1 7 14 26

4

24

23

22

21

19

6

5

2

3

10

18

17

16

15

13

12

11

8

9

Figure 5. Operation efficiency for Figure 4’scircuit. Current limit is set at 15A

or above supply. The LT1160 andLT1162 source pins have been de-signed to take repetitive outputtransients 5V outside the supply railswith no damage to the part. Thissaves the expense of adding high cur-rent Schottky diode clamps.

ConclusionThe LT1160 and LT1162 N-

channel power MOSFET drivers an-ticipate all of the major pitfallsassociated with the design of highefficiency bridge circuits. The de-signed-in ruggedness and numerousprotection features make these driv-ers the best solution for 10V to 60Vmedium-to-high current synchronousswitching applications.

DESIGN FEATURES


LTC®1430 Provides Efficient GTL Supplyby Craig Varga

1.5V output. The initial current was1A and the final current 5A. Thetop trace is the output voltage(100mV/div) and the bottom trace isthe regulator’s inductor current(5A/div). In Figure 4, the load resistoris switched from ground to a 3Vsupply, resulting in bidirectional load-ing. Current is switched from 5A to−4A. Again, the top trace is the outputof the 1.5V supply. The relatively longrecovery period is actually a responseto the perturbation that appears onthe input voltage as a result of thelarge load change. Voltage mode con-trol suffers from inherently poor linerejection. In this case, the lab supplyused for the tests exhibits ratherpoor dynamics. If a better input sup-ply is used, the overall settling timewill be much faster. The LTC1430also has a very accurate referenceand exhibits good static load regula-tion, as can be seen from the verysmall offset between the settled lightload and heavy load conditions on theoutput-voltage trace.

A recent trend in computer busarchitecture is the move toward GTL(Gunning Transition Logic) for rout-ing high speed signals between theCPU and chipset logic. Buses with

Figure 1. Schematic diagram of GTL supply

+

+

+ +

+

L1 2.7µH

VCC

IMAX

SHDN

COMP

SS

SGND

FSET

PVCC21

16

3

5

7

6

13

2

14

12

8

10

9

4

11

15

G1

G2

PGND

–SENSE

+SENSE

FB

IFB

PVCC1

U1 LTC1430

R6 10k

R2, 3Ω

R1 1k

R3 3Ω

R4 51

R5 16.5k 1%

R9 88.7k 1%

R7 15k

R10 3.3k

R8 130k

C7 0.1µF

C8 22µF 35V

C12 330 6.3V OSCON

C13 330

6.3V OSCON

C4 330 6.3V OSCON

1.5V OUT

C5 330 6.3V OSCON

S/D

5V

C6 0.01µF

C16 0.1µF

C2 680pF

C10 1µF

C9 1µF, 25V

C11 0.068µF

C3 220pF

C1 0.1µF

Q1 Si4410

Q2 Si4410

D2

D1 MBRS330T3

di1430_1.eps

330 6.3V TANT

330 6.3V TANT

ADDITIONAL DECOUPLING AT THE LOAD

++

MBR0530T1

DESIGN IDEAS...

LTC1430 Provides EfficientGTL Supply ................... 17Craig Varga

Battery Charger SinksConstant Current .......... 18Mitchell Lee

Low Noise Charge PumpBiases GaAsFET Gates .. 19Mitchell Lee

C-LoadTM Op AmpsConquer Instabilities ..... 20Frank Cox and Kevin R. Hoskins

Lowpass Filters with Rail-to-Rail Input and Output ... 23Philip Karantzalis and Jimmylee Lawson

speeds in excess of 66MHz are usingthis technology. Several different GTLsupply voltage standards seem to beevolving, with voltages from 1.2V to1.5V being discussed most often. Insome implementations of the logicfamily, the GTL supply must bothsource and sink currents of 5A–10A.This causes a great deal of difficultyfor a typical current mode controlpower supply design that is incapableof controlling currents less than zero.The LTC1430 is a voltage mode con-troller, and as such, does not exhibitany problems with negative load cur-rents.

Most GTL incarnations, however,appear not to require current-sinkingsupplies, and the LTC1430 is wellsuited for these applications as well.The circuit shown in Figure 1switches at 300kHz, regulating a 5Vinput down to a 1.5V output whilemaintaining good efficiency. (See Fig-ure 2. Figures 2–4 are on the bottomof page 18.) Figure 3 shows the effectsof a fast 4A load transient on the

DESIGN IDEAS


Battery Charger SinksConstant Current by Mitchell Lee

the current available from the source.The circuit shown in Figure 1 doesjust that, while charging 2 NiCd cells.Input current is sensed by a PNPtransistor and applied as feedback toan LTC1174 stepdown converter. TheBurst Mode™ converter draws justenough power to hold the input cur-rent at 50mA, matching the abilitiesof the wall adapter.

Output current into two cells isnominally 180mA, but climbs to over300mA when the batteries are com-pletely discharged. Efficiency is about78% at 3V output. If the batteries areremoved from the charging circuit,the output voltage could climb tolevels destructive to the load. To

DC-to-DC converters make veryefficient constant current sources forcharging NiCd batteries. Unfortu-nately, the input of a switcher exhibitsa negative impedance, and this cancause problems in systems where thesource is power or current limited.Various schemes are used to precludeinput foldback. Undervoltage lockoutworks well with power -limitedsources, keeping the converter OFFuntil the input voltage has risen tothe point where it can develop ad-equate power.

Undervoltage lockout does not workwith current-limited sources. A bet-ter approach is to control theconverter’s input current to match

prevent this, a second feedbackpath is applied to Pin 1 via a diode,limiting the output voltage to approxi-mately 3.7V.

Another condition that can spelltrouble for simple circuits such asthis one is the loss of input powerwhile the batteries are still connectedin circuit. An extra Schottky diodeplaced in series with the switch (Pin 5)blocks reverse current flow into theLTC1174 and prevents damage. Theoutput can also be short circuitedwithout damage to the device. Shut-down (Pin 8) allows logic control of thecharger, so the charging current canbe interrupted without disconnectingthe source supply.

Figure 2. Efficiency plot of the GTL supply

LOAD CURRENT (A)

55

65

60

70

75

80

85

90

95

EFFI

CIEN

CY (%

)

7

di1430_2.eps

0 1 2 3 654

LTC1430, continued from page 17

di1430_3.eps

INDUCTORCURRENT

5A/DIV

OUTPUTVOLTAGE

100mV/DIV

OUTPUTVOLTAGE

100mV/DIV

INDUCTORCURRENT

5A/DIV

di1430_4.eps

Figure 4. Oscillograph showing effects ofbidirectional loading on the GTL supply

Figure 3. Oscillograph showing effects of4A load step on the 1.5V output

Figure 1. Schematic diagram of constant-current battery charger

+

+

+ +

di1174_1.eps

50mA INPUT

2× 100µF

16V OS-CON

100µF 16V

OS-CON

47µF 6.3V OS-CON

2 NiCd CELLS

11Ω 100Ω

5

62N3906

1

1N5818

1N5818

LOAD

100pF *COILTRONICS (407) 241-7876

7 2 3 4

8

1N4148

CTX100-1*SHDN SW

FBIPGM GND

VIN

LTC1174HV-3.3


DESIGN IDEAS


Low Noise Charge Pump BiasesGaAsFET Gates by Mitchell Lee

ure more typical of generic chargepumps. A plot of spot noise is shownin Figure 2. Above 20kHz, broadbandoutput noise is gradually eliminatedby the 10µF output capacitor.Throughout the audio range the noiseis 600–700nV/√Hz.

Figure 3 shows the spectral re-sponse of the LTC1550 over a range of100kHz to 10MHz. Switching noise at900kHz and related harmonics areless than 10µV. The ferrite bead usedduring these tests attenuates har-monics above 5MHz. Depending onthe application, a series resistor of 1Ωto 10Ω may be just as effective atattenuating noise.

The LTC1550 is part of a largerfamily of devices including multipleversions and the LTC1551. The onlydifference between the LTC1550 andLTC1551 is shutdown polarity (theLTC1550 has inverting shutdown, andthe LTC1551 has noninverting shut-down). Packages with higher pincounts include features such as ad-justable output and a “power good”pin for controlling power to theGaAsFET drain. All devices exhibitthe same characteristic low outputnoise. See the LTC1550/LTC1551data sheet for a complete descriptionof all available options.

Figure 2. Spot noise plot of the circuit shown in Figure 1

tra, rendering the usual charge pumpunsuitable for this application.

A new generation of charge pumpswith on-chip post regulators has beendeveloped to satisfy this application.Although the charge pump is retainedas the most economical and compactmethod of generating a negative volt-age from a positive input, a linearpost regulator is added to “purify” theotherwise noisy charge-pump output.The result is a negative bias supply soquiet you’ll need to build a low noisepreamp in order to observe the noiseon a spectrum analyzer.

Figure 1 shows the LTC1550 lownoise, regulated, switched-capacitorvoltage inverter. Total noise over a40MHz bandwidth is approximately200µVRMS with a 1mA load. Comparethis with 10–100mVRMS noise, a fig-

Gallium arsenide MOSFETs arewidely used in wireless applicationsas RF output amplifiers, especially incellular telephones. Although theyprovide excellent power gain, outputpower and efficiency on low voltagebattery supplies, GaAsFETs are deple-tion-mode devices and require anegative gate voltage for proper bias.Since only a small current (less than10mA) is required, a charge pump isthe best solution for generating anegative gate-bias supply.

Charge pumps normally conjurevisions of noise, and rightly so. Theirvery operation relies on transferringenergy from input to output by alter-nately charging and discharging atransfer capacitor—a decidedly noisyprocess. Noise on the gate-bias linetranslates directly to the output spec-

di1550_1.eps

FERRITE BEAD*

C1 0.1µF

C2 0.1µF

CL 0.1µF

COUT 10µF

5V

4.7µF

1

2

3

4

8

7

6

5

*DALE ILB-1206-19

3900Ω

VOUT = –4.1V

+

LTC1550-4.1

SHDN

VCC

C1+

VOUT

SENSE

CPOUT

GND

C1–

+

Figure 1. Circuit diagram: LTC1550 low noise, regulated, switched-capacitor voltage inverter

FREQUENCY (Hz)

1n

10n

100n

1µ

10µ

100µ

NOIS

E (V

/√Hz

)

100k

di1550_2.eps

1k 10kFREQUENCY (Hz)

–40

0

–10

20

30

50

–20

–30

10

40

60

NOIS

E (d

BµV)

10M

di1550_3.eps

1k 10k 100k 1M

BW = 100Hz

Figure 3. Spectral response of the LTC1550, 100kHz to 10MHz

DESIGN IDEAS


C-Load™ Op AmpsConquer InstabilitiesIntroduction

LTC has taken advantage of pro-cess technology advances and circuitinnovations to create a series ofC-Load operational amplifiers. All ofthese amplifiers are tolerant of ca-pacitive loading, and a majority ofthem remain stable driving any ca-pacitive load. This series of amplifiershas a bandwidth that ranges from160kHz to 140MHz. These amplifiersare appropriate for a wide range ofapplications such as coaxial cabledrivers, video amplifiers, analog-to-digital converter (ADC) input buffer/amplifiers and digital-to-analog con-verter (DAC) output amplifiers.

This article looks at a few of themany applications for which theC-Load amplifiers are appropriate,including overcoming the large junc-tion capacitance of surge protectiondiodes on the outputs of videoamplifiers, driving the dynamicallychanging capacitive loads of highspeed ADC converters’ analog inputs,and driving an ADC’s reference withvarying voltages.

Table 1 lists LTC’s unconditionallystable voltage-feedback C-Loadamplifiers. Table 2 lists other voltage-feedback C-Load amplifiers that arestable with loads up to 10,000pF.

by Frank Cox andKevin R. Hoskins

current of many amplifiers causesslew limiting. The excellent outputdrive (70mA minimum at ±15Vsupplies) and the C-Load stabilityfeatures of the LT1363 make thevideo performance of this circuitalmost indistinguishable from thesame driver without the added ESDprotection. Of course, the 75Ω seriesmatching resistor on the outputhelps to isolate the capacitive load.For the best high speed performance,we recommend that the coax cablebe matched in this way. However,there are cases where the maximumoutput swing is required and thisresistor cannot be used (of course, forquality video a termination resistoris always used on the end of thecable), so tests were run with com-posite video to quantify the effect ofthis component. Table 3 gives differ-ential gain and phase measurementsfor the amplifier circuit with andwithout the output series resistor.

Video Performancewith ESD Protection

The circuit shown in Figure 1 is asimple gain-of-2 line driver that iscapable of withstanding high levels ofelectrostatic discharge (ESD) whileretaining its excellent video perfor-mance. The output of the LT1363amplifier is protected with transientvoltage suppressors. These suppres-sors are Zener diodes designed forcircuit protection and specified forhigh peak power dissipation, stand-off voltage and maximum surgecurrent. The P6KE15A diodes used inthis circuit have a peak power dissi-pation of 600W, a minimum stand-offvoltage of 12.8V and a maximum surgecurrent of 28A. This presents an ex-traordinary level of protection forthe output of the amplifier, but thelarge junction diodes present a prob-lem for some video circuits.

The series connection of the twoP6KE15A diodes has 500pF of totalcapacitance in the OFF state(the normal operating condition). Normally, large capacitive loads areto be avoided in video amplifiers. Notonly can capacitive loading cause os-cillation in amplifiers that are notdesigned for it, but the limited output

Figure 1. Line driver circuit diagram

dicl_1.eps

–

+

+

+

15V P6KE20

4.7µF TANT

0.1µF CER

15V

1000Ω

75Ω

4.7µF TANT

0.1µF CER P6KE20

LT1363

–15V

–15V

1N571275Ω

1N5712

510Ω

510Ω

P6KE15A ×2

OUT

IN

Table 1. Unity-gain stable C-Load amplifiersstable with all capacitive loads

GBW IS /AmpSingles Duals Quads (MHz) (mA)LT1012 — — 0.6 0.4

— LT1112 LT1114 0.65 0.32LT1097 — — 0.7 0.35

— LT1457 — 2 1.6

Table 2. Unity-gain stable C-Load amplifiersstable with CL ≤ 10,000pF

GBW IS /AmpSingles Duals Quads (MHz) (mA)

— LT1368 LT1369 0.16 0.375LT1200 LT1201 LT1202 11 1LT1220 — — 45 8LT1224 LT1208 LT1209 45 7LT1354 LT1355 LT1356 12 1LT1357 LT1358 LT1359 25 2LT1360 LT1361 LT1362 50 4LT1363 LT1364 LT1365 70 6

DESIGN IDEAS


of Figure 3. The LT1206 high currentCFA has no measurable performancedegradation for video signals with thesuppressor diodes on the output,although the restriction of the input-clamp network still holds. The 250mAoutput drive (over 0°C to 70°C tem-perature with ±5V to ±15V supplies)capability and the C-Load feature ofthe LT1206 make it a natural fordriving difficult loads.

Driving ADCsMost contemporary analog-to-digi-

tal converters (ADCs) incorporate asample-and-hold (S/H) using a topol-ogy exemplified by the circuit in Figure4. The hold capacitor’s (C1) size var-ies with the ADC’s resolution, but isgenerally in the range of 5pF–20pF,10pF–30pF or 10pF–50pF for 8-, 10-or 12-bit ADCs, respectively.

At the beginning of a conversioncycle, this circuit samples the appliedsignal’s voltage magnitude and storesit on its hold capacitor. Each time theswitch opens or closes, the amplifierdriving the S/H’s input faces a

Differential DifferentialPhase Gain

LT1363 with 0.11° 0.03%output clamp and75Ω output RLT1363 with 0.11° 0.3%output clamp and no RLT1206 with 0.1° 0.1%output clamp and75Ω output RLT1206 with 0.11° 0.1%output clamp and no R

dicl_2.eps

15V

20k

–15V

1N581975Ω

1N5819

NONINVERTING INPUT OF LT1363

>> –AV

C1

AIN

GND

dicl_4.eps

TO SAR

FROM INTERNAL DAC

dynamically changing capacitive load.This condition generates currentspikes on the input signal. Thesespikes, along with the capacitive load,are a very challenging load that canpotentially produce instabilities in anamplifier driving the ADC’s input.These instabilities make it difficultfor an amplifier to settle quickly. If theoutput of an amplifier has not settledto a value that falls within the errorband of the ADC, conversion errorscan result. That is, unless the ampli-fier is designed to gracefully andaccurately drive capacitive loads, asare Linear Technology’s C-Load lineof monolithic amplifiers.

As can be seen in Figure 5a, anamplifier whose design is not opti-mized for handling large capacitiveloads has some trouble driving thehold capacitor of the LTC1410’s S/H.Although the LT1006 has other very

Except for a slight rise in the LT1363’sdifferential phase, there is no mea-surable degradation. The output andpower supply pins of the driver circuitwithstand repeated air discharges of17kV (the limit of the test equipment)without failure. The input is sensitiveto the loading effects of the diode andresistor network because the junc-tion capacitance of the clamp diodesworks against the impedance of thecable-termination resistors. For thisreason, low capacitance clamp di-odes were used. The input of thedriver circuit still stood off 10kV.If some reduction in the frequencyresponse (approximately 3dB at5MHz) can be tolerated, the networkshown in Figure 2 can be added to theinput to increase the standoff voltageto 17kV.

For even more demanding applica-tions, we recommend the driver circuit

Figure 3. 250mA output line driver

Table 3. Differential gain and phasemeasurement summary

Figure 2. Circuit with modified clamp forhigher voltage

Figure 4. Typical ADC input stage showinginput capacitors

CONVST1V/DIV

INPUT200mV/DIV

dicl_5a.eps200ns/DIV

dicl_5b.eps200ns/DIV

INPUT50mV/DIV

CONVST1V/DIV

Figure 5a. Input signal applied to anLTC1410 driven by an LT1006

Figure 5b. Input signal applied to anLTC1410 driven by an LT1360

dicl_3.eps

–

+

+

+

15V P6KE20

4.7µF TANT

0.1µF CER

0.01µF

V+

1k

75Ω

4.7µF TANT

0.1µF CER P6KE20

LT1206

V–

–15V

1N571275Ω

1N5712

1200Ω

1200Ω P6KE15A ×2

DESIGN IDEAS


desirable characteristics, such lowVOS, high slew rate and low powerdissipation, it has difficulty accuratelyresponding to dynamically changingcapacitive loads and the currentglitches and transients they produce,

as indicated by the instabilities thatappear in the lower oscilloscope tracein Figure 5a.

By contrast, Figure 5b shows theLTC1360 C-Load op amp driving thesame LTC1410 input. The photo continued on page 25

+

+ +

–

+

–5V

–5V

5V

0.1µF

5V

dicl_6.eps

0.1µF

INPUT

0.1µF10µF

+AIN 12-BIT DATA

–AIN

AVDD

DVDD

VSS

AGND

REFCOMP

VREF

10µF

OGND CONTROL LINES

LTC1410

DGND

0.1µF

0.1µF 10µF

LT1006 OR LT1360

CONVST

Figure 6. Test circuit used to measure LTC1410 input signal waveform

Figure 7. The 12-bit LTC8043 DAC and the C-Load LT1220 op amp create variable reference voltages, enhancing the 12-bit LTC1410 ADC’sinput range flexibility

AVDD

DVDD

VSS

BUSY

CS

CONVST

RD

SHDN

NAP/SLP

OGND

D0

D1

D2

D3

+AIN

–AIN

VREF

COMP

AGND

D11 (MSB)

D10

D09

D08

D07

D06

D05

D04

DGND

1

2

3

4

5

6

7

8

9

10

11

12

13

14

28

27

26

25

24

23

22

21

20

19

18

17

16

15

U4 LTC1410

dicl_7.eps

5V

–5V

–5V

C8 0.1µF

BUSY

CHIP SELECT

CONV START

READ

SHUTDOWN

SHUTDOWN MODE

DATA BIT 0

DATA BIT 1

DATA BIT 2

DATA BIT 3

DATA BIT 4DATA BIT 5DATA BIT 6DATA BIT 7DATA BIT 8DATA BIT 9DATA BIT 10DATA BIT 11

SERIAL CLOCKSERIAL DATADAC LOAD

R1 5.1k

C5 0.1µF

±2.5V ANALOG INPUT RANGE

C4 0.1µF

5V

C2 0.1µF

C3 0.1µF

C1 0.1µF

C9 0.1µF

–

+

+

VDD

CLK

SRI

LD

VREF

FB

IOUT

GND

1

2

3

4

8

7

6

5

U2 LTC8043

U3 LT1220

U1 LT1004-2.5

C7 0.1µF

C6 10µF

2

3

6

5V

–5V

7

4

shows that the LTC1360 is an idealsolution for driving the ADC’s inputcapacitor quickly and cleanly withexcellent stability. Its wide 50MHzgain bandwidth and 800V/µs slewrate very adequately complement theLTC1410’s 20MHz full-power band-width. The LTC1360 is specified for±5V operation.

Figure 6 shows the circuit used totest the performance of op amps driv-ing the LTC1410’s input and measurethe ADC’s input waveforms.

Adjustable ADCReference Voltage

An ADC’s reference voltage shouldalways be set to a value that maxi-mizes the number of codes generatedduring input signal conversion. If thereference voltage is set for input sig-nals with the widest full-scale range,this might be too large for input sig-nals with a much smaller range. Forexample, an ADC’s full-scale input

DESIGN IDEAS


Lowpass Filters with Rail-to-Rail Inputand Output

The proliferation of systems oper-ating exclusively with single 5Vsupplies has created a need for ana-log signal conditioning circuits withrail-to-rail input and output. Opamps, A/D converters, and D/A con-verters with rail-to-rail inputs andoutputs are readily available, but nothigh-order lowpass filters. High-order lowpass filters are used toprovide antialiasing protection forA/D converters or to bandlimit theoutputs of D/A converters.

Rail-to-Rail, DC-Accurate,Butterworth and Bessel 7th-Order Lowpass Filters.

Ther circuit of Figure 1 uses adual rail-to-rail op amp and a DC-accurate switched-capacitor filter toprovide a 7th-order lowpass filter.The rail-to-rail op amps accommo-date the nonrail-to-rail outputoperation of the switched capacitorfilter. Overall rail-to-rail operation isachieved by attenuating and levelshifting the input signal and by levelshifting and amplifying the output ofthe switched-capacitor filter.

difilt_1.eps

+

R1 10k

R3 10k

R4 49.9k

0.1µF

R2 20k

C3* 0.039µF

IN

5V

5V1

2

3

4

8

7

6

5

LTC1063 OR

LTC1065

ROS* 200k

1µF TANT

4.99k

4.53k5V

0.1µF

COS 510pF

R8 20k

R9 10k

OUTC1

0.1µF

R10 20k

R7 20k

5V

C2 0.1µF

R6* 11.3k

R5* 5.62k

*THE VALUES FOR THESE COMPONENTS DETERMINE THE FILTERS –3dB FREQUENCY (SEE TABLE 1). NOTE: ALL RESISTORS ARE 1% METAL FILM AND ALL CAPACITORS ARE 5%.

–

+1/2

LT1366–

+1/2

LT1366

The circuit of Figure 1 operateswith a single 5V supply and attenu-ates the input signal by a factor oftwo. A 0-to-5VP–P input signal ap-pears as 1.25–3.75V at the input ofthe filter. The attenuation factor oftwo is chosen as a convenient num-ber, and does not fully exploit the3.5V output-voltage range of theswitched-capacitor filter. The penaltyis a mere 2dB loss of signal-to-noiseratio, yet the circuit is still usable for12- to 13-bit applications.

The input op amp of Figure 1provides buffering. A single poleantialiasing filter is added in front ofthe switched capacitor filter throughcapacitor C3; its cutoff frequency is3.3 times the cutoff frequency of theswitched-capacitor filter.

The output op amp of Figure 1provides an additional 2-pole, con-tinuous-time filter and restores therail-to-rail output swing. Table 1shows the component values of Fig-ure 1 for three different cutofffrequencies. The passive componentswere carefully selected to work forboth filter types, Butterworth andBessel.

Other cutoff frequencies can beobtained by scaling the componentvalues of Table 1. The 7th-order am-plitude response of the circuit is shownin Figure 2. Figure 2 was traced witha 1.7VRMS input (4.76VP–P), and themeasurement bandwidth was 22kHz.The filter cutoff frequency was set to1kHz. Attenuation as high as 80dB isobtained with the linear-phaseLTC1065 filter. Figure 3 shows thedynamic range (S/N + THD) of thefilter of Figure 1.

by Philip Karantzalis and Jimmylee Lawson

FREQUENCY (Hz)

–90

–40

–50

–60

–70

–80

10

0

–10

–20

–30

GAIN

(dB)

10k

difilt_2.eps

100 1k

VS = 5V

LTC1063

LTC1065

Figure 1. 100Hz 7th-Order Butterworth or Bessel lowpass filter with rail-to-rail input and output

Figure 2. Amplitude response of Figure 1’scircuit for both LTC1063 and LTC1065

DESIGN IDEAS


–3dB Freq C3 ROS COS R5 R610Hz 0.39µF 2M 510pF 56.2k 113k

100Hz 0.039µF 200k 510pF 5.62k 11.3k1kHz 0.0039µF 19.1k 510pF 562Ω 1.13k

Table 1. Component values for Figure 1's circuit for threedifferent cutoff frequencies

20Hz Rail-to-RailLinear-Phase LowpassFilter with 60Hz Notch.

The circuit of Figure 5 is a 20HzBessel filter with an additional 60Hznotch. This type of circuit is useful inapplications where excessive 60Hznoise is present and, on top ofbandlimiting, additional attenuationof 60Hz is required. The same schemeof rail-to-rail op amps is used forinput and output buffering, exceptthe output of the first op amp is fedforward to the input of the Sallen-Keylowpass filter. The output of the

LTC1065 is connected to the nonin-verting input of the Sallen-Key filterthrough R11. When the output of thefirst op amp is 180 degrees out ofphase from the output of the LTC1065,a notch depth of 50dB or greater canbe achieved. In order to achieve thisnotch depth for 60Hz, the resistorsR5, R6, R11 and ROS must be 0.1%and the COS capacitor must be 1%tolerance. Figure 6 shows the gainresponse of the circuit in Figure 5.

The 76dB maximum dynamic rangeoccurs around 4.75VP–P and thisshould be the full scale of the system.Figure 4 shows the transient responseof both Butterworth and Bessel 7th-order filters.

The measured DC gain error ofFigure 1’s circuit is less than 0.5%.This depends on the matching of re-sistors R1–R4 and R7–R10. Themeasured DC gain nonlinearity ofthis circuit is ≤0.1% (–80dB).

difilt_5.eps

+

1

2

3

4

8

7

6

5

R1 10k

R3 10k

R4 49.9k

0.1µF

R2 20k

C3 0.039µF

IN

5V

5V

LTC1065

ROS 200k 0.1%

1µF TANT

4.99k

4.53k5V

0.1µF

COS 510pF 1%

R8 20k

R9 10k

OUT

C1 0.1µF

R10 20k

R7 20k

5V

C2 0.1µF

R6 150k 0.1%

R5 49.9k 0.1%

NOTE: ALL RESISTORS ARE 1% METAL FILM AND ALL CAPACITORS ARE 5% UNLESS SPECIFIED OTHERWISE.

–

+1/2

LT1366–

+1/2

LT1366

R11 113k 0.1%

Figure 5. Schematic diagram of 20Hz Bessel lowpass filter with 60Hz notch

Figure 3. Dynamic range (THD + Noise) versusinput voltage plot for the circuit of Figure 1

Figure 6. Amplitude response of the circuitdetailed in Figure 5

Figure 4. Oscillograph of square waveresponse; top trace is the LTC1065 Besselfilter, bottom trace is the LTC1063 Butter-worth filter

BUTTERWORTH

BESSEL

INPUT VOLTAGE (VP-P)

RLOAD = 100k

f–3db = 100Hz fin = 50Hz

0.5–90

THD

+ NO

ISE

(dB)

–40

–45

5.55.0

difilt_3.eps

4.54.03.53.02.52.01.0 1.5

–80

–85

–70

–75

–60

–65

–50

–55

RLOAD = 5kVS = 5V

FREQUENCY (Hz)

–90

–40

–50

–60

–70

–80

10

0

–10

–20

–30

GAIN

(dB)

1k

difilt_6.eps

10 100

DESIGN IDEAS


C-load, continued from page 22

span is set to 5V. Input signals withmagnitudes that cover this span willuse all of the ADC’s codes. However,an input signal that only spans 2Vwill use only 2/5 of the codes, result-ing in a loss of resolution.When the reference voltage is re-duced to 2V, the 2V signal will nowuse all of the ADC’s codes. But withthe 2V reference, an input signalhaving a 4V span is likely to exceedthe input span, resulting in clippedsignals and missed information. Incases such as this, an adjustablereference is needed to adjust theADC’s full-scale gain.

input bypass capacitor C5. TheLTC1410 can easily handle a refer-ence voltage as low as 1V (±2VP-Pinput range) with only a 0.5 LSBchange in linearity error.

ConclusionUsing advanced process technol-

ogy and innovative circuit design,LTC has created a series of C-Loadoperational amplifiers that not onlytolerate capacitive loading, but re-main stable driving any capacitiveload. C-Load amplifiers meet thechallenging and difficult capacitiveloads by remaining stable andsettling quickly.

The circuit in Figure 7 uses theLTC8043 12-bit CMOS 4-quadrantDAC driving the LT1220 C-Loadamplifier to generate variable ADCreference voltages that allow the userto adjust the ADC’s gain. An LT1004-2.5 micropower 2.5V reference is usedas the LTC8043’s reference source,setting its output span to 0V–2.5V.The LT1220’s output is applied to theLTC1410’s VREF input. The LT1220C-Load amplifier ensures that, whenchanging the LTC1410’s referencevoltage, the amplifier does not oscil-late even while driving the reference

PFC, continued from page 11 (Design Features)

probe to see the PFC in action. Scopeprobes don’t read duty factor verywell, and probing on the FET drainreveals a waveform that doesn’t looklike anything you have ever seen be-fore, and may awaken you in themiddle of the night. It’s not that the

waveform isn’t pretty, it’s that thewaveform is very busy, and it is hardto see anything like a sine wave in it.The input current waveform is fareasier to enjoy, it’s just harder to getto without a good current probe.

ConclusionPFC is fun. It offers technical chal-

lenges not because it is complicatedbut because there are so many littleweirdnesses, all happening togetherin concert to provide a beautiful func-tion. One of the biggest challenges isto find a place to hook your scope

CONTINUATIONS


Included with this issue is acomplete index of the circuitsfeatured in Linear TechnologyMagazine, issues I:1 through V:4.This index evolved as a result of arequest from one of our FieldApplications Engineers, who wasunable to locate a circuit hebelieved had appeared in LinearTechnology Magazine. We set out

to create an index in which thereader could “browse” for a circuitthat fit his or her design needs orsearch for a specific LinearTechnology part; we believe thatthe table design we used herethoroughly fulfills this mission. Thereader can page to the desiredcircuit category, scan the columnsfor the circuit type and/or part

number, and easily find theappropriate article, issue, pagenumber, and figure number(s).

Over 250 circuits from the firstfive years of Linear Technologymagazine are included in thisindex. Hereafter, we will update theindex yearly, so new circuits won’tget lost. Pleasant Reading!

Index of Circuits in Linear Technology, Issues I:1 Through V:4


New Device CameosLTC1409 and LTC1415:High Speed, Low Power,12-bit A/D Converters

LTC has two new 12-bit A/D con-verters that set a new standard forhigh performance at low power. TheLTC1415 is a 12-bit, 1.2Msps A/Dconverter that operates on a single5V supply and draws just 60mW ofpower. The LTC1409 is a 12-bit,800ksps A/D converter with out-standing dynamic performance(SINAD = 71dB at Nyquist) that oper-ates on a ±5V supply.

The LTC1409 and LTC1415 bothhave differential input sample-and-hold circuits that reduce unwantedhigh frequency common mode noise.The LTC1415 has a unipolar inputrange of 0V to 4.096V (1mV per LSB).The LTC1409 has a bipolar inputrange of ±2.5V. The input ranges ofboth devices are set with an on-chip2.5V precision voltage reference.

Both devices have parallel outputinterfaces that allow easy connectionto microprocessors, DSPs or FIFOs.Two power-shutdown modes allowflexibility in further reducing thepower during inactive periods. Bothdevices come in 28-pin SO packages.

The LTC1446/LTC1447:World’s Only Dual12-Bit, Rail-to-Rail DACsin SO-8 Packages

The LTC1446 and LTC1447 arecomplete dual, 12-bit micropowerDACs with rail-to-rail buffer amplifi-ers and an onboard reference. Bothparts have a maximum DNL error of0.5LSB. They have an easy-to-usecascadable serial interface that is SPIcompatible. Applications for theseparts include industrial processcontrol, digital calibration, automatictest equipment, cellular telephonesand portable battery-poweredapplications where low supplycurrent is essential.

The LTC1446 operates on a singlesupply ranging from 2.7V to 5.5V,dissipating a mere 1.35mW from a 3V

supply. It has a wide output swing of0V to 2.5V. The LTC1447 operates ona single supply of 4.5V to 5.5V, dissi-pating 3.5mW. It has an output swingof 0V to 4.095V. Both parts have anonboard reference that is connectedinternally to each DAC. A DOUT pinallows several parts to be daisy-chained, and a built-in power-on resetclears the output to zero-scale atpower-up.

Both the LTC1446 and theLTC1447 come in an SO-8 package ora small 8-pin PDIP, allowing efficientuse of circuit board space. These dualDACs offer excellent DNL, low powerand a convenient serial interface, allin a tiny SO-8 package.

LT1464: First MicropowerJFET Op Amp for DrivingC-Loads

The LT1464 op amp combines fea-tures that would normally be uniqueto three different types of op amps.The LT1464 is the first micropowerJFET op amp in our C-Load family ofop amps. The output can drive ca-pacitive loads up to 10nF with anyload resistance. At the input side ofthe op amp, input bias currents areless than 1pA for demanding track-and-hold circuits and active filtersthat use high-value resistors. Inputcommon mode range includes thepositive supply. Power supply cur-rent is typically 250µ A, allowingmicropowered battery applications.For single supply requirements, a fullset of specifications are included for5V operation. The slew rate is 1V/µsand the gain bandwidth is 1MHz.

The LTC1458 and LTC1459:Quad 12-Bit, Rail-to-RailMicropower DACs

The LTC1458 and the LTC1459are complete quad 12-bit micropowerDACs with rail-to-rail buffer amplifi-ers and an onboard reference. Both of

these parts have a maximum DNLerror of 0.5LSB, and an easy-to-use,SPI compatible, cascadable serial in-terface. They also offer severalconvenient features that make theseparts flexible and easy to use. Indus-trial process control, automatic testequipment and digital calibration arejust a few of the applications forthese parts.

The LTC1458 operates on a singlesupply ranging from 2.7V to 5.5V anddissipates a mere 2.5mW from a 3Vsupply. It has a 1.22V onboard refer-ence. The LTC1459 operates on asingle supply of 4.5V to 5.5V, dissi-pating 6mW. Its onboard reference is2.048V. Both of these parts have aseparate reference input pin for eachDAC, which can be driven by an ex-ternal reference or connected to theonboard reference output. The usercan also offset the zero-scale voltagefor each DAC by means of a REFLOpin. Each of the rail-to-rail bufferamplifiers will swing to within a fewmillivolts of either supply rail and canbe connected in a gain-of-one or again-of-two configuration, giving anoutput full-scale of 1×REFIN or2×REFIN respectively. There is a built-in power on reset and a CLR pin thatresets the output to zero-scale.

Both the LTC1458 and theLTC1459 come in small 28-pin SSOPor 28-pin SO packages. These quadDACs offer the user flexibility with ahost of useful features, excellent DNLand low power dissipation.

LT1537 5V PoweredRS232 Transceiver

The LT1537 is a new three-driver/five-receiver RS232 interface trans-ceiver designed to serve as a serial I/Oport for AT-compatible computers.The LT1537 is pin compatible withthe LT1137A. Its integral charge-pump power generator uses lowcost 0.1µF and 0.2µF capacitors to

NEW DEVICE CAMEOS


NEW DEVICE CAMEOS

generate RS232 levels from standard5V power supplies. The device haslow power consumption: 8mA innormal operation and 1µA in shut-down mode.

The LT1537 is RS232/RS562 stan-dard compatible. Operation at datarates in excess of 250kBd for 1,000pFloads and 120kBd for 2,500pF loadsis guaranteed. Slew rate with 3kΩ,2500pF loads is a minimum of 4V/µs.When powered down or in shutdownmode, the driver outputs remainhigh impedance for line voltages to15V. Both driver outputs and receiverinputs can be forced to ± 25Vwithout damage.

Whereas most other RS232 trans-ceivers require external ESDprotection devices, the LT1537 isprotected against ±5kV ESD strikesto the RS232 inputs and outputs.This integrated ESD protection savesthe expense and space of externalprotection devices. This level of ESDprotection is adequate in most appli-cations. In applications where highESD level protection (IEC-801-2) isrequired, our LT1137A provides thatsolution. The LT1537 is available in28-lead SO and SSOP packages.

The LTC1449/1450 ParallelInput, 12-Bit Rail-to-RailVoltage Output MicropowerDACs

The LTC1449 and LTC1450 areLTC’s first pair of parallel-input, volt-age output DACs. These devices aredesigned as complete DACs with in-ternal references and buffered, truerail-to-rail voltage outputs. The ana-log specifications are equivalent tothose of the LTC1451 serial DAC,with excellent DNL performance ofless than ±0.5LSBs.

The buffered true rail-to-rail volt-age output can be configured for twodifferent full-scale settings. When theinternal reference is used and thegain-setting pin is tied to ground, thefull-scale range is 2.500V for theLTC1449 and 4.095V for the LTC1450.When the gain-setting pin is con-nected to VOUT, the full-scale ranges

The LTC1516 can be substitutedfor the MAX619 with improved per-formance. The part is available in8-pin PDIP and 8-pin SO packages.

LT1307 1-CellMicropower 675kHzPWM DC/DC Converter

The LT1307 is a micropower, fixedfrequency DC/DC converter that op-erates from an input voltage as low as1V. The first device to achieve truePWM performance from a 1-cell sup-ply, the LT1307 automatically shiftsto power-saving Burst Mode opera-tion at light loads. High efficiency ismaintained over a broad 100µA to100mA load range. The device in-cludes a low-battery detector with a200mV reference, and consumes lessthan 5µA in shutdown mode. No-loadquiescent current is 50µA, and theinternal NPN power switch handles a400mA current with a voltage drop ofjust 240mV.

Unlike competing devices, theLT1307 does not require large elec-trolytic capacitors. Its high frequency(675kHz) switching allows the use oftiny surface mount multilayer ceramic(MLC) capacitors and small surfacemount inductors. It works with just10µF of output capacitance and re-quires only 1µF of input bypassing.

The LT1307 is available in theSO-8 package.

change to 1.220V for the LTC1449and 2.047V for the LTC1450.

The LTC1449 and LTC1450 aredesigned to run off a single supplywhile consuming very little power.The LTC1449 can be powered from asingle supply as low as 2.7V; whenpowered from such a supply, it dissi-pates only 0.75mW. The LTC1450 ispowered from a single 5V supply anddissipates only 2mW of power. Thislow power dissipation makes theLTC1449 and LTC1450 ideal for bat-tery-powered applications.

Separate pins are provided for bothends of the voltage-scaling resistornetwork to allow for versatileconnection in multiplying andexternal-reference applications.Double-buffered parallel digital in-puts provide flexibility and easyinterface to popular DSPs and micro-processors.

The LTC1449 and LTC1450 areavailable in 24-lead PDIP and SSOPpackages.

The LTC1516: Micropower,Regulated, 5V Charge-PumpDC/DC Converter

The LTC1516 is a micropowercharge-pump DC/DC converter thatgenerates a regulated 5V output froma 2V to 5V supply, without inductors.Only four external capacitors (two0.22µF and two 10µF) are required todeliver a 5V ±4% output with up to50mA of output load current. Ex-tremely low supply current (12µAtypical with no load and less than 1µAin shutdown conditions) and low ex-ternal parts count make the LTC1516ideal for small, battery-powered ap-plications. The LTC1516 achievesefficiency greater than 75% with loadcurrents as low as 100µA, due to itsultralow quiescent current.

To further improve overall effi-ciency, the LTC1516 operates as eithera doubler or tripler, depending on VINand VOUT load conditions. The partalso has thermal shutdown protec-tion and can survive an indefiniteshort from VOUT to GND. In shutdownconditions, the load is disconnectedfrom VIN.

Burst Mode, Bits-to-Nits and C-Load aretrademarks of Linear Technology Corporation.

, LTC and LT are registered trademarksused only to identify products of Linear Tech-nology Corp. Other product names may betrademarks of the companies that manufac-ture the products.

Information furnished by LinearTechnology Corporation is believed to beaccurate and reliable. However, LinearTechnology makes no representation thatthe circuits described herein will not infringeon existing patent rights.

For further information on theabove or any of the other devicesmentioned in this issue of LinearTechnology, use the reader servicecard or call the LTC literature ser-vice number: 1-800-4-LINEAR. Askfor the pertinent data sheets andapplication notes.


AppleTalk is a registered trademark of Apple Computer, Inc.

© 1995 Linear Technology Corporation/ Printed in U.S.A./32K

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(408) 432-1900For Literature Only: 1-800-4-LINEAR

DESIGN TOOLSApplications on DiskNOISE DISKThis IBM-PC (or compatible) progam allows the user to calculate circuit noiseusing LTC op amps, determine the best LTC op amp for a low noise application,display the noise data for LTC op amps, calculate resistor noise, and calculatenoise using specs for any op amp. Available at no charge.

SPICE MACROMODEL DISKThis IBM-PC (or compatible) high density diskette contains the library of LTCop amp SPICE macromodels. The models can be used with any version ofSPICE for general analog circuit simulations. The diskette also containsworking circuit examples using the models, and a demonstration copy ofPSPICETM by MicroSim. Available at no charge.

Technical Books1990 Linear Databook • Volume I — This 1440 page collection of data sheetscovers op amps, voltage regulators, references, comparators, filters, PWMs,data conversion and interface products (bipolar and CMOS), in both commer-cial and military grades. The catalog features well over 300 devices. $10.00

1992 Linear Databook Supplement — This 1248 page supplement to the1990 Linear Databook is a collection of all products introduced since then.The catalog contains full data sheets for over 140 devices. The 1992 LinearDatabook Supplement is a companion to the 1990 Linear Databook, whichshould not be discarded. $10.00

1994 Linear Databook • Volume III — This 1826 page supplement to the1990 Linear Databook and 1992 Linear Databook Supplement is a collectionof all products introduced since 1992. A total of 152 product data sheets areincluded with updated selection guides. The 1994 Linear Databook Volume IIIis a supplement to the 1990 and 1992 Databooks, which should not bediscarded. $10.00

1995 Linear Databook • Volume IV — This 1152 page supplement to the1990, 1992 and 1994 Linear Databooks is a collection of all products introducedsince 1994. A total of 80 product data sheets are included with updatedselection guides. The 1995 Linear Databook Vol IV is a companion to the 1990,1992 and 1994 Linear Databooks, which should not be discarded. $10.00

Linear Applications Handbook • Volume I — 928 pages full of applicationideas covered in depth by 40 Application Notes and 33 Design Notes.This catalog covers a broad range of “real world” linear circuitry. In addition todetailed, systems-oriented circuits, this handbook contains broad tutorialcontent together with liberal use of schematics and scope photography.A special feature in this edition includes a 22 page section on SPICEmacromodels. $20.00

1993 Linear Applications Handbook • Volume II — Continues the streamof “real world” linear circuitry initiated by the 1990 Handbook. Similar in scopeto the 1990 edition, the new book covers Application Notes 41 through 54 andDesign Notes 33 through 69. Additionally, references and articles from non-LTC publications that we have found useful are also included. $20.00

Interface Product Handbook — This 424 page handbook features LTC’scomplete line of line driver and receiver products for RS232, RS485,RS423, RS422, V.35 and AppleTalk applications. Linear’s particularexpertise in this area involves low power consumption, high numbers ofdrivers and receivers in one package, mixed RS232 and RS485 devices, 10kVESD protection of RS232 devices and surface mount packages.

Available at no charge.

SwitcherCAD Handbook — This 144 page manual, including disk, guidesthe user through SwitcherCAD—a powerful PC software tool which aids in thedesign and optimization of switching regulators. The program can cut days offthe design cycle by selecting topologies, calculating operating points andspecifying component values and manufacturer's part numbers. $20.00

1995 Power Solutions Brochure, First Edition — This 64 page collectionof circuits contains real-life solutions for common power supply designproblems. There are over 45 circuits, including descriptions, graphs andperformance specifications. Topics covered include PCMCIA power manage-ment, microprocessor power supplies, portable equipment power supplies,micropower DC/DC, step-up and step-down switching regulators, off-lineswitching regulators, linear regulators and switched capacitor conversion.

Available at no charge.

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