master thesis: forward error control assisted detection

8/14/2019 Master Thesis: Forward Error Control Assisted Detection

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Forward-Error-Control-Assisted Detection

Petra Deutgen and Frode Rand ers

Lule University of TechnologyDiv. of Signal Processing

S-951 87 LuleSweden

Ericsson Radio Systems ABS-164 80 StockholmSweden


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Abstract

In the North American Digital Cellular Mobile Telephone System, D-AMPS, described by the stan dard TIA/ EIA IS-54, only a part of the d igi-

tal speech information is protected against transmission errors by anerror correcting code. This thesis considers a scheme th at u ses coded bitsto decrease the bit error rate of both uncoded bits and time-delayedcoded bits during detection. The detection scheme has been imple-men ted in a simulation environment for D-AMPS.

A p erformance gain of up to 3 dB of C/ I has been experienced for theunprotected speech data on a frequency selective fading channel for aresidual bit error rate of 1%. Correcting time-delayed coded bits is not assuccessful as correcting uncoded bits; coded bits showed only a verysmall decrease in bit error rate.


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Acknowledgements

The first part of the Masters thesis, written as a conference contribu-tion, could not have been written without the generous help of Karim

Jamal of Ericsson Radio Systems, d epar tment of Radio Access and Sig-nal Processing Research and Hkan Eriksson and Per dling both ofLule University of Technology, division of Signal Processing. Theopp ortunity to w rite such kind a pap er gave us great inspiration an d joy.We are most gra teful to them for their help.

We also wan t to th ank ou r ad viser, Johan von Pern er of Ericsson RadioSystems, department of Radio Base Station Development, he gave usvaluable sup por t, when w e were developing the reference simu lator. Asthe new detection scheme was developed, we experienced a largeamoun t of inspirat ion and h elp from our ad viser Karim Jamal. Thereforewe want to thank him once more, for his excellent gu idan ce.

It has been a stimu lating and pleasant tim e at Ericsson Radio Systems.We especially want to thank Omar Ryde and Erol Incenacar, both of Eric-sson Radio Systems, d epar tment of Radio Base Station Developm ent,wh o have contributed to this.

Petra Deutgen and Frode Randers


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Introduction

This Master s thesis p roject w as performed at Ericsson Radio Systemsin Kista, Sweden , as a p artial fulfillmen t of the degree of Master of sci-

ence in comp uter science and electrical engineering a t Lule Universityof Technology. This project is a result of a cooperation between twodep artm ents of Ericsson Radio Systems: the departmen t of Rad io Accessand Signal Processing Research (section of Rad io Access Research ERA/T/ UR) and the d epartmen t of Radio Base Station Development (sectionof Digital Modem Design AR/ RB).

The aim of this thesis has been to eva luate the substan ce of a re-detec-tion scheme herein denoted the multi-pass re-detection scheme, pro-posed by Paul Dent, of Ericsson-GE Mobile Communications Inc., RTP,NC. In order to measure the performance gain of the above-mentionedmethod , we h ave implemented a m ulti-pass re-detection scheme in theSysSim1 simu lation environment. We have been able to simu late the per-formance of both th e original receiver (a prod uct simu lator of TRX884, apart of a base station for the D-AMPS system) and the multi-passreceiver, run under actual conditions. Through these simulations, wehave been able to study various facets of the performance of the multi-pass receiver.

This thesis comprises four parts, of which part I is intended to be aself-contained conference contribution. The other parts, parts II-IV,describe the multi-pass re-detection scheme and the implementation ofit in greater d etail. Anoth er possible application of the method th an d oc-um ented in p art I is described in par t III.

Part I is written in the form of a conference paper that presents theresults of our w ork with re-detection ofuncodedbits by means of a mu lti-pass re-detection scheme.

Part II considers the possibility of using the multi-pass re-detectionscheme with d ifferent schemes of signalling. We concentrate on coherentQuad riphase Shift Keying (QPSK) mod ulation and Differential Quad -riph ase Shift Keying (DQPSK) that is used in D-AMPS.

Part III investigates the possibility of using error corrected bits todecrease the bit error rate for the time-delayed codedbits.

Part IV is a documentation of the parts of the software simu lator of themulti-pass re-detection receiver. Part IV will be more informative forreaders already familiar with the SysSim simulation environment. Anexplanation of the implementation of the multi-pass re-detectionreceiver is given with linkage to source code. The source code itself is,

1. SysSim is a simulation en vironm ent d eveloped at Ericsson Radio Systemsthat allows system simulation.


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Part 1


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AbstractIn the North American Digital CellularMobile Telephone System, D-AMPS, described by thestandard EIA/TIA IS-54, only a part of the digital speechinformation is protected against transmission errors by anerror correcting code. We evaluate a method, suggested in[1], that uses coded bits interleaved among uncoded bits to

decrease the bit error rate among the uncoded bits. In thismethod information about error-corrected data bits is fedback to aid a subsequent re-detection of the uncoded databits.

Simulations indicate that a decrease in the bit error rate

for unprotected speech data bits is achievable. A perfor-

mance gain of up to 3 dB of C/I has been experienced on a

frequency selective fading channel for a residual bit error

rate of 1%.

I. INTRODUCTION

In this paper, the detection of blocks of binary data in digi-

tal cellular telephone systems is investigated. Many contem-porary communication systems use unequal Forward Error

control Coding (FEC) [2], that is, some bits are protected by

more error correcting codes than other bits. This gives the

receivers access to more information about some bits than

about others. We investigate the potential of a scheme that

takes advantage of this.

We will focus on applying a re-detection method to the

North American D-AMPS system, specified by the standard

EIA/TIA IS-54 [3]. In the D-AMPS transmission format, the

bursts consist of both coded and uncoded bits, and the coded

bits are interspersed among the uncoded. This implies that

after having decoded the surrounding coded bits, an enhancedre-detection of the uncodedbits could be possible.

Assuming that the protected bits have been decoded cor-

rectly, these bits are used as known bits when re-detecting the

burst. Due to the memory in the modulation (differential

modulation), the re-detection can improve the Bit Error Rate

(BER) for the uncoded bits. Memory induced by the channel

(ISI) also allows for improvement, as does the use of decision

directed channel tracking.

Within a burst, the bits from the current frame are mixed

with bits from the next frame, due to interleaving. Therefore

it could also be possible to improve the BER for both the

uncoded andcoded bits belonging to this next frame [8].

The speech distortion known as warbling, sometimes

encountered in the VSELP speech decoder of D-AMPS, is

mainly due to bit errors among the uncoded bits. A method of

improving the BER for uncoded bits, with either a differential

detector or with a maximum likelihood sequence estimator

(MLSE), has recently been put forward in [1]. With a suitably

chosen interleaving pattern, a significant BER reduction is

claimed for the uncoded bits.Thus it is expected that such a

re-detection scheme would enhance the speech quality notice-

ably. In this paper we evaluate the scheme suggested in [1] for

the transmission format of D-AMPS.

Related work can be found in e.g. [4], [5] and [6] where

decoded data have been fed back to aid the detection process.

However, none of these papers exploits unequal error control

coding.

This paper proceeds as follows: Section II gives a shortdescription of the channel coder and the interleaver employed

in the reference transmission system. Section III presents the

one-pass receiver. Section IV outlines the re-detection of

uncoded bits with the aid of coded bits. Section V studies an

implementation of the re-detection scheme, denoted multi-

pass receiver, with the one-pass receiver, as a building block.

Depending on the interleaving, three types of multi-pass

receivers are studied, called the two-pass, three-pass and

composite receiver, respectively. The simulation setup is

described together with the simulated performances of the

multi-pass receivers in Sections VI and VII. An assessment of

the increase in complexity of implementing the multi-passreceivers is given in Section VIII, and finally, conclusions are

presented in Section IX.

II. CHANNEL CODING AND INTERLEAVING

The IS-54 standard discriminates among three levels of

FEC-induced protection, see Figure 1. For the bits with the

lowest significance, referred to as class 2 bits, no protection is

invoked. For the bits at the next level of significance, class 1

bits, a convolutional code has been used. Among the class 1

bits, the partition corresponding to the perceptually most sig-

Forward-Error-Control-Assisted Detection of Uncoded Bits in IS-54


Ericsson Radio Systems AB

S-164 80 StockholmSweden

Lule University of Technology

Div. of Signal ProcessingS-971 87 LuleSweden


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nificant bits, here denoted class 1a bits, have been addition-

ally protected by means of a cyclic redundancy check code

(CRC). All other class 1 bits are referred to as class 1b bits.

In the block-diagonal interleaving scheme of D-AMPS,

both coded and uncoded bits of a frame are written column-

wise into a bit matrix, able to contain one frame. The bits arethen transmitted row-wise in a burst B[k].

Figure 2 showsB[k] being assembled. The bits are selected

row-wise, alternatingly, from the matrix containing the cur-

rent frame, C[k], and the matrix containing the previous

frame, C[k-1]. Thus, the interleaving depth is one frame. In

terms ofC[k] andB[k], the interleaving can be described as

where and correspond to the set of even and odd

rows of the interleaving matrix, respectively.

III. THE ONE-PASS RECEIVER

Figure 3 shows the one-pass receiver scheme used in this

work. The detector is an adaptive MLSE with decision feed-

back channel tracking and with a subsequent differential

Figure 1. The channel coding and interleaving [2]

Figure 2. The interleaving

12 most

perceptuallysignificant bits

12

coded

class 1

bits 260260

7178

82

77

2-slotinterleaver

voice

cipher(optional)

Convolutional

encoder

(2,1,5)

Linear

block code

(19,12)

class 2 bits

class 1 bits

speech

coder

5

tail bits

10 bits

...

.

.. .

.

...

.

. . .

C[k-1] C[k]

c [k-1] c [k]1 2

c [k-1] c [k]1 2

c [k-1]1

0

2

1

3

B[k]

C k[ ] c1 k[ ] c2 k[ ]{ , }=

B k[ ] c1 k 1[ ] c2 k[ ]{ , }=

c1 k[ ] c2 k[ ]

decoder. The sequence estimation is done with a soft output

Viterbi detector. After the detection of the current burst, B[k],

de-interleaving is done, creating the frame C[k-1]. The class 1

bits are then decoded, while the class 2 bits are taken directly

from the de-interleaved frame, C[k-1].

IV. DETECTION OF UNKNOWN BITS AMONG KNOWN IN IS-54

To take advantage of the unequal protection of the trans-

mitted bits, the protected bits are, when successfully decoded,

used for restraining the trellis of the Viterbi detector (thenumber of valid transitions between the states in a trellis is

reduced) in a subsequent re-detection of the unprotected bits.

The restraining of the trellis, based upon knowledge of

coded bits, is determined by the method of modulation. With

differential phase shift keying, as used in D-AMPS, the infor-

mation lies in the differential phase shifts, so the restraining

will be in terms oftransition restraining.

Specifically, a set of data bits is associated with a specific

phase shift, see Figure 4, or equivalently a transition in the

trellis of the Viterbi algorithm. Any two parallel paths

through the trellis are equivalent as they give rise to the same

sequence of phase shifts. If the detector has knowledge of

some or all of the corresponding bits prior to detecting a dif-

ferential symbol, the trellis may be restrained, thus reducing

the number of valid transitions.

Figure 3. The one-pass receiver structure

Figure 4. A differential symbol representing two bits

F[k-1]

detector decoder

class 1 bits

class 2 bits

B[k] C[k-1]deinterleaver

c [k-1]2

c [k]2

Z-1

~

^

^B[k]^

^ ^

Q

I

(-1,-1)

(-1,1)

(1,1)(1,-1)

/2

3/2

0

~

~~

~


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Figure 5 shows different examples of transitions when a

symbol represents two bits; Both bits are unknown, the first

bit is known, both bits are known, the last bit is known.

V. THE MULTI-PASS RECEIVER

This section presents a receiver structure, denoted the

multi-pass receiver, that uses FEC-assisted re-detection.

Three types of multi-pass receivers will be studied, called

the two-pass, three-pass and composite receiver, respectively.

Due to the interleaving, the two-pass receiver uses some, but

not all, of the coded bits as known in the re-detection. In the

three-pass receiver all coded bits are considered as known

when re-detecting the uncoded bits, at the cost of adding a

time delay of one burst in the decoding process. The compos-

Figure 5. The restraining of the trellis

Figure 6. The multi-pass scheme

(x,1)(-1,y)(x,y) (1,1)

Q

I

Q

I

Q

I

Q

I

Prepare

feedback

information

Z-1

One-pass

receiver

B[k]~

B[k-1]~

Two-pass

receiver

Three-pass

receiver

F[k-1]^

F[k-1]^

F[k-1]^

F[k-2]^

c [k-1]^1

c [k-1]^2

Combiner

ite receiver, which is a combination of the two- and three-pass

receivers, avoids the time delay, but is able to use more pro-

tected bits than the two-pass receiver, see Figure 6.

A. The two-pass receiver

To extract information about the protected bits, a primary

detection, de-interleaving and decoding are done by the one-pass receiver, resulting in a complete set of class 1 bits as

contained in the frame . Given that the CRC check

for the class 1a bits in is affirmative, all of the coded

class 1 bits of , resulting from re-encoding

, are considered known1. Applying the two-pass re-

detection scheme, we initiate a re-detection of the class 2 bits

in the received burst by means of the two-pass receiver,

similar to the one-pass receiver. Due to interleaving, only the

fraction of the coded class 1 bits of found in

are known. Consequently, the re-detection of is

only based on these bits. The re-detection is done by restrain-

ing the trellis at the location of these known bits in the ViterbiAlgorithm (VA) of the two-pass detector. The two-pass re-

detection of burst is initiated only when the CRC check

of the class 1a bits corresponding to , in the one-

pass receiver, is affirmative.

B. The three-pass receiver

To be able to use all class 1 bits when re-detecting the class

2 bits, the three-pass receiver imposes an extra delay of one

burst. The class 1 bits in and found in

are used to re-detect the class 2 bits in .

That is, after having received and processed burst ,

is output. The restraining method is analogous tothat used in the two-pass detector. In the three-pass case,

when re-detecting burst , there are demands on an

affirmative CRC check for the class 1a bits corresponding to

as well as for the class 1a bits corresponding to

.

C. The composite receiver

The composite receiver selects the bits of from

different detectors. The class 1 bits are selected from the one-

pass receiver. The class 2 bits are taken from the two-pass and

the three-pass detectors. The class 2 bits from the two-pass

detector are found in the set and the class 2 bits

from the three-pass detector are found in the set ,

resulting in no inherent extra delay for the composite receiver.

This receiver was originally suggested in [1].

VI. SIMULATION SETUP

Figure 7 shows the simulated transmission system with

transmitter, interferer, channel and receiver. The receiver uses

1. A study using only coded class 1a bits in the restraining is

found in a related paper [8].

F k 1[ ]

F k 1[ ]

C k 1[ ]

F k 1[ ]

B k[ ]

C k 1[ ]

B k[ ] B k[ ]

B k[ ]

F k 1[ ]

C k 1[ ] C k 2[ ]

B k 1[ ] B k 1[ ]

B k[ ]

F k2[ ]

B k 1[ ]

F k 1[ ]

F k 2[ ]

F k 1[ ]

c1

k 1[ ]

c2 k 1[ ]


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antenna diversity as is indicated in the figure.

A. The Transmitter

Each transmitter block consists of a random data generator

and, as described by the standard IS-54, a channel encoder, an

interleaver, a burst generator and a modulator. Co-channel

interference is modelled by a single interference transmitterwhich is symbol-synchronized with the desired signal. The

influence of the co-channel interference upon the received

signal is signified by the carrier to interference ratio, C/I.

B. The Channel

Figure 8 shows the simulated channel model, with Ray-

leigh fading and co-channel interference.

The base band equivalent radio channel is modelled by a

two-path FIR model. The paths being uncorrelated and com-

plex Gaussian, with the second-order statistics described by

the isotropic scattering model [7]. This model is characterized

by the parameters (,,fd), where is the delay interval, i.e.

the distance between the paths in fractions of the symbol

time, TS, see Figure 8. The relative attenuation of the second

path is , and fd is the Doppler spread of the Rayleigh fading

Figure 7. The simulation transmission system

Figure 8. The simulated channel

Transmitter Channel Receiver

TX

(Carrier)

TX

(Interferer)

RCH

RCH

C+I RX

Transmitter

Interferer

Rayleighfader, f

Rayleighfader, f

Rayleigh

fader, f

Rayleighfader, f

Rayleighfader, f

Rayleighfader, f

Rayleighfader, f

Rayleighfader, f

(1 )2

"C/I"

A

B"C/I"

(1 )2

d

d

d

d

d

d

d

d

in Hz. Receiver antenna diversity is simulated by using two

paths with channel correlation =0.7. White Gaussian noise is

added to each path, prior to receiver filtering, with a constant

signal to noise ration (Eb/N0) of 35 dB.

C. The Receiver

Between the transmitter and receiver in Figure 7, the sam-pling rate is TS/8. The receiver filters in the simulations are

base band models of actual filters. The other parts of the

receiver are described in section V.

VII. SIMULATION RESULTS

The simulations were done to compare the performance of

the multi-pass receivers with the one-pass receiver. The per-

formance of the one-, two- and three-pass receivers, together

with the composite receiver was examined considering the bit

error rate among 15.000 frames containing 159 data bits each.

For C/I values above 14 dB, 20.000 frames were used.

Only the Residual Bit Error Rate (RBER) of the class 2 bits

is considered, i.e. the rate of bit errors in frames having an

affirmative CRC check. To achieve comparable statistics, we

required the same amount of successfully received frames for

each of the four receiver structures. Thus the RBER in the

three-pass detector will actually be taken from the burst

detected by the one-pass detector if the CRC check of frame

C[k-1] is negative and the CRC check of frame C[k-2] is affir-

mative.

Time dispersion of up to one symbol time has been simu-

lated. The attenuation of the second path, , is 0 dB unless

otherwise stated. The simulations have been done on both aslow fading channel (fd = 7 Hz) and a fast fading channel (fd= 77 Hz), corresponding to vehicle speeds of 10 and 110 km/

h, respectively, at a carrier frequency of 800 MHz.

A. Analysis of class 2 bit performance

Figure 9 shows the performance gain in C/I at 1% RBER of

the different receivers, compared to the one-pass receiver,

simulated on a fast fading channel for different time disper-

sions, up to one symbol time (here, time dispersion divided

with the sample rate, 8 samples/symbol). According to the

simulation results shown in Figure 9, the class 2 bit perfor-

mances of the multi-pass receivers are more or less indepen-dent of the degree of time dispersion.


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Performance simulations with fading channels without

time dispersion and varying C/I are shown in Figures 10 and

11.

The Frame Erasure Rate (FER) is measured as the number

of erased frames per total number of simulated frames. In the

simulations, the performance gain of the re-detection scheme

tends to increase with the Doppler spread. The figures shows

that the three-pass receiver has a higher performance gain

than the two-pass receiver. At 1% RBER, the composite

receiver has a performance gain of about 2 dB compared to

the one-pass receiver but about 1 dB performance loss com-

pared to the three-pass receiver. At very low C/I values, the

Figure 9. Performance Gain in C/I at 1% RBER, for time dis-

persion in the range [0,TS], fd=77 Hz

Figure 10. Class 2 RBER; fd=7 Hz, =0

0 1 2 3 4 5 6 7 80.5

0

0.5

1

1.5

2

2.5

3

3.5

Time dispersion (tau)

Performancegain(dB

)

Threepass

Composite

Twopass

Onepass

4 6 8 10 12 14 16 18 2010

5

104

103

102

101

100

C/I (dB)

RBER

x FER

Onepass

Twopass

Composite

Threepass

composite receiver is as good as the three-pass receiver.

VIII. ASSESSMENT OF THE COMPLEXITY

The computational complexities of the multi-pass receivers

are higher than the complexity of the one-pass receiver (due

to the re-detection). About 75% of the computations in the

detector, such as the decision directed channel tracking, have

to be performed separately for each detector. This adds sub-

stantially to the increase in complexity. However, because of

the restraining, not all transition metrics of the Viterbi algo-

rithm need to be computed. This will limit the increase incomplexity. The fact that the synchronization has to be done

only once for all detectors will also limit the complexity.

The increase in computational complexity for the two- and

three-pass detectors is about 80%. The composite detector

(including all three detectors) has an increase in complexity

of about 160% compared to the one-pass detector.

Furthermore, algorithm-specific computations, e.g. the cal-

culation of feedback data, will contribute to an additional

increase in complexity that is difficult to estimate, but we

assess this as small.

Memory usage is also a consideration. An additionallyreceived burst has to be stored; interleaver memory must be

added to contain restraining information between bursts;

memory for detector-specific restraining information must be

provided, etc.

The main reason for the high complexity of the multi-pass

receiver is that decision-directed channel-tracking is used in

each detectors. If this can be avoided, e.g. by channel interpo-

lation, complexity will decrease substantially. In systems

where the channel can be considered stationary, e.g. the

Figure 11. Class 2 RBER; fd=77 Hz, =0

4 6 8 10 12 14 16 18 2010

5

104

103

102

101

100

C/I (dB)

RBER

x FER

Onepass

Twopass

Composite

Threepass


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GSM, or systems where differential detection may be used,

e.g. the PDC, the complexity is expected to be less of a prob-

lem.

IX. CONCLUSIONS

A scheme for FEC-assisted detection of the uncoded bits of

the D-AMPS format has been investigated. The performanceat 1% RBER of the uncoded bits is improved by approxi-

mately 2 dB C/I on a fading, time-dispersive channel. The

performance gain is more or less independent of the amount

of time dispersion1. Other simulations indicates that the effec-

tiveness of the FEC-assisted detection of the uncoded bits is

dependent on the interleaving employed. The improvement is

assessed to have a noticeable impact on the speech quality of

the system, since the speech distortion problem known as

warbling is largely an effect of bit errors among unpro-

tected bits.

The complexities of the studied multi-pass detectors are

relatively high. Compared to a conventional detector, theincrease in computational load is about 160%. This is mainly

due to the use of equalizers with decision directed channel

tracking. For systems where this is not required, e.g. the GSM

(Global System for Mobile Communication) or the PDC (Per-

sonal Digital Cellular) systems, the total complexity is

expected to be significantly smaller.

ACKNOWLEDGEMENT

We wish to express our gratitude towards Mr. Karim Jamal

of Ericsson Radio Systems AB for his significant contribu-

tions to the present work.

REFERENCES

[1] P. Dent, Invention Disclosure for Decodulation, Internal Document

Ericsson-General Electric, RTP/EGE/CT/Y 93:0009, May 1993.

[2] S. Lin and D.J. Costello Jr., Error Control Coding: Fundamentals and

Applications, Englewood Cliffs, NJ: Prentice Hall, 1983.

[3] EIA/TIA Interim Standard IS-54-B, Cellular System Dual-Mode

Mobile Station - Base Station Compatibility Standard, Electronic

Industries Association, 1992.

[4] R. Mehlan, H. Meyr, Combined Equalization/Decoding of Trellis-

Coded Modulation on Frequency Selective Fading Channels, Proc.

5:th Tirrenia Int. Workshop on Digital Communications, pp. 341-352,

Elsevier Science Publishers B.V., 1992.[5] S. Chennakeshu, R.D. Koilpillai, E. Dahlman, Enhancing the Spectral

Efficiency of the American Digital Cellular System with Coded

Modulation, Proc. 44:th IEEE Vehicular Technology Conference, June

8-10, 1994, Stockholm, Sweden, pp. 1001-1005.

[6] R. Sharma, W. D. Grover and W. A. Krzymien, Forward Error Control

(FEC) Assisted Adaptive Equalization for Digital Cellular Mobile

1. Simulation not reported here shows that the effectiveness of the

FEC-assisted detection of the uncoded bits depends on the inter-

leaving employed.

Radio, IEEE Transactions on Vehicular Technology, Vol. 42, No. 1,

February 1993, pp. 94-102.

[7] R.H Clarke, A Statistical theory of Mobile Radio Reception, Bell

System Technical Journal, 1968, pp. 957-1000.

[8] P. Deutgen & F. Randers, Forward-Error-Control-Assisted Detection,

Masters Thesis 1994:149E, Lule University of Technology, Div. of

Signal Processing, 1994.


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Part 2


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I. INTRODUCTION

The knowledge gained by a prior detection and decoding of

protected bits can be used in different ways depending on the

modulation employed. This paper is a continuation of part 1

and will briefly discuss different methods of doing FEC-

assisted re-detection based upon different types of modula-tion, detection and symbol constellations.

Re-detection issues for the two modulation forms Quad-

riphase shift keying (QPSK) and Differential QPSK

(DQPSK) is discussed in the following section. QPSK modu-

lation produces 4 possible symbols to be transmitted. These

symbols represent 2 bits. One or a both of these bits could be

considered as known during re-detection. Thus we distinguish

between two cases of symbol constellations containing

known bits:

Afully known symbol, where both bits of the symbol

are known.

Apartially known symbol, where one bit of the sym-bol is known.

Besides the two different modulation forms and the two

symbol constellations this paper also considers two channel

cases, withIntersymbol Interference (ISI) and without ISI. In

the former case a maximum likelihood sequence estimator

(MLSE) is used, and in the latter case a memoryless threshold

detector is used. The results are presented in a table, consist-

ing of the following sections:

QPSK modulation and channel with no ISI

QPSK modulation and channel with ISI

DQPSK modulation and channel with no ISI

DQPSK modulation and channel with ISI

II. POSSIBILITES FOR PERFORMANCE GAINS

No estimations of possible gains are made in the following

sections. Only the question if there will be a gain or not in the

detection, using a specific method, is considered.

QPSK MODULATION AND CHANNEL WITH NO ISI

This system exhibits no kind of dependencies between the

symbols. The channel has no memory, therefore a memory-

less detector is used in this section. In Subsection A re-detec-

tion of the QPSK modulated fully known symbols is

discussed and in SubsectionB re-detection of partially known

symbols is discussed.

A. Fully known symbols

All known bits are grouped into fully known symbols, and

consequently all unknown bits make up the surrounding

unknown symbols. Since there is no coupling between known

and unknown bits, the only gain possible is an enhanced

channel (phase) estimation for the surrounding unknown

symbols.

B. Partially known symbols

Due to the Gray coding normally used, there will be no

decrease in bit error probability, for the unknown bits, see

Figure 1.

The QPSK modulated symbol is equivalent to two inde-

pendent BPSK symbols each representing one bit and the

decision boundary does not change for one of the bits when

knowing the other. The only gain possible is an enhanced

channel (phase) estimation for the surrounding unknown

symbols, as in Subsection A.

Figure 1. The restraining of signal points

a) Symbol unknown b) Symol partially known;first bit known to be 1

M = 4Q

I

Q

I

(-1,-1)(-1,1)

(1,1) (1,-1) (1,1) (1,-1)

Detection of Unknown Bits Among Known Bits.


Ericsson Radio Systems ABS-164 80 Stockholm

Sweden

Lule University of TechnologyDiv. of Signal Processing

S-971 87 LuleSweden


15/41

2(3)

QPSK MODULATION AND CHANNEL WITH ISI

This system exhibits dependencies between the symbols.

The channel has memory and therefore an equalizer is used in

this section. In SubsectionA, re-detection of the QPSK modu-

lated fully known symbols is discussed and in Subsection B

re-detection of partially known symbols is discussed.


The detection of the unknown symbols will benefit from

the surrounding known symbols due to the ISI. The trellis of a

Viterbi algorithm is forced to go through a single state at each

known symbol, see Figure 2. Thus a gain is possible due to

better ISI-estimation and better channel-tracking because of

more reliable decisions.

B. Partially known symbols.

The trellis is forced to go through 2 states at the locations of

partially known symbols, see Figure 3.

There is a re-detection gain of the unknown bits in the par-

tially known symbol because of better ISI-estimation and bet-

ter channel-tracking, but there is no re-detection gain attained

by the decreased number of possible symbol-choices.

DQPSK MODULATION AND CHANNEL WITH NO ISI

This system exhibits dependences between the symbols,

Figure 2. A fully known coherent symbol

Figure 3. A partially known coherent symbol

Fully known

symbol

Unknown

symbolM = 4 Unknown

symbol

Partially known

symbol

Unknown

symbolM = 4 Unknown

symbol

due to the differential modulation. The channel has no mem-

ory and therefore a detector without equalizer is used in this

section. In this section the re-detection method of the QPSK-

modulated fully known symbols is described.


The re-detection method applied to this form of detectionwill consist of knowing (any number of) phase changes up to

some point and thereafter use an enhanced initial vector for

detecting the following unknown symbol. Based upon a

known phase shift 1, i.e. the angle between vector z[k-1] and

vectorz[k], a better estimate of the starting vector for the sub-

sequent unknown phase-shift 2 is feasible [1]. Figure 4

shows three incoming coherent symbols z[k-1], z[k] andz[k+1]. After detection, the output will be two differential

symbols representing the phase-shifts between z[k-1] and

z[k], and z[k] and z[k+1], respectively. In this case the first

symbol, representing the phase-shift 1, is known. The sec-

ond symbol represents the phase-shift 2. Due to the known

phase-shift, z[k] could be enhanced, which is explained in

Figure 5. With an enhanced start-symbol z[k], a better esti-

mate of the symbol representing the phase-shift 2 could be

gained.

Figure 5 shows the enhancement of the start-vector when

the phase shift is known to be . In Figure 5, z[k-1] and z[k]

are the incoming coherentsymbols when x[k-1] and x[k] aretransmitted. The phase-shift between z[k-1] and z[k] is known

as . Since the phase-shift is known to be , z[k-1] shifted

could be used instead of z[k], but this shifted symbol would

have equal statistics as z[k-1]. An enhanced symbol is

obtained by taking the average of the two symbols, the shifted

z[k-1] and z[k], as shown in Figure 5. Also, known symbols

following an unknown symbol may be used (in reverse) to

enhance the final vector of the unknown phase change (z[k+1]

in Figure 4). If infinitely many preceding phase-shifts to an

unknown phase-shift are known the coherent start-symbol of

Figure 4. Differential detection

/2

z[k-1]

z[k]

I

Q

M = 4 1 =

z[k+1]

1

(Known phase shift)

~ 2


16/41

3(3)

this unknown phase-shift would be correct. In this case the

differential detection would have the same performance as acoherent detector. The maximum increase is thus less than 3

dB. If an unknown symbol is surrounded by infinitely many

known symbols, the performance increase has no limit.

DQPSK MODULATION AND CHANNEL WITH ISI

This system exhibits dependences between the symbols,

due to the differential modulation and memory in the channel,

therefore an equalizer is used in this section. In SubsectionA

the re-detection method of the DQPSK modulated fully

known symbols is described and in Subsection B re-detection

of partially known symbols is described.


Instead of restraining the trellis to a single state, a single

transition is allowed out of each state, see Figure 6 showing

Figure 5. Symbol enhancement

Figure 6. A fully known differential symbol

Fully known

phase shift

z[k-1]

z[k]

z[k]=x[k]+n2

I

Q1

n

Shifted z[k-1]

n21n +( ) / 2

n2

Enhanced

vector

M = 4

Fully known

symbol

Unknown

symbolM = 4 Unknown

symbol

Q

I

Q

I

Q

I

three symbols, first symbol and last symbol are unknown

while the second symbol is fully known. One effect of this

kind of restraining is that the trellis is forced apart at the loca-

tion of a fully known symbol. As in the case of coherent mod-

ulation there is a performance gain possible due to better ISI

estimation and better channel-tracking. A performance gain

because of noise averaging is also made possible, due to thedifferential modulation.

B. Partially known symbols

There are 2 transitions out of each state, see Figure 7, lead-

ing to two possible transitions at the location of the partially

known symbol. A case dealing with both fully known sym-

bols and partially known symbols is studied and simulated in

[2].

III. SUMMARY

To achieve any performance gain by re-detection of unpro-

tected protected and unprotected bits must be sufficiently

mixed and there has to be a memory of some kind in-between

received bits. This memory can be inherent in the modulation,

e.g. in differentially encoded modulation, or it may be due to

the channel introducing ISI. It may also be a result of how the

detector works; if the detector uses adaptive channel tracking

the unprotected bits could be re-detected with a better channelestimate, due to more known pilot symbols.

IV. REFERENCES

[1] P. Dent, Invention Disclosure for Decodulation, Internal Document

Ericsson-General Electric, RTP/EGE/CT/Y 93:0009, May 1993.




Figure 7. A partially known differential symbol

Partially known

symbol

Unknown

symbol

M = 4 Unknown

symbol

Q

I

Q

I

Q

I


17/41

Part 3


18/41

1(4)

I. INTRODUCTION

This paper is written as to develop some of the ideas that

came up in part 1 and is meant to be a complement. In con-

trast to [1], this paper will treat the performance of the multi-

pass re-detection scheme for re-detecting codedbits. Due to

the interleaving [2], the unknown class 1 bits transmitted inthe current burst, belonging to the current frame, can be

helped by the error corrected class 1 bits transmitted in the

same burst, but belonging to the previous frame.

In Section II the performance of the two-pass receiver is

analyzed, using the fading channel in [1]. In Section III fur-

ther studies on the re-detection scheme, using an AWGN

channel, are done. Three probable effects on the performance

of the re-detection of the coded bits will be discussed:

1) How the use of information, on incorrectly received bits,

in the re-detection affects performance of the multi-pass

receiver.

2) How the adaptive channel tracking influences the re-detection.

3) How the soft information provided to the channel

decoder during detection influences the multi-pass receiver.

II. SIMULATIONS ON A FADING CHANNEL WITH CO-CHANNEL

INTERFERENCE

The simulations are done in order to compare the perfor-

mance of the multi-pass receivers with the one-pass receiver.

The performance of the one-, two- and three-pass receivers,

together with the composite receiver is examined considering

the bit error rate among 15.000 frames containing 159 data

bits each. For C/I values above 14 dB, 20.000 frames wereused in the simulations. Time dispersion of up to one symbol

time has been simulated. The attenuation of the second path,

, is 0 dB. The simulations have been done on a fast fading

channel (fd = 77 Hz) [1].

Figure 1 shows the performance gain in C/I at 1% RBER of

the class 1 bits, of the different receivers, in comparison with

the performance of the one-pass receiver, simulated on a fast

fading channel for different time dispersions [1]. The simula-

tion results indicate that the performance of the multi-pass

receivers considering re-detection of class 1 bits, is worse

than of the one-pass receiver and more or less independent of

the degree of time dispersion.

A. Analysis of class 1 bit performance for the three-pass

receiver.

Figure 2 shows the RBER [1] for the class 1 bits experi-

enced with the one- and three-pass receiver respectively.

According to the simulation results, the three-pass receiver

exhibits no improvement in class 1 bit performance compared

to the one-pass receiver. This confirms the theory that sinceall class 1 bits are considered known (and correct) and the

three-pass receiver in consequence is forced to detect the

class 1 bits in compliance with this information, no improve-

ment will occur. According to the simulations the three-pass

receiver is slightly worse than the one-pass receiver for low

C/I values. This may have an explanation in that for low C/I

values, an entire new (valid) CRC code word can be formed

resulting in an affirmative CRC check from the one-pass

receiver, although bit errors are found among the class 1a bits.

Figure 1. Class 1 performance gain in C/I at 1% RBER, for

time dis persion in the range [0,Ts], fd=77 Hz

0 1 2 3 4 5 6 7 81

0.5

0

0.5

Time dispersion

Performancegain(dB)

Threepass

Twopass

Onepass

Re-Detection of Coded Bits for IS-54


Ericsson Radio Systems AB

S-164 80 StockholmSweden

Lule University of Technology

Div. of Signal ProcessingS-971 87 LuleSweden


19/41


20/41

3(4)

B. Simulations with and without channel tracking

In the two- and three-pass detectors, the paths in the trellis

are forced to follow predestinated transitions at locations cor-

responding to known class 1 bits. Because the detection is

coherent with a subsequent differential decoding, at least one

transition out from each state will always be allowed (corre-

sponding to a location with a fully known phase shift) [1].

Due to this, the quick merge of nodes in the two- and three-

pass detectors is prevented where the trellises are pruned.Instead, the paths are forced apart. This side-effect, when

pruning the trellis, might affect the channel tracking, when

the channel-tracking is done according to the coherent states

in the trellis.

Figure 5 shows simulations done with channel tracking and

without channel tracking, using the AWGN channel with the

same conditions as in section C where all class1 bits are con-

sidered as known during the re-detection. According to the

simulations the channel tracking has no negative effect on the

performance of the two-pass re-detection of the coded bits.

Though the side-effect, that the trellis are forced apart, when

pruning the trellis, might still be a cause to the bad perfor-mance for the re-detection scheme in some other sense than

that the channeltracking is not negatively affected.

Figure 4. Class 1 RBER, AWGN, soft decision, [all class1,

class 1a] used

0.0001

0.001

0.01

0.1

1

0 1 2 3 4 5 6

RBER

SNR (dB)

One-passTwo-pass, class 1a+1bTwo-pass, class 1a

C. Simulations with hard and soft information to decoder

The detectors used in the simulations can produce either

soft or hard information to the decoder. The soft information

provided by the detector depends on the certainty of the deci-

sions during detection [3]. In the re-detection scheme, loca-

tions in the trellis, corresponding to known bits and also

surrounding locations, will produce re-detected bits with a

soft value implying higher certainty than the corresponding

bits have got during the one-pass detection. Thus the decodercould get erroneously detected bits with soft values indicating

high certainty. The utilization of soft information in this case

may decrease the performance gain of the decoder compared

to when hard information is used, instead of increasing it.

Therefore, differences in performance of the re-detection

scheme when using soft information versus hard information

is analyzed.

Simulations are done to compare the performance of the

two-pass receiver passing soft information versus hard infor-

mation to the decoder. Figure 6 shows the performance of the

two-pass receiver when using soft information, all class 1 bits

used as known or just the class 1a bits used as known andhard information with the same classification of known bits.

The simulations show that for increasing C/I values, the gain

increases for the two-pass receiver using hard decisions, i.e.

the performance is better than the performance of the two-

pass receiver using soft information. It seams likely that the

current soft decision algorithm is incompatible with this re-

detection scheme.

The comparisons between the performance gain of the two-

and one-pass receiver both using hard information and only

Figure 5. Class 1 RBER, AWGN, soft decision, [channel-

tracking, no channel-tracking] used

0.0001

0.001

0.01

0.1

1

0 1 2 3 4 5 6

RBER

SNR (dB)

Two-pass, soft dec., class 1a, no estTwo-pass, soft dec., class 1a, est


21/41

4(4)

the class 1a bits as known, is shown in Figure 7. According to

the simulations, the gain increases when multi-pass re-detec-

tion is used, even though the increase is almost zero.

IV. CONCLUSIONS

The soft decision algorithm employed today [3] should be

modified if a re-detection of the class 1 bits should be useful.

For low to moderate C/I values, the utilization of information

Figure 6. Class 1 RBER, AWGN, [hard, soft decision], [all

class 1, class 1a] bits used

Figure 7. Class 1 RBER, AWGN, hard decision,

class 1a bits used

1e-05

0.0001

0.001

0.01

0.1

1

0 1 2 3 4 5 6

RBER

SNR (dB)

Soft decision, class 1a+1bSoft decision, class 1aHard decision, class 1a+1bHard decision, class 1a

1e-05

0.0001

0.001

0.01

0.1

1

0 1 2 3 4 5 6

RBER

SNR (dB)

One-pass, hard decisionTwo-pass, hard decision, class 1a

on class 1a bits only in the re-detection scheme is preferred

before using all class 1 bits. The way the pruning of the trellis

is done is not affecting the channel tracking negatively, but

might affect the performance of the multi-pass receiver in

some other extent. Comparison of the performance of the

one- and two-pass receiver, using hard decision and only the

bits protected by the CRC, indicates that re-detection can beused to increase the performance of the class 1 bit detection.

The performance gain is negligeable though. The perfor-

mance for different types of interleaving has not been simu-

lated since the interleaving is stipulated by the standard IS-54.

The performance can probably be increased further by using a

more suited scheme of interleaving.

REFERENCES




[2] EIA/TIA Interim Standard IS-54-B, Cellular System Dual-Mode Mobile Station - Base Station Compatibility Standard, Electronic

Industries Association, 1992

[3] G. Bottomley, Soft information in ADC, Internal Report RTP/EGE/

CT/Y 93:0024, Ericsson-General Electric, July 1993.


22/41

Part 4


23/41

1(19)

A Functional Description of the Multi-pass

Equalizer

This document is a description of the implementation of the multi-passequalizer as proposed by Paul Dent [1]. The program is based on theREQU, the reference equalizer developed by T/ UR, mod ified in order torun together with other components of the TRX884 DVC simulator.Therefore we will describe the REQU core as w ell.


24/41

2(19)

Contents

1 The REQU core 3

1.1 The module Compute R&Phi . . . . . . . . . . . . 41.2 The module Process R&Phi . . . . . . . . . . . . 41.3 The module Synchronize . . . . . . . . . . . . . 51.4 The module Demod . . . . . . . . . . . . . . . 5

2 The Multi-pass Equalizer Scheme 6

2.1 The one-pass demodulator . . . . . . . . . . . . 62.2 The channel decoder . . . . . . . . . . . . . . . 62.3 The channel encoder . . . . . . . . . . . . . . . 72.4 The index calculators . . . . . . . . . . . . . . 72.5 The two- and three-pass detectors . . . . . . . . . . 7

3 Calculation of bit errors 11

4 Parameters defining multi-pass operation 11

5 References 12

6 The SysSim simulation model 13

6.1 The top layer . . . . . . . . . . . . . . . . 136.2 The REQU . . . . . . . . . . . . . . . . . 14

6.3 The multi-pass equalizer . . . . . . . . . . . . 156.4 Bit error calculation . . . . . . . . . . . . . . 16

7 Template for simulations 17

7.1 simparam parameters file . . . . . . . . . . . . 177.2 simparam template file . . . . . . . . . . . . . 17


25/41

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1 The REQU core

As a part of implementing and simulating the multi-pass equalizer wehave ad apted th e reference equalizer, REQU, created at T/ U to the

RBS884 product simulation environment. The REQU itself consists offour blocks: Compute R&Phi, Process R&Phi, Synchronize and Demod(Present code is found in mpass_demod), see Figure 1 and Figure 7. Themodule Compute R&Phi is intend ed as a compat ibility unit where differ-ent types of conversion is made between polar complex representation,cartesian complex representation etc., on samples delivered to theREQU. Compute R&Phi delivers two flows of data, R and Phi, to Process

R&Phi for scaling and frequency adjustment. A feedback mechanismexists in order to compensate for frequency error on the received vectorof samples. Synchronize correlates the a priori known synchronizationword with the received v ector of samples in order to find the p osition of

the synchronization word in the received vector of samples. Also thesynchronizer delivers a channel estimate wh ich is u sed in th e dem odu la-tor to initiate the adaptive process of channel estimation. Demodis thedemodulator and detector. The term demodulator is used because thetransition from the radio signal, simulated as a complex base band sig-nal, to binary (soft) bits takes place inside Demod. The detector is anadaptive maximum likelihood sequence estimator.

1.1 The module Compute R&Phi

Because the REQU is meant to be independent of the environment in

which it is used, a compatibility block exists that p erforms one of several

Figure 1. A schematic illustration of the REQU core

Sequence of

samples from

diversity

path 1

Synchronize

Synchronize

Process

R & Phi

Process

R & Phi

Compute

R & Phi

Compute

R & Phi

Demod

Feedback information on estimated frequency error

Feedback information on estimated frequency error

Sequence of

samples from

diversity

path 2

Detected

bits

Adaptive

channel

equalizer

based upon

SSVE, FSVE

or PFVE

scheme

Carthesian complex

Initial channel estimate

together with position of

known syncword in received

sequence of samples.

Polar complex


26/41

4(19)

available conversions. The REQU u ses polar comp lex samples internallyso for instance cartesian complex samples could be converted to polarcomplex samp les in th is block. Compute R&Phi also limits th e phase to a

value between 0 and 2. The samples will at this point be / 4-DQPSK,see Figures 2.

1.2 The moduleProcess R&Phi

The main function of this block is to subtract the / 4 shifts from thesamp les, see Figure 3. As these were introd uced in ord er to enhan ce theperformance of the linear transmitter filter, they are removed withoutaffecting the detection of the signal. Thereafter the signal will be ordi-nary DQPSK. Scaling of the sam ples may also be done w ith one of sev-eral available methods, e.g. scaling to normalize the rout mean square(RMS) value, scaling w ith m ax amp litude or n o scaling. We recomm endscaling with RMS because the delta metrics of the Euclidean distanceViterbi detector may otherw ise take on very h igh values, poten tially cre-

ating numerical problems. A feedback loop exist from the demodulatorand detector, Demod, back to this block and thus makes automatic fre-quency control possible, i.e. automatic adjustments to the phase basedup on information from the d emod ulator/ detector, see Figures 1 and 6.

1.3 The module Synchronize

Synchronize will try to find the beginning of the synchronization w ord inthe sampled sequence employing a cross correlation method. Since thesynchronization w ord has an cyclic autocorrelation feature the cross cor-relation between the sync word in the received sequence and the sync

word itself turns ou t to be a channel estimate. This estimate w ill be used

Figure 2. An example of a signal produced by Compute R&Phi

5 0 55

4

3

2

1

0

1

2

3

4

5pi/4 DQPSK

Inphase

Qua

drature


27/41

5(19)

as an initial value for the adaptive channel estimation scheme, see Fig-ures 1 and 6.

1.4 The moduleDemod

Demodis the m ain block of the REQU, see Figure 1 and Figu re 7. It com-prises different channel estimation schemes, a predictor to be used to

combat problems arising from the decision delay in the equalizer andthe equalizer itself.Demodmay be configu red in d ifferent w ays to act asa symbol spaced Viterbi equalizer, SSVE, with a fractionally spaced pre-filter also as a prefilter Viterbi equalizer, PFVE and as a fractionallyspaced Viterbi equ alizer, FSVE.Demodalso takes into account diversitythrough two different diversity channels (distance diversity). The infor-mation from th e diversity and the oversampled (fractionally spaced) sig-nal is combined in the calculation of the delta metrices. For moreinformation on this confer [2] and [3].

Figure 3. An example of a signal produced byProcess R& Phi

1.5 1 0.5 0 0.5 1 1.51.5

1

0.5

0

0.5

1

1.5DQPSK

Inphase

Quadrature


28/41

6(19)

2 The Multi-pass EqualizerScheme

The multi-pass equalizer consists of a modified REQU with severalinstances of theDemodmod ule extended with a chann el decoder, a non-interleaving channel encoder and special routines for extracting class 1information from the re-encoded sequence. As is, this system simulatesboth the tw o- and th ree-pass equalizer and a combination of these two,the comp osite multi-pass equalizer.

2.1 The one-pass demodulator

The one pass demodulator consists of the Demod module running inone-pass d etector modus, i.e. it makes no u se of information on the errorcorrected class 1 data bits. Different metrics may be used even thoughthe Euclidean distance metric (SqViterbi) were used in the simulations.Thus a prefilter Viterbi equalizer, PFVE, can be u sed to detect the class 1da ta bits together with th e first estimated class 2 data bits.

2.2 The channel decoder

We use the u sual channel decoder in ou r imp lementation. What we areinterested in here is the class 1 data together with the CRC check flag.Even thou gh w e can not p revent the other p arts of the multi-pass equal-izer from being executed in SysSim w hen the CRC check fails, the gen-eral idea is to only initiate the multi-pass equalizer scheme when we are

reasonably sure of having detected the class 1 data bits correctly. For th is

Figure 4. The multi-pass re-detection scheme

0

detector 1

detector 2

detector 3

"channel"

convolutional

encoder

de-interleaver

convolutional

encoder

(Two-pass)

(Three-pass)

(Class 2 only)

(Class 1&2)

f [.] corresponds to the even rows andf [.] to the odd rows in terms of the

interleaving matrix.

interleaver

F[k] = { }f [k]2

f [k],1

{ }S[k] = f [k]2

f [k-1],1

{ }S[k] =~

f [k-1],1

~f [k]2

~

S[k]~

f [k]1

f [k-1]1

f [k]2

^f [k-1]2

^

{ }f [k-1],1

~f [k]2

~

{ }f [k-1]2

~f [k-2],1

~

f [k-1]2

^

f [k-2]1

^f [k-1]1

^

0f [k-1],1

^{ }

{ }f [k-1]2

^f [k-2],1

^

{ }f [k-1]2

^f [k-2],1

^

f [k-1],1

^{ }f [k]

2

^

f [k-1],1

^{ }f [k-1]

2

^

Z-1

Z-1

Z-1

Z-1

1

2

Z-1

B[k] ={ }b[k]

^B[k-1]

convolutional

decoder

{ }f [k-1]2

^f [k-1],1

^F[k-1] =^

if (CRC[i-1] is OK & CRC[i-2] is OK)

if CRC[i-1] is OKprepare

feedback

prepare

feedback

(Two-pass)

(Three-pass)


29/41

7(19)

purpose we use the CRC check flag from the decoder. Keep in mindthough that the CRC check involves the class 1a bits only. There is aninherent p roblem with th is because we may u se incorrect information on

class 1b bits when we experience (heavy) garbling of class 1b bits with-out having a CRC fail situation for th e class 1a bits.

2.3 The channel encoder

After decoding a frame from the ordinary demodu lator/ detector we re-encode it under the assumption of having received all class 1 bits cor-rectly. Observe that the usual interleaving is omitted to avoid a timedelay.

2.4 The index calculators

The index calculator codes all channel encoded class 1 data bits, or a p ar-tition thereof, received from the channel encoder in such a manner thatall zeros are converted to -1 and all ones are converted to +1 while at thesame time setting all class 2, SACCH and CDVCC bits to zero. Also itinserts zeros in p lace of the synchronization w ord. This results in a 312bit vector reflecting all data bits in the received frame, i.e. 16 bits UCH,28 bits SYNC, 122 bits UCH , 12 bits SACCH, 12 bits CDVCC an d 122 bitsUCH. When the CRC check is non-affirmative, the index calculatordelivers all zeros, reflecting that no inform ation is sup plied for the corre-spon ding frame. The tw o-pass index-calculator sets every second row tozero to indicate the lack of knowledge of these bits du e to the interleav-

ing. The three-pass index-calculator interleaves the current burst withthe previous to create a frame, which produces a time-delay. The two-pass index calculator only considers the CRC for the current frame,while the three-pass index calculator also has demands for an affirma-tive CRC for the previous fram e.

2.5 The two- and three-pass detectors

When re-detecting B[k] consisting of the parts {c1[k-1], c2[k]} the frameC[k-1] holding {c1[k-1], c2[k-1]} has been produced by the secondencoder, see Figu re 4. Thus only the class 1 bits in the fraction c1[k-1] can

be used as known . The three-pass re-detection hold s a time-delay whichimplies that when re-detecting B[k] both C[k-1] and then C[k] have beenproduced by the second encoder. All class 1 bits can be used as known,where c1[k-1] is taken from C[k-1] and c2[k] from C[k]. See Figures 4 and7.

The two pass detector makes use of information on class 1 data bits com-puted in the one-pass detector and converted during the process ofchannel encoding and index calculation (see above). Only th e Euclideandistance metric Viterbi is modified to make use of this information andthe two- and three-pass detectors may only be run in this mode. The

idea is that given inform ation on convolut ional- and block coded class 1


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8(19)

data bits, i.e. data bits that we may check for correctness, we prune thetrellis to reflect that w e for some samples know wh at ph ase shifts shouldoccur and thu s may increase the ability to detect the interm ingling class

2 data bits correctly. In the actual pruning process we use the 312 bitindex vector, that is aligned with the 312 bit information part of thereceived fram e, wh ile processing th e received signal. At all locations inthe ind ex vector w here the valu e differ from zero, i.e. positions held byknown bits, we assume w e know th ese bits with absolute certainty thussetting the sp ecific -metr ics to infinity. This forces the Viterbi algorith mto exclude all paths not corresponding to the known phase shiftsthrou gh the trellis.

In the three-pass case we perform a regular interleaving after havingencoded the data. The received frame on w hich the re-detection process

is initiated mu st therefore be delayed one frame. All class 1 bits are con-sidered to be known in this scheme.

The composite multi-pass detector combines the class 2 bits from thetwo-pass d etector an d the th ree-pass d etector. The class 1 bits are takenfrom the one-pass receiver. After having received B[k] by the one-passreceiver, both a re-detection ofB[k] by the two-pass detector and a re-detection ofB[k-1] by the three-pass d etector, is initiated. From these tw ore-detected bursts the class 2 bits are taken to form th e class 2 bits frac-tion ofF[k-1]. B[k], holds {c1[k-1], c2[k]} and B[k-1], holds {c1[k-2], c2[k-1]}thus the class 2 bits ofc1[k-1] are taken from the two-pass detector and

the class 2 bits ofc2[k-1] are taken from the three-pass d etector, see Fig-ures 4 and 7.

For a fur ther discussion on this su bject, please see [4].

2.5.1 The restraining of the trellis implemented in the two- and

three-pass detectors

The detection is done by means of a coherent Viterbi equalizer with asubsequent differential decoder, implying coherent symbol detectionwith the aid of the trellis, and then d ifferential decoding of the coherentsymbols.

In differential quadrature phase shift keying (DQPSK), as used in D-AMPS, the information lies in the differential phase shifts rather th an inthe absolute ph ase of the signal. A set of information bits correspond toa certain phase shift, see Figure 5.

If we, pr ior to detecting a d ifferential symbol, have knowledge of oneor both of the correspond ing bits we m ay restrain, or p rune, the (num berof) transitions in th e trellis.

Since some causes to an erroneous detection have been left out by

means of the pruning, we may re-detect a differential symbol with an


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increased possibility of m aking the correct d ecision. In th e specific envi-ronment of IS-54 we experience both fully known and partially knownsymbols. Due to u nequal p rotection of transmitted bits, we m ay u se theprotected bits, if they are successfully received, for restraining the trellisin a subsequent re-detection of bits with no protection. Based upon thedepth of the interleaving scheme employed, such a re-detection may beperformed on the currently received burst, or on previously received

bursts. Also the effectiveness of this multi-pass re-detection scheme isdep endent on the intra-burst interleaving employed. From the p oint ofview of this method , the performan ce gain is depend ent of the pattern ofinterspersing (fully or partially) known sym bols among un known sym-bols (symbols with a lower degree of error p rotection).

2.5.2 The influence that the restraining has on the trellis

The detector is a TS / 2 fractionally sp aced maximum likelihoodsequ ence estimator (MLSE), with a 4-state Viterbi algor ithm (VA). In th etwo-pass and three-pass detectors the trellises of the VA are forced to

take pred estinated p aths correspond ing to know n class 1 bits. The differ-ential detection of the DQPSK signal is imp lemented by a coherent Vit-erbi detector with a subsequent differential decoder. This implies thatparallel transitions in the trellis are equal since they represent an equaldifference in phase. If bits corresponding to a certain phase shift areknown, the trellis is restrained so that only transitions corresponding tothis known phase shift are allowed. Due to this, the quick merge ofnodes in th e two- and three-pass d etectors are prevented w here the trel-lises are prun ed. Instead , the paths are forced ap art.

Figure 5. A symbol represented by two bits

Q

I

(-1,-1)

(-1,1)

(1,1)(1,-1)

/2

3/2

0

~

~~

~


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5 References

[1] P. Dent, Invention disclosure of Decodulation, Internal Docum entEricsson-General Electric, RTP/ EGE/ CT/ Y 93:0009, May 1993.

[2] K. Jamal, Equalization for the ADC standard, Internal report T/ U91:260, 1991.

[3] K. Jamal, Fractionally Spaced Viterbi Equalization, Internal reportT/ BU 92:075, 1992.

[4] P. Deutgen & F. Rand ers, Forward-Error-Control-Assisted Detec-tion, Masters Thesis 1994:149E, Lule University of Technology,Div. of Signal Processin g, 1994.


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7 Templates for simulation

7.1 The simparam parameters file

FADE 7 77TAU 0 2 4 8CIR 4 5 8 10 12 14 16 18SNR 35FREQ_ERR 0CH_SEP 0STEP 3101

7.2 The simparam template fi le

sim build

/* Set RATE */

set **/RATE 1

/* Set simulation parameters */

set **/DOPPLERFREQ FADEset **/dopplerFreq FADEset **/snrDb SNRset **/cOverIdB1 CIR

set **/delay TAUset **/ST1/freqErr FREQ_ERRset **/channelSep1 CH_SEP

/* Set other parameters */

set /LogStat/SimName SIM_NAMEset /LogStat/SimRunName SIM_RUN_NAMEset /LogStat/ResultPathName RESULT_PATH_NAME

set /mpass_dvc/ratrx?/mf1/INFILENAME rat/filter/butt4_165e2_194e3.hzset /mpass_dvc/ratrx?/mf2/INFILENAME rat/filter/iir2gaus3_194e3.hz

cd /mpass_dvc/dfiltAsource dfilt/dfilt.inicd /mpass_dvc/dfiltBsource dfilt/dfilt.inicd /

/*---------------------------------------------------Parameters used to force dfilt to produce (i,q)samples,i.e. carthesian complex samples instead of any

obfuscated (rss,phi) samples that we would have to


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invert anyway.*--------------------------------------------------*/set /mpass_dvc/dfiltA/t2/bypass TRUEset /mpass_dvc/dfiltA/t2/eqmode FALSE

set /mpass_dvc/dfiltB/t2/bypass TRUEset /mpass_dvc/dfiltB/t2/eqmode FALSE

/*---------------------------------------------------Parameters used to force dfilt to skip the suspiciousladder-mode. Instead we use a simple filter whichonly amplifies the signal with a constant gain = b0*--------------------------------------------------*/set /mpass_dvc/dfilt?/mf22/bypass FALSEset /mpass_dvc/dfilt?/mf22/ladderMode FALSEset /mpass_dvc/dfilt?/mf22/b0 1000

/*---------------------------------------------------The correlation between the carrier and the interfererin the two ray model. This together with the amplitudeof the interferer relative the carrier amplitude maybe used to simulate a flat non fading channel(correlation=1, amplitude=-200).See below for FADE == 0.*--------------------------------------------------*/set **/CORR/correlation 0.7

/*---------------------------------------------------Statistical interval is now set to 200. That meansthat the number of simulations will be broken downinto small pieces on which we may build more robuststatistics*--------------------------------------------------*/set **/statInterval 200set **/COVERICHA/skipNr 100set **/COVERICHB/skipNr 100set **/TXCMOD/skipNr 100

/*---------------------------------------------------Since dmt comprises a program written in pascal aswell as one written in C++, neither one being possibleto instantiate, we must choose different models for

the modulator of the carrier and the modulator of theinterferer. For the carrier we choose the C++ model(implementation model TRUE) and for the interferer thePascal program (implementation model FALSE).*--------------------------------------------------*/set /mpass_dvc/dmtCarr/imp_model Trueset /mpass_dvc/dmtCarr/imp_length 12set /mpass_dvc/dmtCarr/ampl 5100set /mpass_dvc/dmtInt/ampl 5150

/* Set dequdect parameters */

cd /mpass_dvc/dequdect


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set crphi?/ConversionMode 1set prphi?/AGCMode 1set sync?/FseFactor 2set sync?/ExpectedPos 244

set sync?/CorrelationMode 1set sync?/WndLen 32set sync?/WeightFactor 1.5

/* Set mpass_equ parameters */

cd /mpass_dvc/dequdect/mpass_equ/one_pass_ddtset Weighting TRUEset IsiLp 0.02set FeedbackMode 1set FseFactor 2set NmbDivCh 2

set ExpandedMode 0set SLength 1

cd /mpass_dvc/dequdect/mpass_equ/two_pass_ddt/demodset Weighting TRUEset IsiLp 0.02set FeedbackMode 1set FseFactor 2set NmbDivCh 2set ExpandedMode 3set SLength 1

cd /mpass_dvc/dequdect/mpass_equ/three_pass_ddt/demodset Weighting TRUEset IsiLp 0.02set FeedbackMode 1set FseFactor 2set NmbDivCh 2set ExpandedMode 3set SLength 1

cd /

/* BER calculation */

set **/errDataOut 0set **/statInterval 200set **/skipNr 100

cd /mpass_dvc/ber_calcset one_pass_ber_calc/skipNr 100set two_pass_ber_calc/skipNr 100set three_pass_ber_calc/skipNr 101cd /

#if FADE==0set **/CORR/correlation 1.0

set **/amplitude -200


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set **/statInterval 200#endif

#if FADE==7

set **/statInterval 500#endif

#if FADE==77set **/statInterval 200#endif

sim cons

/* Simulate */sim starts

s STEPsim stopexit

master thesis: forward error control assisted detection

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