Photonic measurement of microwave frequency using a siliconmicrodisk resonator

Li Liu, Fan Jiang, Siqi Yan, Shucun Min, Mengying He, Dingshan Gao, Jianji Dong n

Wuhan National Laboratory for Optoelectronics & School of Optical and Electrical Information, Huazhong University of Science and Technology,Wuhan 430074, China

a r t i c l e i n f o

Article history:Received 29 July 2014Received in revised form5 September 2014Accepted 9 September 2014Available online 23 September 2014

Keywords:Microwave photonic filter (MPF)Microdisk resonator (MDR)Microwave frequency measurementIntegrated opticsOptical microwave signal processing

a b s t r a c t

A simple photonic approach to the measurement of microwave signal frequency with adjustablemeasurement range and resolution is proposed and demonstrated. In this approach, the unknownmicrowave signal is converted to an optical signal with single sideband modulation. Subsequently,a notch microwave photonic filter (MPF) is implemented by employing a high-Q silicon microdiskresonator (MDR). The MPF is tunable by changing the frequency interval between the optical carrier andthe MDR notch so as to obtain different amplitude responses. A fixed frequency-to-power mapping isestablished by obtaining an amplitude comparison function (ACF) of the microwave power ratio and themicrowave frequency. A proof-of-concept experiment demonstrates a frequency measurement range of10 GHz, with measurement error of 70.1 GHz. Different frequency measurement ranges and resolutionsare also discussed.

& 2014 Elsevier B.V. All rights reserved.

1. Introduction

Conventional microwave architectures usually make the measure-ment systems very bulky and costly. So photonics has been rapidlypenetrating into the radar systems for the transmission and proces-sing of microwave signals, with its competitive advantages of lightweight, inherent low-loss, high bandwidth and immunity to electro-magnetic interference (EMI) [1,2]. For frequency measurement of aunknown signal, microwave frequency measurement techniquesbased on optical or microwave power monitoring have attractedwide attention, owing to their large frequency measurement rangeand high measurement resolution [3–16]. For these various schemes,the basic principle is to construct an amplitude comparison function(ACF) which is the ratio between two different optical or microwavepower functions. However, different problems exist. Utilizing multi-ple modulators may degrade the measurement accuracy [4–6] sincethe modulators don’t perfectly match. In [7,8], measurement errorincreases when the Mach–Zehnder modulator (MZM) was notperfectly biased to obtain a complete suppression of the opticalcarrier. The major limitation of the approaches in [9,10] is that it stillrequires two photodetectors (PDs) to get the ACF, which lead to theincreased noise of the PDs and the measurement errors. For a givensystem, the measurement range and resolution are changeless in [11].

However, the systemwith tunable measurement range and resolutionis significant for many applications.

In this letter, a flexible and simple photonic approach to micro-wave frequency measurement with adjustable measurement rangeand resolution is proposed and experimentally demonstrated basedon a low-loss silicon microdisk resonator (MDR). Different from mostof the previous approaches, the system only requires one lasersource, one phase modulator and one PD. Hence, the measurementerror and cost of the system are affirmatively reduced. What’s more,adjustable frequency measurement range and resolution are achiev-able by tuning the wavelength interval to get different ACFs.

2. Operation principle and experimental setup

The schematic diagram for the proposed photonic microwavefrequency measurement system is shown in Fig. 1(a). A tunable lasersource (TLS) emits a continuous wave (CW) beam corresponding toangular frequency ω0, which is modulated by an optical single-sideband (OSSB) modulator driven by an unknown radio-frequency(RF) signal ωRF. Subsequently, the OSSB signal is sent to the MDRwhich acts as a tunable notch filter. As a result, the optical spectrumof the MDR is mapped to RF spectrum by the PD. As the frequencyspacing between the TLS wavelength and MDR resonant wavelengthis f, a response of microwave photonic filter (MPF) with the notch at fis achieved. By shifting the optical carrier, different MPFs can beobtained to calculate the corresponding ACFs. Finally, we can get afixed frequency-to-power mapping in the post processing.

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Optics Communications

http://dx.doi.org/10.1016/j.optcom.2014.09.0300030-4018/& 2014 Elsevier B.V. All rights reserved.

n Corresponding author. Tel.: þ86 18696172900.E-mail address: jjdong@mail.hust.edu.cn (J. Dong).

Optics Communications 335 (2015) 266–270

To theoretically demonstrate the operation of the proposed mea-surement system, mathematical deduction has been given as follows.Assume that the optical carrier is modulated in a phase modulator(PM) by the RF signal. In the case of phase modulation, the outputoptical field of the PM can be described by

EoutðtÞ ¼ E0ej½ω0tþ γ cos ðωRFtÞ� ð1Þ

where E0 is amplitude of the input optical field and γ is the phasemodulation depth. Neglecting the high order side-band, the outputoptical field of the PM under a small signal model can be described by

Eout tð Þ ¼ E0 jJ1 γ� �

ej ω0 �ωRFð Þtþ J0 γ� �

ejω0tþ jJ1 γ� �

ej ω0 þωRFð Þth i

ð2Þ

where Jn is the nth-order Bessel function of the first kind. Thenwe use a rectangular bandpass filter to eliminate one sideband

(ω0�ωRF). After the signal is transmitted in the CMRR (or MDR)with an amplitude transmission function of H(ω), each frequencycomponent will multiply a different frequency weight of H(ω), sothe output optical field can be described as

E ωð Þ ¼ 2πE0 J0 γ� �

H ω0ð Þþ jJ1 γ� �

H ω0þωRFð Þ� � ð3ÞAfter detecting of the square-law photo-detector (PD), and

neglecting the J12 term, we obtain the alternative current (AC) term

of the current, which is expressed by

iAC ω0ð Þp4π2jP0J0 γ� �

J1 γ� �

Hn ω0ð ÞH ω0þωRFð Þ ð4Þwhere P0 is the optical power, expressed byP0 ¼ jE0j2. So theelectrical responses of iAC (ω1) and iAC(ω2) can be similarlyobtained when the TLS is tuned to ω1 and ω2, as shown in Fig. 1(b).

Based on Eq. (4), the ACF between the detected two currentscan be expressed as

ACF ωRFð Þ ¼ iAC ω1ð ÞiAC ω2ð Þ ¼

Hn ω1ð ÞH ω1þωRFð ÞHn ω2ð ÞH ω2þωRFð Þ ð5Þ

As can be seen from Eq. (5), the theoretical ACF is calculatedbased on the amplitude response of MDR while the measured ACFis based on the RF responses from the PD. The difference betweenthe theoretical and measured ACFs represents the measurementerrors.

The relationship between the power ratio and microwavefrequency is unique in the monotonic increasing region ωRF1=2π

��ωRF2=2πÞ of the ACF, as shown in Fig. 1(c). So the microwavefrequency can be accurately calculated from the function of ACF inEq. (5). The feature of our technique is the employment of thehigh-QMDR, which results in different microwave power penaltiesof high resolution for the two laser wavelengths, leading tononunity power ratios. Besides, the frequency measurement rangeand resolution can be adjusted if the TLS wavelength spacing istuned. Therefore, to have a specific measurement range or resolu-tion, one only need to tune the TLS wavelength.

To experimentally demonstrate the feasibility of our microwavefrequency measurement scheme, an MDR has been fabricated on

Fig. 1. (a) Schematic diagram for microwave frequency measurement. (b) and(c) are the measured MPF responses and ACF, respectively.

Fig. 2. SEM images of the silicon MDR (a), the grating coupler (b), zoom-in views of the coupling region (c) and the grating coupler (d).

L. Liu et al. / Optics Communications 335 (2015) 266–270 267

the commercial silicon-on-insulator (SOI) wafer. Fig. 2(a) showsthe scanning electron microscope (SEM) image of the fabricatedMDR whose radius is 10 μm. The thickness of top silicon layerof SOI wafers is 340 nm and the upper silicon layer was etcheddownward for 240 nm to form a silicon ridge waveguideand input/output grating couplers, through inductively coupledplasma (ICP) etching (Oxford Instruments Plasmalab System100).The waveguide width of the bus waveguide is about 500 nm andthe coupling gap between the bus waveguide and MDR is about150 nm. The zoom-in coupling region is shown in Fig. 2(c).We employ the vertical grating coupler as shown in Fig. 2(b) tocouple the optical signal from fiber to the silicon waveguide, andthe zoom-in grating coupler is shown in Fig. 2(d). The period, dutycycle and coupling loss for a single side of the grating couplers are630 nm, 56% and 5 dB, respectively.

Fig. 3 shows the measured transmission spectrum of the MDRby using an optical spectrum analyzer (AQ6370B). It is clear to seethat the zoom-in notch region selected in our experiment has alow loss of 12 dB, a relatively high extinction ratio (ER) of 15 dBand a relatively high Q factor of 1.5�104 at the resonancewavelength 1554.17 nm, respectively.

3. Experimental results and discussion

A proof-of-concept experiment for photonic microwave fre-quency measurement has been performed as shown in Fig. 4. Thephase modulator (PM) is used to generate optical double-sideband(ODSB) modulation. A forward erbium doped fiber amplifier(EDFA1) is used to amplify the input optical power. The tunableoptical filter is used to remove the negative first-order sidebandfrom the ODSB signal to produce an OSSB signal. A polarization

controller (PC2) is required since the silicon waveguide can beoperated only in transverse electrical (TE) mode. Then the OSSBsignal is sent into the MDR by the vertical grating coupler whichhas a 3-dB coupling bandwidth of about 30 nm in C band. Afteramplification by the backward EDFA2, the optical signal is con-verted to electrical signal forming a MPF response by the PD with abandwidth of 40 GHz. When the TLS operates at different wave-lengths, the corresponding MPF responses are measured by thevector network analyzer (VNA, Anritsu MS2028C).

Although the PM can be instead of an intensity modulator, it isknown to all that the PM works better because of its bias-drift-freeoperation and low insertion loss. Fig. 5 illustrates the spectrum ofOSSB modulation. For example, if an RF signal at 19.5 GHz is sendto the PM, one can observe an ODSB signal at the PM output portas the blue line in Fig. 5. Subsequently, a tunable optical filter isused to remove one sideband. The transmission response of theoptical filter is shown as the green line. In order to get betterperformance of beat frequency, the optical carrier is located at therising edge of the optical filter. Thus, OSSB modulation is achievedwith the spectrum shown as the red line. The power of the opticalcarrier and positive sideband is almost the same. The optical filterhas special influences on the frequency measurement. First, itsbandwidth must be larger than the maximum frequency of themeasured microwave signal. For example, if the measurementrange is from 10 to 20 GHz, the filter bandwidth need to be largerthan 20 GHz. Second, as the optical carrier is located at the risingedge of the optical filter, the slope of the filter edge should be assteep as possible to get better performance of beat frequency.

Fig. 3. Measured spectrum of the low-loss MDR.

Fig. 4. Experimental setup of the proposed scheme. TLS: tunable laser source, PC: polarization controller, PM: phase modulator, EDFA: erbium doped fiber amplifier, ATT:attenuator, PD: photodetector, EA: electrical amplifier, VNA: vector network analyzer.

ODSB SignalFilter ShapeOSSB Signal

40 GHz

Fig. 5. Optical single-sideband signal generation using an optical bandpass filter.Green line: the transmission spectrum of the optical filter; blue line: the opticalspectrum of the ODSB signal; red line: the optical spectrum of generated OSSBsignal. (For interpretation of the references to color in this figure legend, the readeris referred to the web version of this article.)

L. Liu et al. / Optics Communications 335 (2015) 266–270268

As shown in Fig. 6, by tuning the TLS central wavelength toλ1(1554.098 nm), λ2 (1554.066 nm) and λ3 (1554.018 nm) respec-tively, we obtained three MPF responses whose notch frequenciesare 9, 12 and 19 GHz, respectively. The following measurementranges and resolutions are determined by these notch frequencies.

In the experiment, when the TLS is firstly operating at λ1and λ3respectively, the corresponding powers at different microwavefrequencies are measured to calculate the ACF. The measured ACFwith a notch at 9 GHz agrees well with the one calculated theoreti-cally, as shown in Fig. 7. Based on Eq. (5), a look-up table is set up andthe microwave frequency is estimated by using the monotonicincreasing region of the measured ACF13. Therefore, the measurementrange is 9–19 GHz (Band 1) with an error of 70.2 GHz, as shown inFig. 8. The measurement error is mainly induced by the system noiseand power fluctuations of the TLS.

The proposed scheme has two important features. First, thisapproach is flexible to adjust the measurement range. As themeasurement range is limited by the central frequency of the twonotch MPFs, it can be changed by simply tuning the TLS wavelength.For example, we can tune the short wavelength of λ3 to λ2 whilekeeping the longer wavelength of λ1 unchanged. As a result, asmaller measurement range of 9–12 GHz defined as Band 2 can beachieved, as shown in Fig. 9. Second, the resolution could also betuned. For instance, Band 2 has been measured with a betterreduced error of 70.07 GHz compared with Band 1. It can be seenthat there is a trade-off between the resolution and measurement

range. Namely, a relatively smaller measurement range would resultin a higher resolution.

Moreover, by introducing a third wavelength, the measurementresolution of the proposed system can be significantly improved whilekeeping the wide measurement range. The fundamental principle hasbeen successfully demonstrated in our experiment. It can be observedthat two independent measurement ranges with each having a higherresolution could be achieved by employing three designed wave-lengths. Therefore, by combining the two separate measurementranges, a larger measurement range with a higher measurementresolution can be realized. For example, three wavelengths at λ1, λ2and λ3 are chosen to verify above conclusion. As discussed in theprevious section, Band 2 (9–12 GHz) has been measured with an errorof 70.07 GHz through employing λ1 and λ2. With a measurementerror of 70.1 GHz, the wavelengths at λ2 and λ3 would offer anothermeasurement range of 12–19 GHz (Band 3) as shown in Fig. 10. Hence,the total measurement range is equal to Band 1 (9–19 GHz) with asmaller error of 70.1 GHz as shown in Fig. 11.

5 10 15 20

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: 1554.098 nm: 1554.066 nm: 1554.018 nm

aλ

bλ

cλ

Input microwave frequency (GHz)0

Nor

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Res

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e (d

B)

Fig. 6. Measured MPF responses for three different wavelengths of λ1, λ2 and λ3.

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Fig. 7. Theoretical and measured ACF13 when the system is operating at λ1 and λ3,respectively.

8 10 12 14 16 18 20

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0.1

0.2

8 10 12 14 16 18 208

10

12

14

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Err

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d fr

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z)Fig. 8. Measurement range (a) and error (b) when the system is operating at λ1 and λ3.

6 9 12 15 18-40

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Pow

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Fig. 9. Theoretical and measured ACF12 when the system is operating at λ1 and λ2,respectively.

L. Liu et al. / Optics Communications 335 (2015) 266–270 269

Through this method, the frequency measurement of microwavesignal with low frequency can also be realized. It is influenced bytwo major factors. First, when the measured microwave frequencyis low, the slope of the filter edge should be as steep as possible togenerate a perfect OSSB signal. Second, the Q factor of the MDR canbe significantly improved to 1�105 or even higher. And it is nodifficulty in fabricating such an MDR. By using a higher-Q MDR,we can realize different MPFs of low central frequencies so as toobtain different amplitude responses. Hence, our approach can alsomeasure microwave signal with low frequency. In practical applica-tions, we can employ multiple measurement steps to improve themeasurement resolution as shown in Table 1. For example, in thefirst step (Step 1), the two initial TLS wavelengths are chosen to beλ1 and λ3 respectively. Thus we can achieve a wide measurementrange of 9–19 GHz while the error is about 70.2 GHz. Assume thatthe predicted microwave frequency is around 10 GHz. As frequencyof 10 GHz belongs to Band 2, we shift the wavelength λ3 to λ2 in thesecond step (Step 2) to improve the resolution. Thus the obtainedACF12 could provide a measurement range of 9–12 GHz while itserror reduced to 70.07 GHz. In this case, the microwave frequency

is more accurately measured as 1070.07 GHz. If a higher resolutionis needed, we can accordingly change the wavelength to improvethe measurement accuracy.

4. Conclusions

We have proposed a simple and integrated photonic-assistedapproach to microwave frequency measurement with adjustablemeasurement range and resolution. Experiments were performedto demonstrate different frequency measurement ranges and resolu-tions. A trade-off between the measurement range and resolutionexists, but it could be eliminated by using multiple measurementsteps to provide a significantly improved measurement resolutionwhile maintaining a large measurement range. In addition, since onlyone laser source, one modulator and one PD were needed, thesystem complexity and cost are obviously reduced.

Acknowledgements

This work was partially supported by the National BasicResearch Program of China (Grant no. 2011CB301704), the Pro-gram for New Century Excellent Talents in Ministry of Education ofChina (Grant no. NCET-11-0168), a Foundation for the Author ofNational Excellent Doctoral Dissertation of China (Grant no.201139), the National Natural Science Foundation of China (Grantno. 60901006, and Grant no. 11174096), and the FundamentalResearch Funds for the Central Universities, HUST: 2014YQ015.Graduates' Innovation Fund of Huazhong University of Science &Technology (Grand no. 01-09-070087). The authors would like tothank Prof. Jinsong Xia and Dr. Qingzhong Huang in the Center ofMicro-Fabrication and Characterization (CMFC) of WNLO for theassistance in device fabrication.

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10 12 14 16 18 20

-40

-30

-20

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0

10

20

30

40

Input microwave frequency (GHz)

Pow

er r

atio

(dB

)23

23

Fig. 10. Theoretical and measured ACF23 when the system is operating at λ2 and λ3,respectively.

8 10 12 14 16 18 208

10

12

14

16

18

20

8 10 12 14 16 18 20

-0.2

-0.1

0

0.1

0.2

Input microwave frequency (GHz)

Err

or (G

Hz)

Mea

sure

d fr

eque

ncy

(GH

z)

Fig. 11. Measurement range and error for the three wavelengths of λ1, λ2 and λ3.

Table 1Steps to accurately measure the microwave frequency around 10 GHz.

Stepnumber

TLS wavelength(nm)

Measurement range(GHz)

Error(GHz)

1 λ1, λ3 Band 1: 9–19 70.22 λ1, λ2 Band 2: 9–12 70.07

L. Liu et al. / Optics Communications 335 (2015) 266–270270