Eindhoven University of Technology

MASTER

Horizontale afbuiging met symmetrisch heen- en teruglopende afbuigstroom voor TV-ontvangers

Frensch, A.J.

Award date:1984

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AFDELING DER ELEKTROTECHNIEK

TECHNISCHE HOGESCHOOL EINDHOVEN

VAKGROEP TELECOMMUNICATIE EC

HORIZONTALE AFBUIGING MET SYMMETRISCH

HEEN- EN TERUGLOPENDE AFBUIGSTROOM VOOR

TV-ONTVANGERS

(HORIZONTAL DEFLECTION IN TV-RECEIVERS

USING A SYMMETRIC CURRENT)

door: AdJ. Frensch

van januari 1983 tot maart 1984

Afstudeerhoogleraar: Prof. Dr. I.C. Arnbak

Begeleiders: Dr. Ing. U.E. Kraus,

Hoofdindustriegroep Video,

Nederlandse Philips Bedrijven B.V.,

TV-Lab., Signal Processing Groep.

Ir. A.P. Verlijsdonk.

De afdeling der elektrotechniek van de Technische Hogeschool

Eindhoven aanvaardt geen verantwoordelijkheid voor de inhoud

van stage- en afstudeerverslagen.

CONTENTS

SAMENVATTING

ABSTRACT

-1-

LIST OF SYMBOLS AND ABBREVIATIONS

1 • INTRODUCTION 1. I The horizontal deflection circuit in the con­

ventional colour TV receiver 1.2 Power dissipation, high voltage and radiation

problems in the standard horizontal deflection circuit at different line frequencies

1.3 Symmetrie horizontal deflection currents

2. THEORY 2.1 The effects of sinusoidal horizontal deflection

currents on the geometry of the picture 2.1.1 Introduetion 2. 1.2.Bounds on the deflection angle ~D 2.1.3 Bounds on the deflection current ih 2.1.4 Sinusoidal currents for horizontal

deflection

page

3

5

7

10 12

19

24

28 30

30 32 42 44

2.2 Generation of the sinusoidal current 48 2.2.1 The ser resonance circuit 49 2.2.2 The parallel resonance circuit 50 2.2.3 Removal of pin-cushion effect: Qualitative 52

consideration of an amplitude modulating signal.

2.3 Deflection current phase modulation due to 54 cross-coupling between the deflection circuits 2.3.1 Variation of e 1n the series resonance 55

circuit 2.3.2 Variation of e in the parallel resonance 56

circuit 2.4 Alignment of the lines scanned in opposite direction 58 2.5 The effects of the symmetrie deflection current 67

on the scanning raster

3. HARDWARE REALISATION 71 3.1 Design of a breadboard model for sinusoidal 71

horizontal deflections currents 3.2 The line memories and blanking control (CONTROL) 75 3.3 The memory enabling (VIE/VOE) unit 81 3.4 The clock and state signals (CLOCK/ST) unit 87 3.5 The sinusoidal deflection current (SIN) unit 91 3.6 The horizontal deflection current-measurement 93

processing (IID•i PROC) unit

-2-

4. DISCUSSION AND EXPERIMENTAL RESULTS 4.1 Choice of amplitude and generation methad

of the sinusciclal deflection current 4.2 Choice of alignment 4.3 Power dissipation, high voltage and radiation

in the sinusciclal driven horizontal deflec­tion circuit

CONCLUSIONS

REPERENCES

page

96 96

100 102

105

107

-3-

SAMENVATTING

Het aftastschema dat wordt gebruikt in de huidige TV-ontvangers wordt tot stand gebracht door een zaagtandvormige stroom in zo­wel de horizontale als verticale afbuigspoel. Voor het horizon­tale afbuigcircuit heeft deze golfvorm de volgende nadelen: - Grote verrnogensdissipatie; - hoge spanningsbelasting van de componenten in het horizontale

afbuigcircuit; - straling. Deze nadelen zullen nog toenemen als de lijnfrequentie wordt ver­hoogd om grootvlakflikker en interlijnflikker te elimineren.

Het functiemodel dat hier wordt beschreven gebruikt een sinus­vormige horizontale afbuigstroom gegenereerd met behulp van een serieresonantiecircuit. Omdat zowel de stijgende als dalende gedeeltes van deze stroom worden gebruikt om het scherm af te tasten, moet, om een hornogeen aftastschema te verkrijgen waarin de aftastgedeeltes van het A-raster en het B-raster elkaar niet snijden, de verticale afbuigstroom worden veranderd in een trap­jesvormige stroom. Doordat het scherm in tegengestelde richting­en wordt afgetast, zal de video-informatie om de andere lijn moeten worden omgedraaid. De omdraaioperatie en het richten van de in tegengestelde richting afgetaste lijnen is digitaal gerea­liseerd op kleinsignaal-nivo. De correctie van de speldenkussenvervorming is ook op kleinsig­naal-nivo gerealiseerd, door de samples van het amplitudegemodu­leerde signaal, dat gebruikt wordt om de eindversterker van het horizontale afbuigcircuit te sturen, op te slaan in een EPROM. De hoge spanning voor de beeldbuis wordt onafhankelijk van het horizontale afbuigcircuit opgewekt.

Bij een lijnfrequentie van 31,25 kHz, zoals wordt gebruikt in een 100Hz-TV-ontvanger (26 inch beeldbuis), is het totale in de horizontale spoel gedissipeerde vermogen teruggebracht van 15,2 watt (in het geval van de zaagtandvormige afbuigstroom) tot 8.75 watt voor de sinusvormige afbuigstroorn. De spanning over de horizontale afbuigspoel is teruggebracht van ca. 2500 volt piek­piek tot 700 volt piek-piek met gebruik van dezelfde zelfinduktie als in de conventionele ontvanger bij 15,625 kHz. Ook de door het horizontale circuit veroorzaakte straling is aanzienlijk vermin­derd. De scheiding van de opwekking van de hoge spanning voor de beeld­buis en de opwekking van de horizontale afbuigstroom heeft als voordeel dat beide onafhankelijk van elkaar kunnen worden geopti­maliseerd (b.v. verlaging van de inwendige weerstand van de hoog­spanningsbron). Voordat een volledige vergelijking, betreffende beeldkwaliteit en vermogensdissipatie, kan worden gemaakt, moet het functiemodel worden afgebouwd. Op dit moment blijkt echter al, dat het vermogen dat wordt gedissipeerd in een ontvanger die een sinusvormige af­buigstroorn gebruikt, veel minder is dan het vermogen dat wordt ge­dissipeerd in een conventionele ontvanger, die een zaagtandvormige

-4-

afbuigstroom gebruikt, vooral bij lijnfrequenties van 31,25 kHz en hoger. Deze vermindering in vermogensdissipatie, die ook veroorzaakt wordt door een vermindering van het aantal grootsignaal-circuits, gaat ten koste van extra kleinsignaal-circuits. Door monolithische integratie kunnen de afmetingen en kosten van deze circuits worden gereduceerd.

-5-

ABSTRACT

The scanning raster 1n most TV-sets today is produced by means of sawtooth-current wavefarms in both the vertical and horizon­tal deflection coils. For the horizontal deflection circuit this kind of wavefarm has the following disadvantages: - High power dissipation; - high voltages over the components in the horizontal deflection

circuit; - radiation. These disadvantages will become even larger the line frequen­cy is increased in order to eliminate large-area flicker and in­terline flicker.

The breadboard model discribed here uses a sinusoidal horizontal deflection current generated with the aid of a series resonance circuit. Since both the rising and traili:ng parts of this current are used to scan the screen, the vertical deflection current wa­veform has to be changed into a staircase current wavefarm in or­der to get a homogeneaus scanning pattern, in which the scanparts of the A- and B-field do not intersect. The scanning in opposite directions involves the reversal of the video information in eve­ry other line. The reversing operation and the alignment of the lines scanned in opposite directions, has been realized digital­ly, at small-signal level. The correction of the pin-cushion distartion has also been realized at small-signal level, by sto­ring the samples of the amplitude modulated signal, used to drive the amplifier end-stage of the horizontal deflection current, in an EPROM. The high vol for the picture tube is realized 1n dependently from the horizontal deflection circuit.

At a line frequency of 31.25 kHz, as used in a 100Hz TV-set (26 inch picture tube) the total power dissipated in the horizon­tal deflection yoke has been reduced from 15.2 watt (in case of the sawtooth current waveform) to 8.75 watt for the si:nusoidal deflection current. The voltage over the horizontal deflection yoke has been reduced from ca. 2500 volt peak-to-peak to 700 volt peak-to-peak with the same self inductance (1.35 mH) as at the 15.625 kHz line frequency in standard TV-sets. The radiation caused by the horizontal deflection circuit has also been reduced. The separation of the high voltage souree and the horizontal de­flection circuit makes it possible to optimize them separtely (e.g. reduction of the internal resistance of the high voltage source).

Before a full comparison concerning picture quality and power dis­sipation can be made, the breadboard model has to be completed. Already at this moment, however, it appears that the power dissi­pated in a TV-set which uses a sinusoidal horizontal deflection current is for less than the power dissipated in a conventional TV-set, which uses a sawtooth current waveform, especially at line frequencies of 31.25 kHz and higher.

This reduction in power dissipation which is also caused by a re­duction in large signal circuitry, will be at the expense of ex­tra small signal circuitry. Monolithic integration is a possibi­lity for reducing this circuitry in size and cost.

-7-

LIST OF SYMBOLS AND ABBREVIATIONS

ff field frequency;

fh frequency of the horizontal deflection current;

f 1 line frequency;

h viewing screen height;

~h horizontal deflection current;

~ souree current for the parallel resonance circuit generating s

the horizontal deflection current;

~ vertical deflection current; V

k

s

s s

the sealing factor with which the active time (T ) a

is reduced, the video signals compressed, and the spot

velocity (v) increased;

humer of windings of the horizontal deflection coil; 6 Ta . = ~ , fract~on of the line time (T

1) carrying video

1 information;

coordinate of the spot along the path on the screen;

souree signal for the resonance circuit, generating the

horizontal deflection current (s is v or i ); s s s t time variable;

v spot velocity;

vh voltage over the horizontal deflection coil;

v souree voltage for the series resonance circuit generating s

the horizontal deflection circuit;

v voltage over the vertical deflection coil V

w v~ew~ng screen width

A blanking signal in TV-sets which have ff = 50 Hz;

Ah amplitude of the actual sinusoidal horizontal deflection

current;

A normalised amplitude of the sinusoidal deflection current; n

A amplitude of the sinusoÏdal souree signal (s) driving the s s

resonance circuit;

B "Blue" video information signal for ff 50 Hz or ff 100 Hz;

-8-

B re-arranged vers~on of B used for scanning with a symmetrie sym

horizontal deflection current;

C normalisation constant for the horizontal deflection current

(ih);

c a

capacitor ~n the resonance circuit tuned to Lh such that I

w =---h

LhCa

G "green" video information signal for ff = 50 Hz or ff = 100 Hz;

G re-arranged version of G used for scanning with a symmetrie sym

horizontal deflection current;

H line synchronisation signal for f 1 = 15.625 kHz;

Lh self-inductance of the horizontal deflection coil;

1 self-inductance of the vertical deflection coil; V

"Red" video information signal for ff = 50 Hz or ff

resistance of the horizontal deflection coil;

R magnetic resistance; m

100 Hz;

R re-arranged version of R used for scanning with a symmetrie sym

horizontal deflection coil;

RD distance between the centre of the deflection system and the

screen;

R screen radius; s

T time period during which the spot ~s actually visible on the a

screen (active time);

Tf period of one field;

Tfb time period during which fly-back takes place;

Th period of the horizontal deflection current,in case of the

conventional system Th= T1

, in case of the symmetrie de­

flection system described here Th= 2T1 ;

T1

period of one line;

V field synchronisation signal for ff = 50 Hz;

2A blanking signal in TV-sets which have ff 100 Hz;

2H line synchronisation signal for f1

= 31.25 kHz;

2V field synchronisation signal for ff = 100 Hz

al

e

p

T

-9-

phase difference between the r1s1ng edge of the ST

signal and the following zero-crossing of the s1nus-di

oidal deflection current 1h' such that ~ 1S positive; dt

phase difference between the souree signal s and the s

horizontal deflection current ih; 6 I . = ~ RS' normal1sed screen curvature;

ê vt/~, norrnalised time variable (or normalised dis­

tance) used for calculating the desired horizontal de

flection current;

Wh angular velocity of the sinusoidal horizontal deflec-

tion current;

norrnalised angular velocity of the sinusoidal horizontal

deflection current;

line duet a sequence of one LR line and one RL line including the

overscan parts;

LR line the piece of the video signal (R G B ) which syrn' syrn' syrn

will be used to reproduce the picture with the reprodu-

C1ng spot rnov1ng frorn Left to !ight on the screen;

RL line the piece of the video signal (R G B ) which syrn' syrn' syrn

will be used to reproduce the picture with the repro-

ducing spot rnaving frorn !ight to Left on the screen;

ST signal the logic signal at half the line frequency deciding

whether the line of the i~~2~igg video signal will be­

corne an LR line or an RL line; if ST is "low", then the

line of the incorning video signal will becorne an LR line;

if ST is "low", it will becorne an RL line.

-10-

I. INTRODUCTION

In recent years, the quality of the ?icture displayed by consu­

mer, television receivers has been improved to the extent that

further improvP.ments are difficult to achieve within the frame-urk

of existing display techniques. Additional impravement could be

obtained through improvements in colour picture tubes, better sig­

nal processing, and through modification of current video-signal

transmission standards, i.e. NTSC, PAL and SECAH:[1.2]. Howevermo­

dification of these standards may not be acceptable unless the

changes are compatible with the circuitry in the receivers alrea­

dy installed in millions of homes and do not make these receivers

obsolete. Alternative approaches are being explored that do not re­

quire a change in transmission standards, but provide an imprave­

ment in observed picture quality through the elimination of cer­

tain artifacts in the picture; i.a. large-area flicker and inter­

line flicker. Lare area flicker will disappear completely when

the number of fields per second is dou;.;led; i.e. from 50 to I 00 Hz

for the European systems· (PAL and SECAM), and from 60 to 120 Hz

for the A~erican system (NTSC).

For the PAL system this has already been worked out by

Kraus [3]. In fact, the system which will be considered bere, is to

be used 1n combination with a 100 Hz colour TV-set according

to [3]. The increase in field frequency from 50 to 100Hz

implies an increase in line frequency from 15.625 kHz to 31.25 kHz.

A technique under evaluation for the NTSC system is the impravement

of picture quality by a change in the horizontal scanning system

from the current 525 line interlaced system to a 525 line non-1n­

terlaced system (progressive-scan) system. This would involve an

increase in the line frequency from 15.75 kHz to 31.5 kHz.

Bath quality-improvement techniques described here need a

doubling of the line frequency, but do not require any change 111

the transmitted signal. Note that a combination of both techniques

for the NTSC system would involve an increase in line frequency

-11-

from 15. 7 5 kHz to 63 kHz, and for the PAL sys tem would invo 1 ve an

increase in line frequency from 15.625 kllz to 62.5 kHz, respecti­

vely. The standard horizontal deflection circuit used in televi-

sion-receivers has the following disadvantages:

High power consumption;

high voltage over the deflection coil and other elements ~n the

circuit;

- radiation problems.

These disadvantages become even worse if the line frequency ~s ~n­

creased [ 4l, as will be the case with the impler.1entation of the

described improvements. The disadvantages arise from the big chan­

ges in the first derivative of the deflection current due to the

short fly-back time [ 4}.

A horizontal deflection system, in which both the r~s~ng and trai­

ling part of a syrnmetr.ic deflection current, e.g. a triangular or

sinusoidal current, are used to scan the screen horizontally, can

be used as an alternative. The frequency of such a symmetrie cur­

rent ~s half the line frequency. Such a deflection syste~ will not

have the disadvantages of the conventional deflection system but

there is now a new problem:

The three video signals (R, G and B), userl to drive a colour catho­

de~ray tube, have to be turnen around every other line with shi~t

registers, and at the same time the lines "written" on the screen

from left to right have to be aligned with the lines "written" on

the screen from right to left.

This alignment problem can be solved fundamentally by determining

the starting moment of the read-operation using a direct measure­

ment of the deflection current and its first derivative.

In this report a horizontal deflection system ~s considered, which

uses a sinusoidal deflection current. SDecial attention has to be

payed to:

- The power dissipation;

- the exactness of the control circuitry;

~12-

the possibility of monolithic intergration of the total deflec­

tion circuit.

In chapter 1 a conventional colour TV system and its standard ho­

rizontal deflection system at the receiver is reviewed. The disad­

vantages of this system as a function of the line frequency are

considered, and the system is compared with the alternative system,

which uses a symmetrical deflection current. The theoretical foun­

dations for a breadboard model of the alternative horizontal deflec­

tion system are treated in chapter 2. Chapter 3 gives the hardware

re al ion of this model. In chapter 4 the balance will be drawn

up, to see in what way the theoretical aspects, concerning the

breadboard model, correspond to the practical results. The exact­

ness of the control circuitry will be considered, and the power

d sipation 1n the horizontal deflection circuit, based on a s1nus­

oÏdal current, will be compared with the power dissipation in the

conventional horizontal deflection circuit. Finally we will consi­

der roughly the possibility of monolithic integration of the total

deflection circuit.

1.1 The horizontal deflection circuit in the conventional colour

TV receiver

The problem of television (the transmission and the reproduetion

of moving scenes) is solved by subdividing the image field into a

sufficiently high number of picture elements and by periodically

transmitting electrical signals representing the light intensities

of the picture elements 1n a sequential way. The sequential elec­

trical signals are generated by a raster scan of the image field

by means of a light sensitive probe (exploring element) of

which the aperture determines the size of the picture elements

(see figure 1.1). The reproduetion of the 1mage is realised by

moving a intensity-modulated small light souree (reproducing spot)

1n a raster scan over the display screen. Measures must be taken

to synchronize the recording and reproducing rasters. The probe,which

produces a voltage or current proportional to the light intensity,

starts at point A1

and moves with constant but unequal rates in the horizontal and vertical directions, following the path A1 A2.

-13-

h

Fig. 1.1, [sJ: Scanning raster Cline spacing grossly exaggerated). Solid lines are the A field; dashed lines are the B field.

Upon reaching point A2 , the probe quickly flies back to A3

(the

horizontal fly-back) and proceeds similarly to point A4 . Then

the probe flies back vertically to B1

and follows an interlaced

pattern ending at B2.

The process is then repeated starting aga1n at A1 • The two sets

of lines are called the A and B fields; tagether they constitute

one complete picture or frame. The frame rate 1s just rapid

enough (25 per second) to create the illusion of continuous mo­

tion, while the field rate (twice the frame rate) brings the

large-area flicker effect tri a still annoving but in general

tolerable level, the field sequence being ABABAB .... The com­

plete system for this type of television transmission and recep­

tion can be represented by the five basic elements shown in

figure 1.2. Communication

I channel I

1--- -f--- - f- -1--- --, Scanning >- -

-t._ patterns~ -- -~ -.._, ·'<

Exploring Reproducing spot element

Image field Viewing screen

Figure 1.2, [6]: Functional Representation of a black-and-white TV system.

-14-

.These basic elements are: (I) Exploring element, (2) Scanning

raster on the image field. (3) Communication channel. (4) Repro­

ducing spot. (5) Scanning raster on the v~ew~ng screen.

The system shown in figure 1.2 can only serve as a modelfora

black-and-white TV system. Figure 1.3 suggests how the system

might be generalised in case of a colour TV system.

Figure 1.3 depiets the fact that the colour sensation of nearly

any speetral light distribution within the visible range can

be produced by means of three monochromatic sourees (i.e, red,

green, and blue) which stimulate the three different light­

sensitive recepters in the eye to the same extent as the origi­

nal distribution. In the ~mage piek-up camera the original ~ma­

ge is divided into three color components. These separated ~ma­

ges are explored synchronously in the same way as the single

image, in the black-and-white system. Three signals (R, G and

~) are sent simultaneously to the receiver.

'1J

Red mq fiek:! R G channel ~

B

--LJ -

Green ·lra:Je fetj Vil?';ciirq screen

,_

- -o -

Blue ir1ldge fiekl

Fig. 1.3: Functional representation of a colour TV system.

"J i-'•

()Q . -. ..". ..

(/) <: 1-'• 1-'•

()Q 0.. ::l (I) Pol 0 1-' (/)

0" H'l 1-' 0 Pol 'i ::l

~ Vl 1-'• 0 ::l

()Q ::r: N Pol

Pol .,5. ::l 0.. 1-'

J-1.• ::l

0 (I) 0

Pol ::r: ::l N 0..

H H'l <: 1-'•

(l) 'i 1-' (l) 0.. (') (l) (/)

1-'• '< < ::l (I) (') 'i ::r' (/) 'i

0 ::l 1-'• (/)

Pol ("1"

1-'• 0 ::l

R, G Of B 50 or 100Hz

composi te bldnking A!SOJ Of 2AI100Hz} n

A field

I -------.!: ~ 1 I

fiE>Id-sync: V Of 2V

I B field :

B field

A field

R,G orB I

comp blanking I '-----:---------------------------J

I ine-synr

fi<?ld-sync

) rïr1r1r1r1r1r.-,rïrïr-lrïr-lr-lr-lr-lr-lr-lr-lr-lr-rlr-lr-lrïr-lrïr-lrïr-lr-lr

I

ll-~~----l

l=Tt = 64~s a=12~

5"0 Hz b cc5,51JS C= 1,51JS d= SIJS I!= fiJS

{

L= Tt = 321JS a= 61JS

100Hz b=2,15~J. s c~o?51JS d= 2 S~s e= l51JS

fig. 1.4a

b

(

d

e

f

g

h

Vl I

-16-

In the rece~ver, as shown in figure 1.3, there are three collo­

cated spots which produce the three images.

At a distance of 5 to 6 times the viewing screen height, these

three images are amalgamated by the eye, thus bringing back the

sensation of the original colour picture.

Blanking pulses are inserted before the video signals are sent

over the channel. They are inserted during the fly-back intervals

to blank out the fly-back lines at the receiving tube. The added

synchronisation signals are used to synchronize the horizontal

and vertical scanning mechanism at transmitter and receiver.

Figure 1.4 shows an example of one of the video signals (R, Gor

B), the blanking signal, and the line and field synchronisation

signals as a function of time for 50 and 100 Hz TV sets. The sig­

nals used in this 100 Hz TV-set have the same waveferm as the

signals in the conventional 50 Hz TV-set, but they are compressed

in time.

The reproducing spot on the viewing screen (see figure 1.2)

formed by cathode-ray beams deflected over the display screen by

means of a suitable magnetic deflection system. It is the current

through the horizontal deflection coil which plays a major rêle

~n this report.

If the spot size ~s constant and we want the same resolution over

the whole picture, then the spot velocity of the exploring spot

(as well as of the reproducing spot) has to be constant, too.

In the case when the spot velocity is a linear function of the

deflection current, the deflection currents corresponding to the

scanning raster shown in figure 1.1 would have sawtooth wavefarms

for both vertical and horizontal direction. In practice only the

deflection angle is approximately a linear function of the cur­

rent through its deflection coil. The spot velocity, however,

is not a linear function of the deflection angle. This can be

shown easily for a flat screen. A constant angular velocity

~17-

will nat give a constant spot velocity s~nce the spot will move

faster at the extremes of the screen than in the middle (see

figure 1.5). Thus the linear deflection current must be changed

óS

óS ) 65 e m

centre of deflection

fig. 1.5: Spot displacement on a flat screen in case of a con­

stant angular velocity.

into an S-shaped current in order to obtain a constant spot velo­

city over the whole screen. This is especially felt with respect

to the current through the horizontal coil, since the width (w)

of the picture is greater than its height (h). For standard TV,

w/h = 4/3 (see figure 1.1). In its simplest embodiment the hori­

zontal de.flection system of a conventional television receiver

consists of a magnetic deflection coil, a power souree and one

or more switches [7]. Figure 1.6 shows such a deflection system,

while figure 1.7 showsits associated current and voltage wave­

farms. The switch in this case ~s a combination of a transistor

and a diode (damper). The circuit includes a fly-back capacitor

Cfb' which, tagether with the deflection coil Lh and the fly-back

coil, determines the length of the fly-back interval.

It also includes the capacitor C , which provides the duel func-s

tion of DC blocking and S shaping. An actual deflection circuit

used ~n a receiver would norrnally also include pin-cushion and

linearity correction circuitry plus a winding on the fly-back

coil to provide a scurce of high voltage for the picture tube.

The disadvantages of the described conventional horizontal de­

flection circuit are:

- High power consumption;

high voltage over the deflection coil and other elements ~n the circuit;

- radiation problems.

Ie

HOR. OUTPUT

-18-

Eoc

11 11 ISO LA Tl ON l 11 (Fl YBACK) 11

10 I (fb

DAMPER

(fb ~~OR 1~ YOKE

.. Lh Cs

Fig. 1.6, [4]: Basic horizontal deflection circuit 1n the con­

ventional receiver.

Fig. 1.7 [4]: Current and voltage wavefarms in the

horizontal deflection circuit.

-19-

1.2 Power dissipation, high voltage and radiation problems in the

standard horizontal deflection circuit at different line fre-

quenc~es

In the circuit of figure 1.6 there are 3 eleoents which dissi~ate

power, Le.: (I) The deflection yoke; (2) the transistor; (3) the

damper.

Power dissipation ~n the deflection yoke results from 3 factors,

[ 4] : (a) r 2

R losses in the windings;

(b) eddy current losses in the windings;

(c) hysteresis and eddy current losses ~n the ferrite yoke r~ng.

The power dissipated in the transistor ~s the sum of the power dis­

sipation due to collector current during scan, the dissipation due

to the base drive, and the dissipation during fly-back.

W.E. Babcock and W.F. hledam [4] have detemined the losses ~n the

standard deflection circuit (shown in figure 1.6), at different

line frequencies, under the following conditions:

(a) The DC voltage is the same for all line frequencies;

(b) the peak stared energy in the yoke is constant for all line

frequencies; ~ 4,7 milijoules;

(c) the peak fly-back voltage is about the same (~ 900 volt) at

all line frequencies;

(d) the ratio of scan time and fly-back time ~s constant for all

line frequencies.

The fly-hack voltage is limited by reducing the number of windings

(nh) of the horizontal coil (thereby reducing the self-inductance

Lh; Lh::n~) and at the sametime increasing the current ih through

the coil with the same factor.

Tables I. I a, band c show the horizontal deflection yoke losses

for three ferrite ring materials tested at 15.75, 31.5 and 63kHz.

Figure 1.8 shows theselossesas a function of frequency.

line ency

-20-

Table 1.1, [~: Power losses 1n the horizontal deflection yoke

at d ferent line frequencies

table 1.1a:

LOSSES USING DEFLECTION YOKE lHTH RCA 804 FERRITE

122 total eddy eddy current

frequ- temp rise retrace P-P retrace tot al yoke yoke current & & hysteresis above amb. time voltage loss loss hysteresis loss during

loss retrace

15.75 kHz 19° c 12.3 psec 880 7.6 w 3. 95 1l 3.65 w 3.65 w 31.5 kHz 33° c 6. I psec 875 15.2 H 4.53 w 10.67 1\f 8.19 w 63 kHz 82° c 3.1 psec 865 52 1-1 5.5 w 46.5 w 38.4 w

table l,lb:

LOSSES USING DEFLECTION YOKE WITH PERRITE "A"

total eddy line temp rise retrace P-P retrace total yoke yoke 1

22 current & hys-frequency above amb. time voltage loss loss teresis loss

15.75 kHz 19.4° c 12.2 psec 920 6.7 w 4.05 w 2.55 w 11.5 kHz 14.4° c 6,08 psec 910 13.6 w 4.44 w 9.16 w 61 kHzlt --- --- --- --- ---

ltUnable to run at 63 kHz because core temperature exceeded curie temperature

table 1. Ie:

LOSSES USING DEFLECTION YOKE W!TH FERRITE "B"

line temn rise retrace P-P retrace total yoke yoke !22 total eddy frequency above amb. time voltage loss loss current & hys-

teresis loss

!5.75 kHz 17° c 12.2 psec 920 7.4 j.,r 4.1 w 3.3 w 31.5 kHz 36.6° c 6.05 usec 910 20 w 4.5 w 15.5 w 61 kHz 60° c 3.03 usec 880 41.5 j.] 5.44 w 36.06W

The line frequenc 15.75, 31.5 and 63kHz are related to the

NTSC system. Note that the corresponding line frequenc of the

PAL system (15.625, 31.25 and 62.5 kHz respectively) are approxi­

mately the same.

The curves 1n figure 1.8 show that at the standard 15.75 kHz line

frequency the combined hysteresis and eddy current losses are

about equal to the losses in the windings. As the line fre-

quency is increased and the fly-back time is shortened, core losses

and eddy current losses in the copper become a large portion of

the total loss.

lOOr-----------------------~~~~~ Renace T ~ ~ }'

1';<: ·• Scan R.ate

u.n kHz

ll. S kHZ

12. l 118 1040 "'" ~ ;~;,;.:..: - .

6.1 "'' 26o "'"~Ti: :~~ ·~

f • 1.8a (RCA 804 Ferrite)

• IJ ... !1 I

ll 0 ... ... :t ... 111 111 ...

Q

t ~ ....

' . ' i ' ; i l i

1 10 lS

- -- I -1--1..:.. -- ;__ - 1-

.. ,. : ; ' ~ ; . :I·· i . ::1 , .. .... ; •zt· I ., ..

10 50 llO

Scanning R.ate - kllz

• 1 • 8b (Ferrite Haterial "A")

• ... ... !1

Cl 0 ... ... a. ... • • ...

Q

100 Scan R.He Rctrace T Yoke J:: •·····~·r-

----- ·-·- ....... t'l. n klh 12. 2

~· 1068 ~~~~~ 1-- f-·· f-

)1.) kHz 6.0) ~8 26) ~H ;- ·-·· -I- -

6] !<11; ].0) 6). 1 --· -~- - +-~-

~8 ~~~ 0 ... I·

l

1·, -- ---~ ·-· ... I· .

.. ~1- --·-•·-'----'--- -- --- -· --- .v l

0 vvoq V u - .......... : • V I;' I.;._;_.;. -;-:-- -- .,"' v V" . ..,. ... . ., ..

0 .. -.?vL .. "' . :. ' . ... ~ 1/t--~ o"- -- -~- ....

i I ' . ·y .-v ' . --~ .".

··----- --- -;: V ... "'I;' ~ . ·- i·· .

. . t/V ~~V~~ ' l :

• I J 41 •' 7 .... - 1-.

"'. . . I I f-., . . . f o" . ·--...

2

10

-. I .. :; : cl .--v ; ... ~7- ~--. f-":' l--\"

et- f-·· ·--. (.0~ ~ .. .

\1'-\..O,ses

I---'-- -~ ~- -- -· 1-- ·-:

) 1-1- f-f-

. ' ---- --- --· ~-· ~-- 1-- . . - ~- .. . I-. . .

2 --- --1-;-

' __ .........,. __ ... -· ~--- -- -- -- .,. . ·~-- ~-

l . .. ,.

I . I. i .. 1

10 l5 lJ 1 0

Scannlng R<He - IUiz

fig. 1.9c (Ferrite t·1aterial "B")

. 1.8, [4]: Deflection yoke lossesas a function of line frequency (horizontal scanning rate)

I N

l 2 3

-22-

5 lO 20

Figure 1.9 [4J: Deflection yoke lossesas a function of

fly-back (returne) time

Figure 1.9 shows a tynical curve of yoke dissi~ation as a function

of fly-back (retrace) time. At the very short fly-back time used

for the higher scanning rates, the losses increase rapidly. In the

conventional system not only the power dissi~ation in the yoke in­

creases as frequency increases, but the nower dissipation in the

transistor and the diode of the circuit shown in fi~ure 1.6 will

also increase as frequency increases. These power losses have als0

-23-

been determined by Babcock and Hedam [ 4]. Table I. 2 glves the

result of these calculations.

Table 1.2, [~: Horizontal output transistor and

damper dissipation at different

line frequencies.

scannlng ra te 15.75 kHz 31.5 kHz 63 kHz

retrace time 12 ~sec 6 ~sec 3 ~sec (T ) r peak retrace 990 990 990 '10 1 ts (V )

s DC supply 128 123 128 volts (EDC)

Peak switch 3 A.rnn. 6 A.n:m 12 A.rn'. current

Peak base 1.2 Ar.!~. 2.4 Amp. 4.3 Amp. current

Transistor I. 4-4. 3H 7. 6-28. 7H 35-165\\T d. . . :t) lSSlpatlOn

Damper 0.4 ~v 0.83 H 1.8 ~v

dissiuation -· ------

:t)Lower value is for fall time Tf of 0.4 ~icroseconds. Higher value is for fall time of I microsecond.

The peak voltage across the deflection yoke lS glven by [4]:

where

V h,peak

fly-back time.

(I. I)

The peak voltage on the deflection switch and fly-back capacitor

is:

V S,peak

V C,peak

V S,neak

-24-

TIL I h h,peak

Tfb + EDC (I • 2)

As shown by equations (1.1) and (1.2) tQe peak voltage on the yoke

and on the deflection s1vitch is inversely pro~ortional to the fly­

back time.

Therefore, for a given deflection yoke, if the line frequency is

doubled and the fly-back time is cut in half, the peak yoke volta­

~e will double and the peak switch voltage will nearly double. If

these comnonents cannot withstand this high voltage or do not have

adequate safety factor, then yoke inductance must be decreased and

peak current must be increased to keep the peak voltage within the

desired limit and stil have sufficient scan. That is what Babcock

and Nedam did 1n their calculations and measurements: they kept the

fly-back voltage constant at all line frequencies. In this way the

high fly-back voltage contribute indirectly to the power losses in

the horizontal deflection circuit at the higher line frequencies.

It is obvious that, with increasing line frequency and hence decrea­

sing fly-back time, the radiation will increase, since radiation is

mainly caused by the high frequency components due to the short fly­

back time.

1.3 Symmetrie horizontal de~lection currents

If one of the symmetrie currents, shown in figure 1.10, is used,

then the disadvantages of the horizontal deflection system will get

smaller. During scan time the wavefarm of such a s~etric deflec­

tion current should approximate the wavefaro of the conventional

asymmetrie deflection current, in order to get a constant snot ve­

locity on the screen. Figure 1.10 shows two symmetrie current wave­

farms. Figure I.IOa shows a triangular form, figure !.lOb shows a

sinusoidal form. The parts of the rising-part of the wavefarms in­

dicated with "LR-scan" are approximately the same as the scan part

0

-25-

\ LR- SCël'l \ RL-scan over-scan over-scan

~ ~ / ~

t

..... 1,..... -\ LR-scan \ RL-scan

over-scan over-scan

Tl

Th

Fig. 1.10: Two symmetrie current wavefarms suitable for

horizontal deflection: A trangular wavefarm (a)

and a sinusoÏdal wavefarm (b).

of the wavefarm of the conventional horizontal deflection current.

If we want to use the trailing parts of these waveforms, too, then

the video signals displayed in the corresponding time interval have

to be turned around in time.

-26-

If we maintain the vertical deflection current used in the conven­

tional receiver, then the scanning pattern on the viewing screen

changes from the one shown in figure 1.2 to the one shown in figure

1.11 in which the viewing screen is alternately scanned from left

to right and from right to left.

w

h ~

v

Fig. 1.11: Scanning raster produced by a conventional verti­

cal deflection current and a symmetrie horizontal

deflection current.

The lines scanned from left to right and the lines scanned fro~

right to left should be aligned, that is to say: A straight vert

cal line in the original picture should stay straight in the pic­

ture displayed on the viewing screen. If the nlace of the spot on

the screen is determined unambiguously by the current through the

deflection coil, then the rising and trailing parts of the deflec­

tion current have to be absolutely symmetrie in order to get this

alignment. Note that with the conventional asymmetrie horizontal

deflection current this alignment is accom~lished much easier, be­

cause small deviations fro~ the ideal current wavefarm will not

give disturbing geometrical deformation, since these deviations are

likely to be the same for all lines.

It is obvious that from the two symmetrie currents shown in figure

1.10 the sinusoidal current is preferable, because its first deri-

-27-

vative does not contain any JUmps at all, the rising and trai­

ling parts of the sinusoidal current wavefarm are S-shaped just

like the conventional deflection current waveform, and, moreover,

a ser1es or parallel resonance circuit using an additional capaci­

tor can be created such that the horizontal deflection amplifier

just has to deliver effective power. That is why the description

of a breadboard model using a sinusoidal horizontal deflection

current will be the main subject of this report.

-28-

2, THEORY

In par. 1.1 we assumed the deflection angle to be a linear function

of the deflection current. For the çlat screen shown in figure 1.5

we then demonstrated that the snot velocity is not a linear func­

tion of the deflection angle, but that the deflection current

should have an S-shaned wavefo~ in order to achieve a constant

snot velocity on the screen.

Consicier a cathode-ray tube of a colour TV-receiver (figure 2.1).

x

--z 0

xz-plane

Fig. 2.1: Picture tube ~n the rece~ver

The tube has two planes of s:~etry. Let the z-ax~s of a Carte­

sian coordinate system (x,y,z) be the intersection of these two

planes, the xz-nlane and yz-plane being as indicated in figure

2. I. Let the xy-plane touch the tube in point 0 the origin of the

system of axes.

We will determine the horizontal d~flecrion current ~n the ulane

of intersection of the screen with the xz-plane. We might also

determine the horizontal deflection current for an arbitrary

plane narallel to the xz-plane, but this appears to be nuch

more difficult. Xoreover, the calculated deflection current ~s

used for horizontal scanning for all lines in a field. Although

-29-

this will g1ve pin-cushion distartion of the picture 10 East-West

direction, as shown in :igure 2.13, this distartion may be correc­

ted by amlJlitude modulatinf, the given horizontal de:!:lection cur­

rent with a signal of :ield frequency l~ . In par. 2.1 the exact current wave~orm necessary fora constant

spot velocity, will be determined. This theore.tically and numme­

rically determined S-shaped wavefarm will be compared with the

correspondin3 S-shaped part of the sinusoidal deflection current.

The part of the current of interest for this cornparison is called

the "active" part of the deflection current, corresponding to 'Ji­

sible spot positions on the screen. If the active parts of the

theoretically deterrnined current waveforn and the sinusoidal cur­

rent wavefarm used in nractice do nat coincide, then a geometrie

horizontal distartion o~ the ~icture will result. The magnitude

of this distartion appears to be rather small for the ,icture

tube used bere (A66-540X); this distartion can be minimised

further by signal processing, as we shall see in nar. 2.1. That

opens the way to a horizontal deflection system with a nerfect­

ly sinusoidal deflection current. Thi~ sinusoidal current will

be generated by a resonance circuit. In ~ar. 2.2 the two prin­

ciual types of resonance circuit will be reviewed, and the am­

plitude rnodulating signal of field frequency, necessary for cor­

rection of the pin-cushion distortion, will be considered quali­

tatively. The influence of the Q-factor of this circuit will be

discussed in par. 2.3.

The lines "written" on the screen from left to right will nat be

aligned with the lines "written" on the screen frorn right to

left. ~o solve these alignrnent problerns, treated 1n ryar. 2.4, we

will only consider the active narts of the video signals; so the

blanking partsof the lines (see figure 1.4) are excluded.

By shifting the active parts of the video signal such, that they

coincide, in time, with the active uarts of the sinusoidal cur­

rent, the alignrnent problems can be solved. Finally in nar. 2.5,

the consequences, which the implementation of a sinusoidal hori­

zontal deflection current has on the scanning raster, will be con­

sidered.

-30-

2.1 The effects of sinusciclal horizontal deflection currents on

the geometry of the picture

2.1.1. Introduetion

The desired deflection current wavefarm g~v~ng a constant spot ve­

locity denencis on the place of the observer. An observer ~t a dis­

tanee of 5 to 6 times the viewing screen hei~ht (h) is at the opti­

mal position for looking at a screen [9]. However, it is very diffi­

cult to determine the current in this case. Instead, the current

will be determined in two other c~ses: (I) ~Vhen the spot velocity ~s

constant on the curvature of the screen iffielf; (2) when the spot ve­

locity is constant for an infinitely removed observer.

These two currents provide lower and u~ner bounds on the desired

current giving a geometrically correct picture for an observer at a

standard distance of 5 to 6 times the viewing screen height. For

the cathode-ray tubes used ~n modern TV-sets these two bounds will

appear to be so very close to each other, that the desired current

is determined for all practical purposes.

The vacuum inside the cathode-ray tube is very high. To limit the

resulting farces the screen has been curved. The intersectien of

the screen with the xz-plane is a niece of a circle at least for

the nieture tube considered here (A66-540X, see figure 2.2).

x l

xz-plane

Fig. 2.2: Intersectien of the picture tube (A66-540X) with the horizontal plane (xz).

-31-

The instantaneous horizontal deflection current ih 1

will de-, flect the electron-beam over an angle ~D

1, and the snot will ,

move from s0 ( ih = 0, t = 0) to S 1 • Seen from the screen the electron

bearn seerns to come from the ~oint ~~ on the z-axis known as the

centre of the deflection system. Let th~ distance from f~ to the

point s0 on the screen be ~· and let the screen have radius R8 and centre M

8.

Then we want to know the deflection current ih as a function of

~· R8 and t, giving a geometrically correct picture.

The current direction is assumed to be chosen such, that 1h 1s

positive if ~D is positive, and we then will only consider ~osi­

tive values of ih. The negative values of 1h can then be deter­

mined easily, since ih is an odd function 1n time t.

The phase and the frequency of a sinusciclal deflection current

are already determined, because the sinusciclal current has to be

0 for t = 0, with positive first derivative, and the frequency

is half the line frequency. Thus the sinusciclal current has to be

max1mum at t = ~ T1

, where T1

1s the oeriod of I line. The only

thing which can be changed 1s the arnnlitude of the sinusciclal cur­

rent.

For a given spot velocity v, and given ~ and R8 , the horizontal

deflection current giving a geometrically correct picture is

completely detemined. Now let T be the time interval corresponding a with the active 9art of the deflection current and let w be the x width of the picture measured along the x-axis. Then for a given

picture tube the spot velocity v is determined such that during

T seconds the distance w is covered. So the deflection current, a x

which is theoretically determined by this spot velocity (and by

the given Ru and R8), will cause the spot to reach the extreme

right of the picture at t = !Ta. We now have to choose the am­

plitude of the sinusciclal current such, that 1n the interval

0 < t < lT the sinusciclal current resembles this theoretically 2 a

deterrnined current as closely as possible. Here the amplitude of

the sînusoidal current will be chosen such that the ~agnitudes

of the sinusciclal current and the theoretically determined cur­

rent are the same at t = ~T • a

-32-

Note that in this case the interval lT 2 a < t < !T1 will be used

for overscan.

Normally the spot velocity, which 1s determined by T , cannot be a

changed, because T has to be the same at transmitter and rece a

ver. However, in the breadboard model described here, the video

signals are A/D converted and stared into memory, so they can be

compressed or expanded in time easily, by changing the reacl-out

clock frequency of the memory. By campressing or expanding the

video signals, in fact (at the receiver) is changed. With

each new T a new spot velocity v is determined, and with each a

new snot velocity a new deflection current, giving a geometr

ly correct picture, is determined. This current 1n turn, will

determine a new amplitude for the ap?roximately sinusoirlal cur­

rent. The distartion which is found with this improved T might a be less than the distartion which has been found with the stan-

dard

Note that for the case when T is decreased (compressing of the a video signals whereby the spot velocity is increased), a smal-

ler, more linear, part of the sinusoidal current is used; con­

versely for the case when T is increased (expanding of the video a

signals whereby the spot velocity is decreased), a larger, more

curved, part of the sinusoirlal current used.

In this way, by changing T , and thus changing the snot velocity, a

the geometrie horizontal distartion mi~ht be minimised further.

2.1.2. ~~~~~~-~~-!~~-~~f!~~!!~~-~~g!~-~D

a. Lower bound (for observer at the screen).

Consicier a screen with radius R8 , and a deflection system D with

radius RD. We are searching for the angle ~D as a function of ~·

Rs and t when the spot-velocity on the curvature of the screen 1s

assumed constant, see figure 2.3! In this figure the following

equation holds:

-33-

with 6s = v 6t, v being the spot veloci~y. We find:

V or

ar ~n the limit 6t ~ 0:

V

Let ~S be a function of ~D' then:

-- = dt

ar with (2.1):

V 1 • d~

s d~D

( 2. 1)

(2.2)

(2.2)

Equation (2.2) gives a relation between ~D' R8, t and ~S

d~D Ta determine -at as an explicit function of ~D' ~ and R

5, we

shall first derive a relation between ~S and ~D. d~D

Now let ~ and R8 be fixed, then with -at known as an explicit

function of ~D' we are able tq determine ~D as a function of t,

numerically. Suppose, for instance, that ~D(t) is known at t 1,

then ~D(t 1 + 6t) with 6t + 0, is determined by:

d<bnj ~D(t 1 +6t) ~ ~D(t 1 ) +-at t=tl .öt, (2,3)

for suitably small. Now take ~D(O) = 0, then with (2.3) ~D

can be determined arbitrarily well for every t.

-34-

,Fig. 2.3: Deflection geofletry for a spherical screen

In triangle MsMUs 1 the following equation holds:

(2.4)

In triangle MbS)S 1 the following equations hold:

-35-

or (2.5)

With (2.4) and (2.5) we find:

2 Omitting the subscript "I" and deviding by RS g1ves for an arbi-

trary point

Defining p

Defining

or

. 2'"' s1n '~'s

. 2'"' S1n '~'D

~2 ~ I + (I - --) - 2(1 - --)costjl

Rs Rs s

(',~ =- the

Rs last equation beco~es after squar1ng

. 2'"' s1n '~'s

{ 2 sin tjJD

2 - I - (1-p) }

2

. 2'"' (', . 2 (', ( 2) (', 2 s1n '~'D p, s1n <Ps q, and 1-p u ,

2 2 (3. - I - \! )

p

2 4 \) ( 1-q)'

2 2 2 2 2 2 2 q - 2q(p + \! p - 2\! p ) + (1-v ) p o.

The roots of the last equation are:

(2.6)

q:. = p[(I+p2) - 2}p:_ 2\!v }p2

+ (I+})p + ;] (2.7)

Using the definitions for p and q

sin2

rps = sin2

<t>D[(l-+U2)- 2U2 ~dn2rpD + 2uVu2

sin4

rpD-(I+U2)sin

2rpD+;]

(2.8)

t.Je shall restriet ourselves to configurations as shown 1n figure

2. 3, in which ~ ~ RS < co, or 0 < p ~ I •

-36-

If we do not consider the configuration with p = I, then it can be !::. shown, that the factor between brackets 1n (2.8), U = V ~ IV, 1s

always positive. Then, sirree sin~S must be positive if sin~D 1s

positive, only 2 of the 4 solutions for sin~S can represent the

configuration in figure 2.3, so:

sin~s = sin~D[(l+1})- 2u2 sin2~D + 2uVu2 sin4~D-l(l+u2 )sin2~D + 1]

(2.9)

Note that equation (2.9) also includes the solutions of sin~S for

P = I (u=O), s1nce sin~S = sin~D for p = I in the configuration of

figure 2.3.

Differentiation of (2.9) with respect to ~D g1ves:

d~s cos~s -d,t,

'~'D

I • ,t, 1 2 S1n'I'D

cos~D [U] 2

+

x d: [u J D

[up (2.10)

1T For symmetry reasans we may restriet ourselves to 0 ~ ~S < 2, so with (2.9) we find:

Now, wi~h (2.10), we find:

[I -sin2

n[u]J 2 ruJ 2

d~s~ with (2.11) we find two possible values for ---d~D ~ = 0

D

[v + wl !J<lb 0 I + u 2 - p

d~s [TIJ!\ I --- =

w] 2

1<lb d~D

·For 0 < p < I

configuration

(jJ =0 D [v - 0 I - u = p

d~s only the latter value will do, sirree --- < I,

d~D of figure 2.3.

1n the

-37-

So finally we find: l l • <P d

[ V-~<71 d<Ps co s<PD [ v-W] 2 2s~n D

d<j)D

d<j)D 2 -[I-sin <PD[v-wJ]

2

+

[I-sin2

<PD[V-ivJ] 2

[v-w] 2

with V~ (I+J2

) - 2v2

sin<j)D

w~ 2 v 2 . 4 c 2) . 2 v v s~n <P - !+V s~n <P D D

_d_[v-w] dcjJD

+ 1

(2.12)

\.!ith (2.2), (2.3) and (2.13) cjJD can be determined numerically as a

function of t.

Noting that if t

cjJD givin by:

dqJD

dt t=O

V l =

0 én cp5 = O, the first derivative of

V I v Rs V = --= --

Rs p Rs ~ ~ d~s~ dcjJD cjJD=O

~ vt will vary if v or RD ~s varied, the dimensconless quantity t

has been introduced.

Hith t we find:

dqJD

Hence, s~nce

dcjJD = dt

dtf!s

cjJ =0 D

dt -= dt

p,

V ~/v

..J?_ (2.13) . dij)= dcjls s

dcjJD dtf!D

Using t, all curves cpD can be olotted in the same figure with the

same resolution. t, may be considered a normalised time variable,

since t can be~egarded as: The ratio between the time variable t

and the time -- , which the spot, moving with a constant velocity V .

v, needs to cover the distance RD. t, however, may also be consi-

dered a normalized distance,since t can be regarcled as: The ratio

between the distance vt, which the spot when moving with a con-

-3~-

stant velocity v, has covered during ti~e t, and the distance ~·

Since vt = s (see figure 2.3), the normalised distance T is also

g1ven by:

T = vt s (2.14)

Usinz (2.12) and (2,13), ~D(T) can he calculated numerically 1n

the sa~e way as ~D(t), see (2.3), with:

(2.15)

Figure 2.4 shows the results of these calculations for the norma-

lized curvatures p l,p=.S,p .1, p = .01 and p -+0 (ATN (TAU)).

Fl

1.0

0.5

0 0

/ /

0.5 1.0 1.5

Fig. 2.4: Lower bound on the deflection angle ~D(T)(FI), for

different normalised screen curvatures, p.

-39-

The extremes for p = I and p ~ 0 can be determined inderyendently

of (2.12), (2.13) and (2.15), and can therefore be used to con­

trol the calculations. For p = I that ~s when RS = ~' ~D(T) will

be a linear function, the spot following a circular are with con­

stant speed.

For p ~ 0 (that ~s if RS ~ oo) ~D can be determined with figure

2.5.

s=vt

Fig. 2.5: Deflection geo~etry for a flat screen.

In fi8ure 2.5 the following equation holds:

vt

or vt arctan (~) = arctan T (2.16)

The results for these extreme values of the normalized curvature

p found in this way match well with the results found by numeri­

cal calculation and thus give confidence in the general results.

In this case we do not want the spot-velocity on the screen to be

constant but rather the projection of the spot velocity on the

x-axis. This situation is shown in fi3ure 2,6,

-4U-

Fig. 2.6: Deflection geometry for a curved screen viewed

by a distant observer.

Let this constant spot-velocity in the x direction be v, then the

following equations hold, see fig. 2.6:

or in the limit ~t + 0:

(2.17)

-41-

dcps Since dt

dcpo we find with (2.17):

t

V

With (2.10) we find:

with the same V, \v as in (2.12).

(2.18)

(2.19)

Figure 2.7 shows the results of the numerical calculations of cpD

for this case with p = I, p = .5, p =.I and p = .OI.

Fl

1.0

0.5

t

0 0

------------F I, RO= 1-:::----~-------,' ____ fi, RO=. ___ fi, ROz. __ FI. RO=.

I

/

/

/ ~ / ~

/ </ ;/,/

/ij //#'

// ;_/

0.5

/

/ /

/

/ /

/ / /

l.O 1.5

Fig. 2.7: Upper bound on the deflection angle~(T) (FI), for diffe­rent normalised screen curvatures, p.

-42-

Note that now vt =x (see figure 2.6), so if T ~s considered to be

a normalised distance rather than a normalised time, T is also

g~ven by:

vt x (2.20) 1

2.1.3 Bounds on the deflection current

If the relation between the current ~h and the angle ~D ~s known

then ih can be determined by:

(2.21)

The basic relation between ih and ~D ~s g~ven by Donald G.Fink [8]:

or

. 1 ,r nh ~h = , ;---;;:--' V E

jJ V e/2m

~h

c

0

D a . ,+,

l s~n'VD' f

(2.22)

where ih/C is a dimensionless quantity. The normalization factor C

is given by:

with

nh number of windings of the horizontal deflection coil

E electrical field strength in the cathode-ray-tube

the effective diameter of the coil

the effective length of the coil

space permeability) -9 l-l 4n.IO henry/m (free \~ 5 - 1 -1 VeiLID = 3.10 mV 2 sec

With (2.21) and (2.22) one finds:

-43-

This last equation with (2.13) becomes the lower bound:

pcoscfJD

dcps

d<tJD

and with (2.18) the upper bound:

pcos<PD

d<jJS coscp 5 -d'"'

"'D

(2.23)

(2.24)

Figure 2.8 shows the results, for the normalised curvatures p .5

and p

1/C

1.0

0.5

0

0

.2. Note that for p = .2, which almost equal to the P

0.5 1.0 1.5

pper-twnd

Fig. 2.8: Bounds on the normalised horizontal deflection current

ih/C (I/C), for different normalised screen curvatures, p.

-44-

of the picture tube (A66 - 540X) used in the breadboard model

described here, there is hardly any difference between the up­

per and lower bound on ih/C.

If (2.22) is differentiated with respect toT, we find:

dih d<jlD

dT c . dT cos<jlD (2.25)

For T 0 (2.25) becornes:

dih d<jlD cos<jlDI C- .

dT T=O dT T=0 T=0 c, (2.26)

in this <Pn = o, and d<jlD

s1nce case dT T=0 I.

For the case when the lower bound on ih/C is calculated, and the

nomalised distance T s C can be deterrnined by (2.26) - RD' as:

dihl dih dx~

dih (2.27) c TT T=O = ( dx ' dT ~ ds T=O s=O

For the case when the lower bound on ih/C is calculated, but the

norrnalised distance T x

Ru' C can be deterrnined in a sirnilar way

c dihl ~ dx x=O·

(2.28)

Using equations (2.27) or (2.28) as appropriate C can be deter­

rnined by rneasurernent of the deflection sensitivity coëfficient

dihl dih I ds (or dx ), by rneasuring s (or x) as a function of 1h s=O x=O

for a known ~·

The upper bound on the horizontal deflection current 1s deter­

rnined cornpletely by a given spot velocity v, and given ~ and

R5

(see par. 2,13).

.. 0 .. ... • • c .c u • .. .. ...

3

0

~ .. >

i5 t.J z ... a

3 a

:e ... .a ... f .; z ... a .. ...

-45-

The spot velocity v ~s determined by w and T : x a

V

w x T

a (2.29)

With the normalised deflection currents

~ and RS are implied in the normalised

implied in the normalised time T(=~).

calculated in par. 2.1.3,

curvature p(=:U), and v is s

A normalised sinusoidal deflection current can be written as

A sin~T • n

The normalised frequency ~s g~ven by:

~ Tr /2 Tr ,-;:::-zTl Tl

s~nce this current bas to be max~mum

tl v Tl

~

Now let

(2.30)

for T

(2.31)

(2.32)

then s~nce Ta= T1-a, where a ~s the line blanking time (see fi­

gure 2.4):

If the magnitude of the theoretically determined current is

(ih/C)E for T = ha' then the amplitude An of the normalised

current is given by:

with (2.30) and (3.32) we will find for A : n

A n sin(p .n/2) a

(2.33)

-46-

Figure 2.9 shows ~D and (ih/C) for the normalised curvature

o = .1, both in the upper-bound case. With 1 = 2.34 we find: a

A n

.855 1.09

This normalised sinusoidal current is also included 1n figure

2. 9.

l/C OR FI

1.0

<Pu

( ih/ c )E t -- --

0.5

I

0 ...J<.--;1--l-+··1 I I I I I I I I ++-+-t--+-t I I I -I I' I I I I I j I b 0.5 1.0 'Ta 1.5

TRU (=V*TIRDJ

Fig. 2.9: Normalised sinusoidal horizontal deflection current,

for O= .1

Suppose now that T 1s reduced by a factor k, a

T I a

T /k a (2.34)

-47-

This means that we have to lncrease the spot velocity by the

same factor k:

v' kv

Then also T changes,into:

T' v't kvt

kT RD ~

Thus T 1 ' will be:

v'T ' 1

kT1 Tl RD

Ho wever T does not change, Slnce a

T v'T ' kv 2:.

' a k T

~ RD T a a

With (2.32), (2.37) and (2.38) pa' can be determined by:

p ' a

(2.35)

(2.36)

(2.37)

(2.38)

(2. 39)

In agreement with the change in T • If the modified normalised a

current is given by A 'sin~'T' then: n

A' n sin( p 1T /2k)

a (2.40)

(2.41)

Figure 2,10 shows the upper bounds for ~D and ih/C if p = .1,

(same ~D and ih/C as in figure 2,9), tagether with a few exam­

ples of normalised sinusoÏdal currents.

Figures 2.9 and 2.10 demonstrate clearly the usefulness of the

sinusoÏdal current as a deflection current.

-48-

l!C OR F l

l.O

<~>o,E

0.5

l I

0 I

-f--1--+-t-+-1-+-+-+-l-+---+--+ +++--++--+- + +-+ --t----+':-1 I I I I I + l.O 1'a 1.5 D 0.5

Fig. 2.10: Normal sinusoirlal horizontal deflection currents,

for p = ,1, at different spot velocities.

2.2 Generation of the sinusoÏdal current

It is obvious that the sinusoirlal current can be generated with

the help of a resonance circuit. If the horizontal deflection

coil were to be driven directly by a source, that souree also

would have to exchange the reactive power with the coil. If a

resonance circuit is used, then the souree only has to deliver

the power dissipated in the coil resistance.

-49-

There are two principal types of resonance circuits that can be

used: The seriesresonance circuit and the parallel resonance

circuit.

2.2.1 The series resonance circuit

The series resonance circuit is shown ~n figure 2.11.

+

V

Fig. 2.11: Serie resonance circuit.

For figure 2.11 the following equation holds:

or V(jw) lh (jw)

v(jw)

. ( I Rh + J WL - -) h wc .

a

We want this circuit to be in resonance for w

(2.42) has to be real. Thus

c a T • 2f 2 -h'+TT h

w n

(2.42)

2TTlh' where

(2.43)

-51-

For figure 2.12 the following equation holds:

\(jw) V(jw) V(jw)

Ic(jw) + Ih(jw) = + . I Rh + JWLh

or It(jw)

V(jui)

jwc a

We want this circuit to be in resonance for w

(2.46) has to be real~ Then

c = a R 2

h

2 Normally Rh

2 2 << w Lh

c = a

That is the same value we found ~n (2.43).

(2.46)

(2.47)

The relation between the souree current ~s and the current ~h will

be found with:

I s

V

l.Jîth

IR + IC Ih V . V

+ =- + JWC V R. a Rh + jwLh ~

I s

I -+ jwC + R. a Rh + jwLh ~

V ~--~~--- the last equation changes ~n: ~ + jwLh

I s

( R ~ j wC) (Rh+ j wLh) + I ~

I s

or

..

(2.48)

-52-

So if l_ Assinwht, then s

l_h = Ahsin(wht-8) (2.49)

A Ah

s =

with

2.2.3 B~~~~~l-~~-Ei~:~~~~i~~-~ff~~~~-g~~li~~~iY~-~~~~i~~E~~i~~-~=

~~-~~Eli!~~~-P2~~~l~!i~g-~i~~~l

The souree signals v and i of the described resonance circuits s s .

should be amplitude modulated in order to correct for the pin-

cushion distortion. This amplitude modulation can be realised on

small signal level. The amulitude modulated small signal then

simply is fed to a power amplifier, which directly excites the

resonance circuit.

Note that with the conventional deflection circuit (described in

par. 1.1) the amplitude modulation has to be clone on large signal

level.

The modulating signal can be detemined qualitatively with the help

of figure 2.13 showinga picture with pin-cushion distortion in the

horizontal direction. This picture 1s s~etric with respect to the

x-asis, the modulating signal m(y) 1s determined qualitatively by

the even-order Taylor series in the space domain.

m (y) 2

= ao + a2y

-5.3-

y

Fig. 2.13: A picture with pin-cushion distortien in the hori­

zontal direction.

Here v is the spot velocity in the direction of the y-axis, and y

Tf the period of one field.

With y = v t the modulating signal m(t) can be written as: y

m(t) . •. . . . . , (2.50)

1n the time domain. Since the latter function is periodic with

the modulating signal m(t) can also be written as a Fourier

series:

where

m(t) b0 + b 1coswft + b2cos2wft + b3cos3wft + •••• ,

(2.51)

for all t,

2îT w =·

f

With the rnodulating signal m(t) given by (2.50) or (2.51) the

sinusoidal amplitude modulated horizontal deflection current is

given by:

-54-

(2. 52)

Note that equation (2.52) holds for -~Tf < t < !Tf if (2.50)

is used, and for all t if (2,51) is used.

2.3 Deflection current modulation due to cros

between the deflection circuits

Figure 2.4 shows the deflection yoke with the ring of ferronag­

netic material. Inside this ring one fincis the yokes for hori-

Fig. 2.14: Deflection yoke with ring of ferromag­

netic material.

zontal as well as for vertical deflection. Although the magnetic

fields for horizontal deflection ~ and for vertical deflection

H are orthogonal they influence each ether via the ring of ferro­v

magnetic material. The ring serves as a ~agnetic short-circuit for

both fields.

Consicier the magnetic circuit for the horizontal deflection. The

magnetic souree ih.nh sees two magnetic resistances R . and R t' m, ~ m, The farmer the magnet resistance inside the shield; the lat~

ter is the magnetic resistance of the ring itself.

Although the latter is much smaller than the farmer, its variatien

may cause trouble. This variatien in R is due to the variable m,r

field H caused by the vertical sawtooth current ~ . This variable V V

field H saturates the shield thus enlarging R . If R becomes V m, r m, r

-ss-

larger than the flux ~h for the horizontal deflection becomes

smaller. And with ~h' Lh becomes smaller.

Although this variatien in Lh is rather small, it will cause a

noticeable variatien in the phase e between the exciting souree

signal and the current ih of the resonance circuits, described

1n par. 2.3. The impact of this variatien of Lh on the phase 8

will be described below for both the series resonance circuit

and the parallel resonance circuit.

2.3.1. Variatien of 8 in the series resonance case -------------------------------------------The relation between the souree voltage v

5 and the current 1h 1s

given by (2.4S).

Let the variatien on Lh o.ue to variatien 1n 1 be óL , and let v m L h,m be the self indoetanee of the horizontal deflection coil for

i o, so that: V

Lh = L - L.lL h,m m (2.53)

with 0 < óL < óL m m,max

óL = 0 for i 0 m V

óL = L for l = + l m m,max V - v,max

2 In practice resonance will not be attained exactly, so Wh LhC

will be somewhat smaller or somewhat larger than I.

Let L = L + óL h,m res res

(2.54)

with L ~ -2-res wh ca

Then with (2.53) and (2.54) we find:

L + 6L - 6L (2.55) res res m

-56-

T.Jith (2.55) '\ and 8 from (2.45) will change into:

8

A s 2 , A ) 2

+ Wh luL -61 res m

wh(6L -61 ) res m arctan( R )_

t

(2.56a)

(2.56b)

In case of resonance 61 and 61 will bath be small. Suooose: res m --

and

lw(6L -61 )I<< R res m t }2.57)

I6L I has the same order of magnitude asi 61 I m res

Then Ah and 8 of (2.56) change into:

8

A s

Rt

wh(6L -61 ) res m

(2.58a)

(2.58b)

(2.58) show that the assumed variatien of Lh due to variatien 1n

iv' with the supposition of (2.57), will have no impact on Ah but

surely some on 8.

Since 61 > 0, see (2.53), 8 always becomes smaller if m

comes larger; 8 of course remains close to 0.

2.3.2.Variation of 8 in the narallel resonance case ----------------------~----------------------

li I be-v

The relation between the souree current 1 and the current 1h s 1S given by (2.49). With R. -+oo (2.49) changes 1n: 1

if 1 A sinwht s s

then 1h '\ sin(wht-8)

with

A s

-57-

1 + l\.1 r res l\.1m those values of Ah and 8 change into:

8

Now suppose:

arctan

A s

jw(l\.1 -l\.1 )[<< w~ C m res -ba

(2.59a)

(2.59b)

(2.60)

and jl\.1mj has the sameorder of magnitude as jll.1resl

Then Ah and 8 of (2.59) change into:

8

A s (2.6Ia)

(2.6Ib)

The variation of 1h due to variation of iv' with the supposition

of 2.60), will have no impact on Ah but certainly has impact on 8.

Here, too, 8 becomes smallerif ji I becomes larger; 8. V

Although the changes in 8 are small, in both cases, they are

noticeable. Especially in a system, which uses a symmetrie deflec­

tion current, as we shall see in par. 2.4.

-ss-

Here no choice is made between the two types of resonance

circuits. Although in the hardware realisation described in

chapter 3 a ser~es resonance circuit is chosen, this choice is

made for practical reasons. In chapter 4 th

discussed.

choice will be

2.4. Alignment of the lines scanned 1n opposite directions

If a symmetrie horizontal deflection current is used the R, G,

B signals, containing the picture information, have to be re

arrangend in time, because the signal coming from the trans~it-

ter corresponds to a scanning raster which

asymmetrie horizontal deflection current.

produced by an

the new types of

1 , necessary for a raster produced by a symmetrie horizontal

deflection current, will be defined.

R G B : The re-arranged vers1ons of R, G resp. B used sym' sym' sym

LR line

RL line

Line duet

LR scan

for scanning with a symmetrie horizontal de­

flection current. See figure 2.15 c and d.

The piece of the video signal(R , G , B ) sym sym sym which will be used to reproduce the picture

with the reproducing spot rnaving from ~eft to

~ight on the screen, see figure 2.15d.

The p of the video signal which will be

used to reproduce the picture with the repro­

ducing spot mov1ng from ~ight to Left on the

screen, see figure 2.15d.

A sequence of one LR line and one kL line in­

cluding the overscan parts, see figure 2.15d.

The scanning piece on the visible part of the

screen with the spot rnaving from Left to ~ight.

See figure 2.15e.

RL scan

-59-

The scanning piece on the visible part of the

screen with the spot rnaving from ~ight to Left.

See figure 2.15e.

If the scann1ng raster 1s produced by an asymmetrie horizontal

deflection current, the lines are written from left to right.

The scanning raster of the receiver described here is produced by

a symmetrie horizontal deflection current and this raster needs a

video signal consisting out of line duets.

The video information of every other 1 ine has to be reversed. \ifl1ich

of the lines of the incoming signal will become an LR line, and

which an RL line, will be determined with the "ST" signal:

ST signal The logic signal at half the line frequency

deciding whether the line of the incoming video

signal will become an LR line or an RL line.

If ST is "low", then the line of the incoming

signal will become an LR line; if ST is "high",

it will become an RL line. Fig. 2.14b shows the

relation between the ST signal and the line syn­

chronisation signal (the latter signal has al­

ready been shown in figure 1.4).

The resulting video information will be delayed by one line due to

the turning operation so the LR line coincides, ~n time, with the

"high" part of the ST signal, and the RL line with the "low"

part of the ST signal. In figure 2.15 this has been shown.

On the rece1ver screen the active parts of the LR lines have to be

aligned with the active parts of the RL lines, 1n order that a

straight vertical line in the original picture rema1ns straight on

the receiver screen and doesn't become rippled.

There are three ways to align these active parts of the LR and RL

1 ines:

-60-

2H

JI---L.---1 ------!.....1 J__l ~. t (a)

ST

(b)

R,G,orB

(c)

(d)

0

LR-line RL -irne

I i neduet ih

0 (e)_

LR-scan \ RL-scan over-scan OYer-scan

Tl Tl

Th

Fig. 2.15: The line synchronisation signal 2H(a); the ST signal

(b); the incoming video signal (R, Gor B)(c); the

resulting video signal (R , G orB ) (d); the sym sym sym horizontal deflection current (ih) (e).

-61-

( 1 ) By changing the current ih;

(2) by changing the spot position with the aid of 'n auxiliary

coil;

(3) by shifting the active parts of the video signals ~n time.

If the current ih is realised with the help of a resonance circuit,

then it will be very difficult to use methad 1. Methad 2 is possi­

ble, but requiers more electrical power than methad 3, and more­

over it might disturb the convergence and the purity of the three

electron-beams.

Methad 3 appears to be the one methad to use. That ~s why four

ways of alignment will be described here which all use shifting of

the active parts of the video signal.

These alignment methods will be described with the aid of figure

2. 16. This figure shows two periods of the line sync signal, one

period of the ST signal, a corresponding line duet and one period

of the sinusoÏdal current ih. The LR scan takes place from tLR,s

till tLR,e and the RL-scan from tRL,s till tRL,e" These four va­

lues of t deterrnine the active parts of the horizontal deflection

current. If the spot is in the extreme left and the LR scan is

going to take place then t = tLR,s" The other values are deterrnined

in similar ways. The time differences tv0 , tv 1, tv2 and tv3 are

related to the starting and stopping moments of the active parts

of the LR line and the RL line. The active part of the LR line

starts tv0 seconds after the rising edge of the ST signal and it

ends tv0 + tv 1 seconds after the rising edge of the ST signal.

The other values tv2 and tv3 indicate the start and the end of

the active part of the RL line.

Consider the four ways of alignrnent described ~n the following:

Linear alignment: The alignment ~s said to be linear if the re­

sulting video signal corresuonds to a linear

function of the incoming videosignal. If this

~s not the case then the alignment is said to

be non-linear.

Static alignment:

Line-duet static

alignment:

Line-duet dynamic

alignment:

ST

c><.l

\. tv

0 I

-62-

The alignment is said to be static if the time

differences tv0

, tv 1 , tv2 and tv3 (see figure

2.16) are the same for all line duets.

The alignment is said to be line-duet static if

the time differences tv1

, tv2 and tv3 are the

same for all line duets and tv0 is not the same

for one or more line duets.

The alignment lS said to be line-duet dynamic

if one of the time differences tv1

, tv2 or tv3 1s not the same for one or more line duets.

[ • I (a)

tv1 I tvz I" tv3 I

I R ,G ar Î sym s,m

I :~ ~: B sym

I I (b) I •t

ih r (c)

· 2 16 Th ST · 1 (a), one corresponding line duet of the re-Flg. • : e s1gna

sulting video signals (R G orB ) (b), and one s)llp.' sym sym

corresponding period of the sinuscaidal horizontal deflec-

tion current (c).

-63-

~~ich of these alignment methods can be used depends upon the

quality of the sinusoidal current ih.

Assume that the current ih satisfies the following conditions:

(a) For a eertaio value i of the vertical deflection the place of V

the spot on the screen is determined unambiguously by the in-

stantaneous value of the sinusoidal current ih through the ho­

rizontal coil.

(b) The amolitude of the sinusoidal current is varied such, that

the pin-cushion distortion (in horizontal direction) is correc­

ted. However, this variatien of the amnlitude is so small that,

for practical purposes, it can be considered constant, within

one line duet.

(c) The phase difference a 1 (see figure 2.16) between the rising

edge of the ST signal and thaifollowing zero-crossing of the

sinusoidal current ih, with d~ positive, is the same for all

lineduets; with a1

~ 100°.

Now suppose an alignment mechanism, which syncronised to the

positive edge of the ST signal, and which determines when and

with which rate the active parts of the LR and RL line are to

be written on the screen. Then, with conditions a, b and c ful­

filled, the intervals of writing on the screen are completely

determined by values of tv0

, tv1

, tv2

and tv3 which are the

same for all line duets. Hence, with conditions a, b and c ful­

filled, static alignment can be applied.

But if a resonance circuit used to drive the horizontal coil

then condition c might not be fulfilled anymore (see par. 2.3

and 2.4). It is obvious that a souree signal s , s being v or s s s is, can be made such that the phase difference, ao, between the

ST signal and ss is constant for all lines. (With a 0 being the

phase-difference between the rising edge of the ST-signal and

the following zero-crossing of the sinusoÏdal signal ss' with ds s dt positive).

Now let us for example take a series resonance circuit with souree

signal v . Then, if the supposition of (2.57) is fulfilled, s

-64-

L al = ao + e ce has been discussed in par. 2.4) will change if the

vertical current, iv, changes. If [ivf becomes larger than a1

will become smaller, and ih will be shifted backwarcis in time with

respect to the rising edge of the ST signal. If [i [ becomes smal-v

ler then ih will be shifted forwards 1n time with respect to the

rising edge of the ST signal. ~ig. 2.17 shows ih for two values

I I ,.

I =I / v v.max ---/

/ /

Fig. 2.17: One peri'od of the sinusoidal horizontal current (ih)

for two different values of [i [: [i [ = 1 and v v v,max [i I = o

V

i • So, if a series re-v,max

sananee circuit is used to drive the coil for horlzontal deflec-

tion, and if supposition (2.57) is fulfilled then a 1 will not be

the same for all line duets and the application of static align­

ment will give distortion. During the LR scan the instantaneous

value of ih will be larger for the case \vhen [i [ = 1 v v,max than

for the case when [i [ = 0; V

value of ih will be smaller

for the case when [i [ = 0. V

during the RL scan the instantaneous

for the case when [i [ = 1 v v,max than

Figure 2.18 shows this kind of dis-

tortion; the diverging (vertical) lines should remain together,

but that is only true in the centre.

Suppose that condition (c) is not fulfilled and static linear

alignment is applied: This gives the distartion just described.

Then replace condition c by condition

(c') The phase-difference a1

between the ST-signal and ih is not

the same for all lineduets but its variatien is so small

that, for practical purposes, it can be considered constant

within one line duet.

-65-

I I I I . I \i ! I i

'I . I l j

~ I 11 ,' I I i

!-·

i\

I ~~ 1\ 11 11

• 2.18: Distartion due toa change in phase in the

sinusoÏdal horizontal deflection circuit, (a):

undistorted grid, and (b): Distorted grid.

ST

0

I I

I l 1h ' I

t : I 1-:-:--1 I

I

0 I I

!

I

!\

(a)

(b)

(c)

Fig. 2.19: Line-duet static linear align~ent: The resulting

video signals (b) are shifted in time, with res­

pect to the ST signal (a), in the same way as the

horizontal deflection current (c).

-66-

t.Jith the conditions (a), (b) en (c') fulf led line-duet stat

alignment can be applied. Fig. 2.19 shows the result of that kind

of alignment for the lineduets with I i I i and I i I = 0. v v,max v How do we de termine tv , which will be different for each 1 ine-

o duet? Within one field, tv0 will be different for each lineduet,

but tv0 will be the same for the corresponding line duets in the

corresponding fields following, since tv0 depends on iv. So the

value tv0 will return periodically. Store these values and let

the line duet number within one period determine the appropr

value of tv0 to be used. Another way of determing tv0 is by de­

terming the phase difference a1

• This can Be clone by measuring

the current ih. Figure 2.20 shows a method to measure ih. In the

s resonance circuit used to drive Lh' the horizontal coil,

an extra resistor is inserted. The voltage over this resistor is

proportional to 1h.

F • 2.20: Series resonance circuit including a measuring

resistance Rhm·

Let the sinusoidal current be max1mum for t t , then with ma x the conditions (a), (b) and (c ') fulfilled (and of cour' se also

with (a), (b) and (c) fulfilled), the sinusciclal current, for

practical purposes, can be considered symmetrie with respect

to the axis t = t , within each line duet. That is why, to ma x achieve alignment, tv

1, tv2 and tv3 can betaken the same for

all line duets. If the sinusciclal current were no longer suf­

ficiently symmetrie, then line-duet dynamic alignment would

-67-

have to be applied.

Non-linear alignment might be used ~n combination with the above

kinds of alignment to reduce small geometrical distortions in the

horizontal direction.

In the hardware realisation described ~n chapter 3 line-duet sta­

tic linear alignment is applied. This choice will be discussed in

chapter 4.

2.5 The effects of the symmetrie deflection current on the scann~ng

Figure 2.22a shows the type of scanning raster which is used in

almast every TV-set at present. For convenience ~n every

frame only 25 lines have been taken; each field thus having 12.5

lines.

(a)

(b)

(c)

Fig. 2.21: The scanning raster (a) produced by two asymmetrie

sawtooth currents, (b) for the vertic al and (c) for

the horizontal deflection.

-68-

For simplicity the time between the A- and B-field has been

taken zero; whereas the time between the B-field from one frame

and the A-field from the next frame is supposed to coÏncidence

with the fly-back of the 25th line.

The deflection currents are shown in 2.2lb and c. These currents

are asymmetrie both for horizontal and vertical scanning, and

will be supposed to be linear. In practice the latter is not

true, as we saw in par. 2.1, but that is of no concern when des­

crihing the effects on the scanning raster.

If the horizontal deflection current is changed from asymmetrie

to symmetrie, the scanning raster will change to the one shown

1n figure 2.22a. Figure 2.22c shows the horizontal current as a

function of time for this situation.

-= F-=-

f:::-·

F==--·-=

~:::=--1----

---= f=·

---= F=-:"

_____-\t c======--=­ r

Vr À À À À À À À À À À À ;, V~ ~ V~~ V'V'V

(a)

(b)

(c)

Fig. 2.22: The scann1ng raster (a) produced by a sawtooth current

(b) for vertical deflection, and a symmetrie triangu­

lar current (c) for horizontal deflection.

-69-

If figure 2.22 is cornpared with 2.21 the following things will

attract attention:

In figure 2.22a the active line segrnents of the A- and B-field

interseet and that is sarnething that doesn't happen in the scan­

ning-raster of figure 2,2la.

The scanning raster shown in figure 2.22a has "holes" in the left

and right sides of the raster, whereas the active line segments

are hornogeneously spread over the screen in figure 2,2la,

If we campare figure 2.21b with 2,22b then we notice that the

maximurn value of the current ih for horizontal deflection in the

symmetrie case is larger than in the asymmetrie case. The diffe­

rence ~s about 6%. This is because, in the symmetrie case, the

fly-back time is "translated" ~n over-scan time and therefore the

current must be made larger.

The only way to get rid of the "holes", while at the sarne time

preventing intersections appears, to be re-arranging the lines such

that the lines of the A- and B-field are parallel. In the case of

symmetrical scanning the only possible way to manage this is by

changing the vertical sawtooth current into a "staircase" current,

such as the one shown in figure 2,23b. The horizontal deflection

current is the sarne as the one shown in figure 2.22b, Figure 2.23a

shows the resulting scanning raster.

The staircase vertical deflection current ~ can be realised by V

superirnposing the "norrnal" sawtooth current i , on a sawtooth cur­v

rent at line frequency in the vertical deflection coil L . V

This has been shown in diagram in figure 2.24.

Another way to achieve the wanted staircase deflection ~n the ver­

tical direction is by superirnposing a "norrnal" sawtooth rnagnetic

field H and a sawtooth rnagnetic V

in the sarne direction. The field

field, H , of line frequency, y,a H should be set up with the

V

"norrnal" sawtooth current through L , and the field H v v,a with a

sawtooth current of line-frequency through an auxiliary coil L v,a

-70-

(a)

r~.m~~ ~! ~ ~ l(b)

fr 1\ A A /\ A 1\ A 1\ 1\ 1\ 1\ G VVV V VVV V\fVVV (c)_

Fig. 2.23: The scann1ng raster (a) produced by a staircase

current (b) for vertical deflection, and a symme­

trie triangular current (c) for horizontal deflec­

tion.

b4 Fig. 2.24: Staircase current waveform realised by superimposing

two sawtooth current waveforms.

-71-

3. HARDWARE REALISATION

~. 1. Design of a breadboard model for sinusciclal horizontal

deflection currents

In par. 2.4 the lines of the resulting video signals (R , sym

G and B ) were divided into two types: LR lines and RL lines. sym sym Now we shall also devide the lines of the incoming video signals

(R, G and B) into two types, by defining:

PLR line: A line of the incoming video signals which 1s Predesti­

nated to become an LR line.

PRL line: A line of the incoming video signals which 1s Predesti-

nated to become an RL line.

The line types of the incoming and outcoming video signals are

determined by the ST signal, see par. 2.4.

Within each line a further subdivision will be made into the

active part and the blanking part. The active part has already

been defined in chapter 2; the blanking part is the complement

of the active part within each line.

The digital hardware described in this chapter has been realised

with TTL components. The two signal levels used are: "0" and ''1 ",

and they correspond with the "low" and "high" level used 1n par.

2.4.

If symmetrie scanning 1s used the video information has to be

reversed in every other line, and the video information of the

lines which are nat reversed has to be delayed one line time in

order to get the same over-all delay. These operations, which are

realised digitally, will be explained with the aid of the block

diagram shown in figure 3.1. The blocks which actually execute

these operations are: (pre) LPF, ADC, LR MEMORY, BLANKING, RL

MEMORY, DAC and (post) LPF. These blocks are present for each video sig-

nal (R, G and B), since the operations are the same for all three

video signals, it suffices to describe them for one video signal:

-72-

r!- LR- MEMORY f-1-M A c R U'

'--

10 r

R,G orB 3 ,....---------1 -1 A 0 C -Î

Cl I

BLANKING ~

[lÜ w I

c!. -1 RL -MEMORY & -

~ A CI.R l'i

I l -ç I

~ 10

A WE OE

,--J-+j-f--l2r CONTROL

I

2 H -----+----!2 H

2V ---,_...~2v

CLI CLCJ ( 1_LR CLRl Sf ST V!E VOE

I I l ' J

I l I I I CLI CLO o...:JR CLLR CLRL S ~ ST VIE VOE CU [lJ] CIDR ST

CLOCK/ST VIE/VOE

Cll ST

L------------1 2V SIN

's

--'-c

IHI1

IHM

PROC m+ii----J

Fig. 3.1: Block diagram of the breadboard model for sinusoidal

horizontal deflection.

-73-

The incoming video signal is low-pass filtered (LPF), AD-converted

(ADC) and stared in a line memory. In the case when the ST signal

is "0", the line of the incoming video signal

and the samples of the ~~ÈiY~ part of this 1

a PLR line,

are written into

the LR MEMORY in order of arrival. In the same time interval the

samples of the ~~!!Y~ part of the RL line are read out the RL

MEMORY in a last-in-f st-out (LIFO) order, and thus the line of

the resulting video s is then an RL line. In the case when

the ST-signal is "I", the line of the incoming video signal is a

PRL line and the samples of the ~~!iY~. part of this line are writ­

ten into the RL MEMORY in order of arrival. In the same time in­

terval the samples of the ~~tiY~ part of the LR line are read out

the LR MEMORY in a first-in-first-out (FIFO) order, and thus the

line of the resulting video signal is then an LR line. During the

time without read out the line memory, the line of the outcoming

video signal is blanked (=made zero). Finally the samples which

come out the BLANKING unit or the LR or RL MEMORY are DA-conver­

ted (DAC) and low-pass filtered (LPF).

The LR and RL MEMORY units are RAMs. Figure 3,2 shows their block

diagram. Figure 3.3 shows the BLANKING unit.

10

RAM f 5. 2l~o7l

WE

i I

\wr ifE fig. 3.3: BLANKING unit

fig. 3.2: MEMORY unit

-74-

In addition to the described basic units, 1.e: LPF, ADC, LR MEMO­

RY, RL MEMORY, BLANKING and DAC, five important blocks can be

distinguished, 1.e.:

I. CONTROL unit.

This unit generates the control signals for the LR MEMORY, RL

MEHORY and BLANKING units. Demultiplexing and multiplexing of

the digital video signals are done by an appropriate choice of

the WE and OE signals.

S Signals in: CLI, CLO, CLLR, CLRL, ST, ST, 2H, VIE and VOE.

Signals out:

ALR, Adresses LR-MEMORY;

ARL, " RL-MEMORY;

WELR, Write Enable for LR-MEMORY;

WERL, " " " RL- "

OELR, .Q_utput Enable " LR-"

OERL, " " " RL- "

OEBL, " " " ELANKIN unit.

2. VIE/VOE unit.

The VIE signal determines when the active parts of the incoming

lines are to be written into the line memories. The VOE signal

determines when the active parts of the resulting signals are

to be read from the line memories.

Signals in: CLI, CLO, CLOR, ST, STIHM.

Signals out:

VIE, Video In Enable;

VOE, Video Out ~nable;

3. CLOCK/ST unit.

This unit generates all the clock signals and the ST and ST s1g­

nal.

Signals 1n: 2H, 2V, STIHM.

Signals out:

CLI, CLOCK _!_n;

~75-

CLO, CLOCK Qut;

CLOR, 11 11 for Resetting the counter with which the VOE

signal is determinded;

CLLR, I! par. LR operations;

CLRL, 11 I! RL I!

ST, Determines the STate of the processor, see par. 2.4

ST, 11 11 I! " 11 11 11 11 11

4. SIN unit.

5.

This unit generates the sinusoÏdal souree voltage v for the s

ser~es resonance c it described in 2.3. This has been chosen

because it was then possible to use an existing audio power am­

plifier to drive the resonance circuit.

Signals in: CLI, ST, 2V.

Signals out: v . s

IHM PROC unit.

This unit uses the voltage IHM over the

If IHM > 0 then "0"; if IHM < 0

The r1s~ng edge of STIHM determines the

and with a 1 also tv0 , see par. 2.4.

Signal in: IHM, I Horizontal ~easured.

resistor Rhm.

then = "1".

phase difference

Signal out: STIHM, STate I Horizontal Measured;

al

These 5 blocks will be described in more detail in the pars~

3.2 through 3.6.

3. 2 The line memorHi!S and blanking control (CONTROL) unit

The addresses for the MEMORY units are generated by counters

(3 for each unit) connected in such a way that they forma 12

bit synchronous up/down counter (of these 12 bits only JO will

be used). These addresses are only generated during the time the

active parts of the video lines are written into or read out the

-76-

line memor~es. The LR COUNTER generates the addresses (ALR) for

the LR MEMORY. The RL COUNTER generates the addresses (ARL) for

the RL MEMORY, see figure 3.4. So to realise the output signals

LR COUNTER IT

CLLR

10

ALR

li CE RL IT

ST UID

CLRL CL

10

ARL

Fig. 3.4: Blockdiagram of the LR and RL COUNTER generating

the addresses for the LR and RL MEMORY units res­

pectively.

ALR and ARL of the CONTROL unit we have to realize the input s~g­

nals of the counters, i.e.:

LELR (LERL), Load Enable LR (RL) COUNTER;

CELR (CERL), Count Enable LR (RL) 11

U/DLR (U/DRL), ~pjQown LR (RL) COUNTER;

CLLR (CLRL), CLock LR (RL) COUNTER,

The two clock signals CLLR and CLRL need not be realized separate­

ly since they are also input signals for the CONTROL unit, so they

can be directly used as input signals for the LR and RL COUNTER

respectively.

Finally we have to realize the following signals: LELR, LERL,

-77-

U/DLR, U/DLR, CELR, , WELR, WERL, OELR, and OEBL.

The first four signals are realised with the ST and the 2H sig­

nals. The rest of the signals are realised with the VIE, the VOE

and the ST signals.

The LR and RL COUNTER are loaded on the rising and trailing edge

of the ST signal. The LR COUNTER 1.s loaded with "O"'s at both

edges of the ST signal. The RL COUNTER is loaded with "0"' s at

the rising edge of the ST signal, and with the content of the

RL COUNTER at the trailing edge of the ST signal. Here the

relation between the ST signal and the 2H signal (see par. 2.4)

has been used, and the counters are loaded by a monostable mul­

tivibrator (MMV) which is started at the trailing edge of the

2H signal, see figure 3.4

In this way the start and the end of the "O" part of the loading

signals (LE) are not well defined with respect to the edges of

the clock signals. However, this will give no timing problems,

because the "0" part of the loading signal will be a subinter­

val of the blanking part of the video lines.

The U/DLR is made "I". So whether there is written into or read

out the LR MEMORY, the LR COUNTER will start with ALR equal to

"0", see figure 3.4, and then count up, in agreement with the

fact that the LR line not reversed. For the U/DRL signal the

ST signal is taken. So when the ST signal is "I" the RL COUNTER

starts with ARL equal to "0" (see figure 3.4), and then counts

When the ST signal 1.s "0" the RL COUNTER starts with ARL equal to

the last ALR, from the preceding ST = "0" part, and then counts

down, in agreement with the LIFO reading sequence for the RL line.

With the VIE signal the active parts of the incoming video signals

are determined. The VOE signal determines the active parts of the

resulting video signals. With the VIE, VOE and ST signals the CE

signals for the LR and RL COUNTER are determined. The CELR and CERL

CLLR

U!IJLR

-78-

signals, in turn, determine, together with the ST signal, the

and OE signals. The WE and OE signals finally carry into effect the

demultiplexing operations on the incoming video signals, and the

multiplexing eperating on the resulting video signals, respective­

ly. Let for example the ST signal be "1", then read-out from the

LR MEMORY and writing into the RL MEMORY will take peace. Figure

3.5 shows the time diagram for this situation.

reaJmg in LR-MEMORY

n o o o n n.n !1D ---··--,-------

output

~i~~oo--F-F-+--~_;','_·D~..-~-J_O__j_,L.O__j_,L.L_ AWELR

CLRL

UltîRL

CERL

writ1ng in RL -MEMORY I

output ati:J~ a;~:~tss 0-FF Rl:ti----L-----L

AWERL

Cl

WERL 0 ___ L ___ __j_......J~-'---1----'------ ...1.---J.~-·-'- JODOODCJOC

Fig. 3.5: LR and RL MEMORY control signals during the time

interval in which read-out from the LR MEMORY and

writing into the RL MEMORY take place.

I

-79-

This figure shows that the WERL signal consists of pulses. In

practice these pulses are made with the help of another signal,

AWERL (~id for WERL s ignal).

This signal keeps the sarne value ("!) during the writing period.

The NAND function of the S\mRL signal and a delayed version of the

clock signal will give imRL. So here we will deterrnine the AWERL

(and also AWELR) signal rather than the \{ERL and suppose that both

signals are clocked with edge-triggered flip-flops (D-FF's).

Let the CERL signal change frorn "I" to "0" during clock period I,

and frorn "O" to "I" during clock period 1; in figure 3.5 this has

been shown. The value of ARL just before the RL COUNTER is enabled,

is in fact the first address sent to the RL MEMORY, because of the

D-FF on the RL MEMORY unit, see figure 3.2. This first address will

arrive at the RAM on the RL MEMORY unit during clock period 2.

Hence the AWERL signal has to change forrn "O" to "I" during clock

period 2. The last address will arrive at the RAM on the RL MEMORY

unit during the (l+l)th clock period. Hence the AimRL signal has to

change frorn "I" to "0" during the (1+2)th clock period. So the rela­

tion between the AWERL signal and the CERL signal is given by:

AWERL =ST. (CERL(-1) + CERL(-2))

Where "(-k)" denctes a delay of k clock periods,

or AWERL = ST.(CERL(-I).CERL(-2)

or AWERL = ST.(CERL(-I).CERL(-2) (3.I)

In a sirnilar way we find for AWELR, and OERL:

AWELR .(CELR(-J).CELR(-2)) (3.2)(3.2)

OELR ST.(CELR(-I).CELR(-2)) (3. 3)

OERL ST.(CERL(-I).CERL(-2) (3 .4)

-80-

\.Jith and known, OEBL can he determinded, since the three

OE-signals have to be mutual exclusive, so:

OEBL OELR + OERL OELR.OERL. (3. 5)

The relation between the CE-signals and the VIE and VOE-signals

are determined by the ST-signal, since:

CELR VIE.ST + VOE.ST = VIE. .VOE.ST

CERL VIE.ST + VOE.ST = VIE.ST.VOE.ST

or CELR = VIE. (3.6)

CERL = VIE.ST + VOE.ST = VIE.ST.VOE.ST (3. 7)

Now with (3. 1) through (3.7) the signals CELR, CERL, OELR, OERL, OE

, WELR and WERL can all he determined as a function of the sig­

nals VIE, VOE, and ST.

Figure 3.6 shows the hardware realisation with NAND-gates and D-FF's.

The WELR and WERL-signals are made with the auxilliary signal AWELR

and AWERL respectively, and a delayed version of the clock.

Fig. 3.6: Hardware realisation of the control signals

WERL, OERL, OEBL, CELR and CERL by means of the signals

VIE, VOE, ST and ST.

-81-

3.3 The memory enabling (VIE/VOE) unit

The time intervals during which the VIE signal is 11 I 11, the 11 1"

parts, determine when the active parts of the incoming video

signals are written into the line memories, whereas the 110" parts

of the VIE signal determine the blanking parts of the incoming

video signals. For the VOE signal the same holds with respect to

the resulting video signals. The VIE and VOE signals are deter­

mined with 12-bit up/down counters, see figure 3.4 (of these 12

bits only 11 bits are used). Each of these countersspansaline

duet, since that is the period of the display process with the

sinusoÏdal deflection current. The VIE and VOE signals have to be

present continuously, hence the counters have to be enabled all

the time. That is why special attention was to be payed to the

periodical reset. The counters are reset at the begin of each

line duet, that is at, or just after, the rising edge of the ST

signal.

Fig. 3.6 shows the block diagram of the unit which determines

the VIE signal. The upper part of the diagram shows the part which

determines the VIE signal during normal action.

The lower part used to load the RAM with a function from the

EPROM. This loading will have to be done when the power swit-

ched on, or if another VIE function is wanted. Note that the EPROM

cannot be used directly since it has an access time of 450 ns max.

while I clock period lasts 41~ ns!

The VIE RAM reset with the ST signal, after every line duet.

This reset takes place with an auxiliary counter: The VIE-RAM

RESET COUNTER. When the ST signal "0 11, this counter is loaded

with "!", at every positive edge of the clock. When the ST signal

becomes "!" the RESET COUNTER starts to countdown. At the moment

this counter reaches "0", a ripple carry out (RCO) resets the VIE

RAM COUNTER.

12 BITS VIE RAM SYNCHRON()J

-82-

VIE RAM COUNTER

12 BITS SYNCHRON()JS

VIE I l ~ c-------fl----ro

VIE RAM O-FFs I

+ VIE RAM __1_ UP/DOWN

+ COUNTER RE SET COUNTER

JP/DOWN 11 11 11 f C 0 U N TER l---t---"-i D- FF 1------+-t-t-i

(2x374)

RAM

1 x 4096 WE~

UI

---'r-- I 21471

J_ -

ViE SELECT I ON

SWITCH

VIE EPROM COUNTER VIE EP ROM O- FFs

11 r---- BCD F

MU X f----- ENCO- ~~ Tof--- DER'=".

12 BITS. SYNCHRONOUS

UP/DOWN COUNTER

~ VIE EPROM EPROM

11 8 x 2048 II 1-

11 11 f------+~---t D- FF 1-++--t->-t

_.__ 12716)

-:- 16

LOAD CONTROL

Fig. 3.7: Block diagram of the VIE unit generating the Video

In Enable (VIE) signal.

:I-

During the rest of the time during which the ST signal is "!" the

RESET COUNTER will nat produce a RCO again. If the RESET COUNTER

would be able to count down all the time, the time interval be­

tween two RCO's would be 4096 x TCLI" In reality the counter can

only count during I line period, that 1s:

for 20 MHz < fCLI < 30 MHzf (see par. 3.4). The clock frequency of

the VIE EPROM COUNTER is ~~I which is low enough to access the

EPROM succesfully, even if fCLI = 30 MHz.

During normal action the EPROM FF's and theEPROMare disabled and

VIE LOAO

BUTTON

-83-

and the RAM FF's enabled. With the VIE LOAD BUTTON a monostable

multivibrator MMV) is triggered, which produces a load pulse.

During the load time the EPROM FF's and theEPROMare enabled and

the RAM FF's disabled. At the same time wr enable pulses are

sent to the RAM. Note that the VIE EPROM COUNTER is never reset,

so the load pulse has to last at least

2048 seconds.

With the VIE SELECTION SWITCH a choice can be made from the 8 avai­

lable VIE functions in the EPROM.

The outcoming (resulting) video-signal enabling (VOE) unit

The VOE unit has the same block diagram as the VIE unit, only the

clock signals and the reset signal differ. The reset signal for

the VOE RAM COUNTER is not determined by the ST signal, since line­

duet static linear alignment is applied. Instead of using the ST

signal, the STIHM signal is used to reset the VOE RAM COUNTER.

If the rising edge of the STIHM signal ~s shifted forward in time

with respect to the rising edge of the ST signal, then the reset

moment of the VOE RAM COUNTER and the VOE signal self will also

be shifted forward in time. Thus tv0 changes; however, tv 1, tv2 and tv3 remain constant, just what line-duet static alignment re­

quiers. The alignment is also linear because the active parts are

read out with a contstant clock frequency.

But the STIHM signal ~s used the clock signal CLI cannot be used.

The clock CLI is locked to the 2H signal, and hence also to the ST

signal. The phase difference between the clock and the ST signal

therefore is constant. The STIHM signal, however, changes in phase

with respect to the ST signal and so also changes in phase with

respect to the clock CLI (The STIHM signal changes in phase because

the horizontal deflection current i changes in phase, see par. n

2.4. There will be line duets, during which the rising edges of the

STIHM signal and the clock signal CLI coincide or almost coincide.

And since there always will be some jitter on both signals there

-84-

will be differences, of 1 clock period in the reset-time of cor­

responding line duets (with the same i ) in fields following. If V

the reset time is shifted 1 clock period, both the LR line ánd

the RL line will be shifted 1 clock period forward ~n time. So on

the screen the LR line will be shifted TCLI'v (v being the spot­

velocity) to the right while the RL line will be shifted TCLI'v

to the left, giving a total difference on the screen of 2 TCLI'v

(=1,6 mm with the A66-540X tube used here). That is too much.

Note that a scanning raster produced by an asymmetrie horizontal

current would have a fault of only TCLI'v. Thus with symmetrie

deflection reset timing faults produce faults on the screen, which

are twice as big as with an asymmetr deflection current.

Instead of using the clock signal CLI, the clock signal CLO is

used as a system clock for reading. This clock signal made by

changing the phase of the clock signal CLI. Figure 3.8 shows the

block diagram of the unit which generates the clock signal CLO.

INTERVAL

DEtECTOR CODER

k

MUX

CL')

Figure 3.8: Generation of the clock signal GLO.

One clock period of the clock CLI is subdivided into k intervals

with the aid of delay elements (T). The k new clocks signals

-85-

created this way are used to detect the interval (1 through k),

in which the STIHM signal changes from "0" to "I". The CODER deter­

mlnes which of the k clock signals finally will be used (which clock

clock will be chosen and how this is done will be treated in par.

3.4).

Now consider three periods of the ST signal as shown in figure 3.9.

(a)

(b)

(c)

STIHM!

,] (d)

u ~~--~--'-----' ~[.' (e)

Fig. 3.9: Three line duets (a) and three corresponding periods

of the ST signal (b), the sinusoirlal horizontal de­

flection current (c), the STIHM signal (d), and the

STIHM signal (e).

In each ST period a new clock signal CLO will be determined.

So the clock signal CLO can be different in each ST period.

If the (i+2)th line duet is read out of the line memories then

the VOE RAM RESET COUNTER is started at the positive edge of

the STIHM signal in the (i+l)th ST period. At the reset moment

of the RAM COUNTER, that is in the begin of the (i+2)th ST

period, the clocks of the RESET COUNTER and the RAM COUNTER

-86-

have to be the same, otherwise reset cannot be done properly.

That is why the following strategy has been used:

The clock signal CLO to be used in the (i+2)th ST period is de­

termined in the ith ST period with the rising edge of the STIHM

signal. This clock signal will also be used by the RESET COUNTER

during the second half of the (i+1)th ST period and the first

half of the (i+2)th ST period.

This clock signal used by the RESET COUNTER is called CLOR. This

clock signal is changed at every rising edge of the ST signal.

The complete realisation of the unit producing the clock signals

CLO and CLOR will be given in par. 3.4.

In figure 3.10 the block diagram of the unit producing the VOE

signal has been shown. Note that the RESET COUNTER is loaded

with "0001100000000" (=384 decimal), so the RCO is produced a

quarter of an ST period after the rising edge of the STIHM sig­

nal, if fCLO = 24 MHz. VOE ..--~---1 u..l----

RCO VOE RAM COUNTER .----------------+---------t6 VOE RAM O·FFs

VOE RAM 12 BITS

SYNCHRONOO UP/DOWN COUNTER

12 BITS SYNCHRONCIJS VOE RAM

bits Q.6•9 11

RESET COUNTER

CLO

12 UP/DOWN

11 COUNTER t--+--+----i

VOE EPROM COUNTER

11

VOE EPRtJ-1 0-FFs

12 BIT:; SYNCHRONCIJS

UP/DOWN 11 COUNTER 1-----+----;

Fig. 3.10: Block diagram of the VOE unit generating the Video Out Enable (VOE) signal.

RAM

1. 4096 0{

(2147)

EPROM

e. 2048 ll'

11716)

LOAO CONTROL

VOE SELECTION

SWITCH

VOE EPROM

V8E

1- LOAD

.,ç BUTTON

CLO

-87-

To extend the static control possibilities for the alignment of

the LR lines with the RL lines (see par. 2.4), an extra delay unit

has been added. This unit enlarges tv2 , see figure 2.16, by delay­

~ng only the RL reading part of the VOE signal, and can be used

in combination with the eight VOE functions in the VOE EPROM.

Tagether they make a variable range of 64 clock periods in tv2 possible. The variation, in tv2 is, of course, a static variation,

that is to say if tv2 is changed then it is changed for all line

duets. Figure 3.11 shows the block diagram of this delay unit.

BCO

CO OER

re ss

fig. 3.11: Block diagram of the RL DELAY unit offsetting the

RL line with respect to the LR line

3.4 The clock and state signal (CLOCK/STl unit

The clock signal CLI has been made with an crystal Phase-Locked­

Loop (XPLL) which uses the line synchronisation signal 2H as re­

ference. The system has been built with a 24 MHz clock. But it

is also possible to use other clock frequencies between 20 MHz

and 30 MHz, if only the ratio of the clock frequency and the line

frequency is an integer. The video signal in the 100 Hz system

has a bandwith of 10 MHz, twice the bandwith of the video signal

in the 50 Hz system. So the clock frequency of 20 MHz, being

twice the signal bandwith, is a lower limit on fCLI' The upper

limit is determined by the components used.

VOE

-88-

The clock signal CLO has the same frequency as the clock signal

CLI but it differs in phase. The phase of the clock signal is de­

termined by the STIHM-signal. tlliy this is necessary and how it

is clone have been explained in par. 3.3. That paragraph (3.3)

also introduces the clock signal CLOR for the RESET COUNTER. The

block diagram of figure 3.7 can be extended to one that gives the

clock signals CLO and CLOR as outputs. Figure 3.12 shows the ex­

tended blockdiagram.

Cll k -r[DrE}.F(Drr······· ... --· ................. ----(D-

w0 cu1 cu2 CLs Q.lk- 1

l J ,.......:......~. ............... _ _,L...,

Si"'iii'M INTERVAL f-- COOER i--- MU X

DE1ECTOR

MUX

Fig. 3.12: Block diagram of the CLO/CLOR unit generating the

clock signals CLO and CLOR.

-89-

Figure 3.13 showshow the block diagram of figure 3.12 has been

realised with TTL components. One clock period of the clock sig­

nal CLI has been divided in 4, nearly equal, intervals with AND­

gates. With the flip-flop's D-FFla through D-FFld the interval

in which the rising edge of the STIHM

ned.

found, will he determi-

Take, for example, the case that the rising edge of the STIHM­

signal is in interval I. Then the Q output of D-FFlb will become

"!" and hence the other flip-flops D-FFla, D-FFlc en D-FFld will

remain "0", since they are reset by D-FFla by means of its Q out-

put.

Note that the propagation time of the AND-gate connected to the

R input of the flip-flop, has to he less than 1 quarter of the

clock period TCLI'

Fig. 3.13: Realisation of the CLO/CLOR unit.

-90-

The CODER of figure 3.12 which determines the clock number is for­

med by the BCD CODER, the ADDER, MMV2, triggered by the field syn­

chronisation signal 2V and FF1a through FF4b. MMV2 divides one

period of the field synchronisation signal into two equal pieces

(see figure 3.14).

2V

·---'----'------' [ ..

Fig. 3.14: Sub-division of one period of the field synchronisation

signal, 2V, into 2 equal intervals.

During the time the upper half of the screen is scanned, the r~s~ng

edge of the STIHM-signal will shift forward in time with respect

to the ST-signal. During the scanning of the lower half of the

screen the STIHM-signal will shift backward in time. The clock s~g­

nal CLOR to be used in the secend half of the (i+l)th ST period and

the first half of the (i+2)th ST period determined in the ith ST-

period (see figure 3.9 in par. 3.3).

During the time interval determined by the moment the clock signal

CLOR is determined, and the moment the STIHM signal starts the RESET

COUNTER (see par. 3.3), the signal will shift in time, in a

direction that is different for the upper and lower part of the

screen, see par. 2.4. Figure 3.15 shows this shift in time.

When the rising edge of the STIHM signal in the ith ST-period ~s

found in the jth interval, then in the upper half of the scYeen the

clock mod(j-~+4 ) will be taken and in the lower half of the screen

the clock j will be taken.

-91-

nsmg

t t t t 4

t edges ~loc~~

CLI 1 CLJ2

CLJ3

r1s 1ng

t t ~ ··± ·-S I .. ---·---·· (i•1) ST-period iS T-penod [i+1l ST-pericd

Ja.;er part screen upper part screen

Fig. 3.15: The change in the STIHM signals for the lower

and upper part of the screen.

The clock signals CLLR and CLRL are made by multiplexing the

clock sig~als CLI and CLO with the aid of the ST signal. The clock

signal CLLR is equal to CLI if the ST signal is "0", and equal

to CLO if the ST signal is "I"; the clock signal CLRL is equal

to CLO if the ST signal is "0", and equal to CLI if the ST

s i:gnal "I".

The ST and ignal are obtained by dividing the 2H-signal by

2, by means of a toggle FF.

3.5 The sinusoirlal deflection current unit

The sinusoÏdal deflection current ih is made with the aid of the

series resonance circuit shown in fig. 2.20 of paragraph 2.3.

The voltage-souree has been realised with a small signal voltage

souree followed by an audio power amplifier.

The signal and the small signal souree has to be:

1. Very stabil in frequency and phase

2. Very little distorted

J. Amplitude modulated, with an adjustable modulation function of

field frequency.

[ ll ( 24Miizl

ST

2V

-92-

4. Adjustable ~n frequency and phase.

The first two requirements are necessary because we want to apply

line-duet static linear alignment. The amplitude modulation ~s

necessary to compensate for the pincushion distartion in East-1-Jest

direction, see par. 2.2.3.

The frequency of the signal leaving the signal souree has to be

adjustable, because we also want to use it for a 50 Hz-system,

where a frequency of! 7.5 kHz is required. Adjustment of the

phase is necessary to cornpensate for signal delays in the ampli­

fier.

The digitally generated sinusoidal voltage from the signal souree

shown in figure 3.16, meets all these requirements.

I.MHzl _,_,

SIN

EP ROM COUNTER

I

R~SET V

SIN-EC

(2HHz)

13 SIN

a::XJres EP ROM

t I

I

2 I SIN- ~I

CONTROL 9 SIN- ~ EPRO~ ~ CONTROl I

COUNTER EPROH

RESET

I I I I

I SIN-CEC

J L_ __ _ ---

'Z

I I 2 V

16 PARALLEL IN DIGITAL

data SERlAl OUT dot a F lL TER

2

I

ANALOG ~ 1-d-a.,ta--t.__F_IL_TE_R--J~ /-

AOC

data

Fig. 3.16: Block diagram of the SIN unit generating the sinus­

aidal voltage driving the series resonance circuit.

-93-

In the SIN-EPROM 16-bit samples necessary to generate an amplitude

modulated sinusoidal voltage are stored. Instead of using a DA-con­

verter followed by an reconstruction filter, here a combination of

a digital filter followed by an DA-converter and an analog filter

~s used. This combination ~s also used in the "Compact Disc".

In this way a sinusoidal voltage with a distartion suppression of

more than 80 dB is obtained. Amplitude modulation is done by storing

all the different sample values of the sinusoÏdal voltage within one

field. The frequency and the phase of the sinusoidal voltage can be

changed by changing the samples in the EPROM's. The necessary control

signals are made with the aid of the SIN CONTROL EPROM.

3.6 The horizontal deflection curYent-measure~ent nrocess~ng

(n1M PROC) unit

This unit employs IHM, the voltage drop over the measurement res~s­

tor Rhm (see figure 3. I) to obtain the necessary infonmatien for the

applied alignment.

In this case line-duet static linear alignment ~s used and thus the wan­

ted information ~s the change in phase (a 1) of the deflection current

ih with respect to the ST signal. The zero crossings of the IHM sig-

nal give this information. The sinusoÏdal IHM signal H converted

into a "TTL" signal, the STIHM signal, by means of a smitt-trigger

(comparator with positive feedback, see figure 3.17). As worked out

in par. 3.3, the rising edge of this STIHM signal can be used to de­

termine the phase variation.

Figure 3.17 also shows an adjust'lhle de1ay, unit included for the

following reason. Consicier one period of the sinusoidal current

as shown in figure 3. 18 and two picture elements to be aligned,

x and y, each having a width of wCLO; wCLO being the width corres­

ponding with one period of the clock.

The time interval t between the "writing" on the screen of the y middle of picture element x and the middle of picture element y,

is given by:

t y (3. 8)

-94-

-- ....... STIHM

LAY UNIT

75

Fig._3.17: Hardware realisation of the IHM PROC unit, process~ng

the voltage drop IHM over the rneasurement resistor ~·

Fig. 3. 18: The position of two aligned picture elernents x and y

with respect to the sinusoidal horizontal deflection

current.

-95-

where k 1s an integer; given by the way the VOE signal 1s gene­y rated (see par. 3.3).

In case of perfect alignment

d d x y

or 2t + t x y (3. 9)

TCLI' where kCLI 1s an integer (kCLI is

an integer because the clock signal CLI 1s locked to the line syn­

chronisation signal 2H). Now s1nce TCLO = TCLI (see par. 3.3 and

par. 3.4), equation (3.9) changes, us1ng equation (3.8), into:

2t x (3. I 0)

6 where kx = 3kCLI - ky 1s an integer.

Thus to get proper alignment, 2t has to be an integer multiple of x the clock period TCLO

The time between the zero cross1ng

and the rising edge of the clocked

dih . of ih where ~ 1s negative,

STIHM signal resetting the

VOE RAM COUNTER, will in general not be an integer multiple of

TCLO; nor will the time between the moment when the sample x is

read out of the line memory and the moment when the picture ele­

ment x is actually written on the screen, be an integer multiple

of TCLO' Thus tx will in general not be an integer multiple on

TCLO' That is why a delay, adjustable between 0 and !TCLO' is

included. This delays range follows from (3.10), and is realised I

in practice, with a seriesof 8 AND gates, each with delay T6 TCLO'

-96-

4. DISCUSSION AND EXPERIMENTAL RESULTS

In chapter 2 the theoretical implecations of the use of a sinus­

aidal current for horizontal deflection have been considered.

In chapter 3 a hardware model has been realised based on these

theoretical considerations. Thereby choices have been made concer­

ning: (I) The amplitude of the sinusoidal current, and the way

the sinusoÏdal current has been generated; (2) the kind of align­

ment applied. These choices will be discussed here. Besides that,

we will consider the power dissipation, coil voltage and radiation

in the resulting deflection circuit and compare these fenomena

with the ones in the conventional deflection circuit.

4.1 g~~i~~-~~-~~E!i~~~~-~~~-g~~~!~~i~~-~~~~~~-~~-~~~-~i~~~~!~~! deflection current

The amplitude of the sinusoïdal current can be determined with the

horizontal deflection current giving a geometrically correct pic­

ture, see par. 2.1. This deflection current can be determined the­

oretically and experimentally. The theoretical current is deter­

mined by the radius ~ of the deflection system and the radius R5 of the screen. For the picture tube used here (A66- 540X), these

values are:

~ = 26.9 cm 114 cm.

The experimental current can be determined by driving the horizon­

tal deflection coil with an adjustable DC current, and measuring

the deflection of the spot.

Such a measurement has been clone. To this end, a ruler without pa­

rallax has been mounted in the xz-plane along the x-axis (see

figure 2.2) in order to obtain the current giving a geometrically

correct picture for a remote observer. The results of this measure­

ment are reproduced in table 4.1.

-97-

Table 4.1: Horizontal deflection current I, as a function

of the spot deflection, x.

1 l.n Ampère x 1.n cm

0 0 • I 05 .85 .206 I. 60 .309 2.40 .407 3.20 .507 4.00 .757 6.05 I. 000 8.10 I. 19 10.00 I. 90 12.00 1. 60 14.00 I. 79 16.00 I. 96 18.00 2. 11 20.00 2.28 22.00 2.41 24.00 2.52 26.00

(extreme right)

With (2.20) and (2.28) the values found 1.n table 4.1 can be used

to plot the measured deflection current 1.n the same way as the

calculated currents are plotted. Figure 4.1 shows the result.

In this figure the theoretically calculated current has also

been shown. For I/C = (ih/C)E the difference in T corresponds

to a difference of circa 5 mrn on the screen (with an A66-540X

tube). Figure 4.1 also shows the normalised sinusoÏdal current

for k=1 correspondig to the measured curve. The norrnalised

amplitude A is 0.7734, corresponding with an amplitude of 2.63 n

ampère in reality. Figure 4.2 shows the normalised sinusoidal

currents for k= 1.05 and k=l.l, both corresponding to the measu­

red curve. The latter sinusoidal current with k=l.l appears to

match the measured current well. For k=l.l the normalised ampli­

tude A would have to be 0.8071, corresponding with an amplitude n

of 2.75 ampère in reality, and the clock CLO would have to be

26.40625 MHz rather than 24 MHz (k=I); the clock-frequency being

-98-

l/C ~R FI

1.0 ____________ MERS, I IC, R~=. 2359649 I 2, C=3. 40 I 39 I 57~ ______ l/C, Rl'J=.235964912 -~------~---

- ___ 11 I. 293*5 IN (I. 32*TRU )----- - -------·- -----~

0.5

I ,'fa

0 -!L-+--f-+--+--+-+-+---+--+--+--+-+--lr---+--+--+--t-t--t-4--t---t-+' ---+---+-+-+-+-+-+ 0 0.5 1.0 1.5

Fig. 4.1: Comparison of the measured normalised horizontal

deflection current (A66-540X tube) with the norrna­

lised sinusoidal current at standard spot velocity.

an integer multiple of the line frequency f1

( = 31,25 kHz).

In the breadboard model described in chapter 3 we have taken k=l,

but with this hardware realisation it is also possible to use different

frequencies for the clock signals CLI and CLO; hence, since the am­

plitude of the sinusoidal deflection current can be continuously va­

ried, it is possible to use other values for k.

The theoretical current calculated 1n par. 2. I and the s1nus­

oÏdal current constructed to match it indicate very clearly that

--99-

1/C OR FI

1.0 T ····· ..... MERS. 1/C. "". 23596'912. c •'-'0 139!57 r ______ 1 / 1 . 2 3 9 * S I N I I . 2 * T RUl. k = 1 1 - -- - -- ---------~ ____ l/!.267*SINI!.258*TRUl, k=105---- --------------\ \

+

0

Fig. 4.2: Comparison of the measured normalised horizontal

deflection current with normalised sinusoÏdal

currents at different spot velocities.

a sinusoÏdal current may be used for horizontal deflection with­

out giving unduly large geometrical distortion. This has been

verified in practice. That is why par. 2. I has been of great

value. But if we want to determine the sinusoidal current gi­

V1ng minimal geometrical distortion, then the following questions

have to be answered first: How is geometrical distartion to be

defined 1n an objective manner, realising that the observer will

apply subjective criteria? Which current do we use as a reference

the theoretically determined current or the measured current?

-100-

The latter question is brought about by the small discrepancy

between the measured and the calculated current.

In the breadboard model described here we used k=l corresponding

to an amplitude of 2.63 ampère. This amplitude value only holds

for the horizontal deflection current in the xz-plane. The ampli­

tude of the deflection current in the other planes parallel to

the xz-plane is determined by amplitude modulation, see par.

2.2.3. In this case (Ah 2.63 ampère) az, of equation (2.50) was

taken equal to -4650, while the other coëfficients of (2.50) were

taken zero.

In this breadboard model the ser1es resonance circuit, described

in par. 2.2.1, has been taken to generate the sinusoÏdal deflec­

tion current. The only reason for this choice was that we then

simply could use an existing audio power ampl . In this re-

port no choice is made, however, a future choice should be made

by considering, inter al

- the power dissipation of the total deflection circuitry;

- the possibility of integration;

- the alignment involved.

The latter subject will be discussed 1n the following paragraph

(par. 4. 2).

4.2 Choice of al

With the hardware realisation discribed in chapter 3 it is also

possible to apply static linear alignment as described in par.

2.4. Therefore the clock signals CLO and CLOR have to be exchanged

by the clock signal CLI, the STIHM signal has to be exchanged by

the ST signal, and the VOE RAM RESET COUNTER (see figure 3.10)

has to be programmed with "1" rather than with "384".

With static linear alignment applied the distartion shown 1n fi­

gure 2.18b 1s obtained. So the supposition (2.57) of par. 2.3

appears to be fulfilled and then the deflection current will not

satisfy the requirements a, b and c statet in par. 2.4.

-101-

The distorted grid eau be used to measure the phase changes in a1

(see par. 2.4). By choosing appropriate VOE functions and by ad­

justing the delay of the RL line with respect to the LR line with

the RL DELAY unit (described in par. 3.3), the lines of the grid

shown in figure 2. 17b eau be made to coïncide everywhere on the

screen. Consider two different periods of the horizontal deflec­

tion current in one field, one corresponding to i = i 1 and V V,

one to iv = iv, 2 . The change in a 1 (~a 1 ) between these two diffe-

rent periods is determined by the number of clock periods, TCLO'

which the RL line has to be shifted in order to bring the lines

of the grid together at the vertical positions corresponding to

1 = i 1 and i = i 2. If this is done for the deflection cur-v V, V V,

rent periods corresponding to the top and the middle of the screen,

then ~a 1 appears to be determined by 8 clock periods, giving:

8 x TCLO a -a = 4 x 360°.

l,middle l,top Th

The factor ! states the fact that a change in a 1 corresponding to

one clock period TCLO produces a fault on the screen of 2v.TCLO

(v being the spot velocity), just like a reset timing fault, see -6

par. 3.3. With fCLO 24 MHz and Th= 64.10 , we find for ~a 1 ,tm:

In the top of the screen, with the situation shown in figure 2.18b,

this corresponds to a difference of 6,7 mm between the LR and the

RL line (with au A66-540X tube). The change in a 1 per line duet,

approximated by:

0 .012 '

where n 1s the number of lines in one frame (n=625).

With line-duet static linear alignment, as described theoretically

in par. 2.4, and practically in par. 3.3, the distortien as shown

-102-

1n figure 2.18b will disappear. Thus the deflection current seems

to satisfy the requirements a, band c',described in par. 2.4.

It should, however, he noted that the phase difference a 1 1s deter­

mined bye from (2.58b):

w(6L -61 ) res m

where the denominator Rt is given by:

Rt = Ri + Rh.

For the deflection coil used here (AT1270) Rh= 1.35Q. Ri is quite

large, however, since an extra resistor of 2.75Q is inserted in

the resonance circuit in order to match the deflection circuit to

the audio amplifier. By designing a power amplifier especially for

this resonance circuit, R. could he decreased, thereby increasing 1

the Q factor and decreasing the dissipated power. The decreasing

Ri howeve~would result in an increasing 6a 1, and hence it just

might he possible that the sinusoidal deflection current did not

satisfy conditions a, band c', stated in par. 2.4, anymore; so

that the application of line-duet static linear alignment would

give distortion. In this case line-duet dynamic linear alignment

could he considered, see par. 2.4.

4.3 Power diss and radiation 1n the sinusoÏdal

driven horizontal deflection circuit

In the breadboard model described here the losses in the horizontal

deflection yoke have been determined at a line frequency of 31.25 kHz.

The results are reproduced in table 4.2. The peak stored energy in

the yoke was 4.8 milijoules, about the same amount of energy as Bab­

cock and Wedam kept up during their measurements [3] (see also par.

1.2). If the results obtained by Babcock and Wedam (table 1.1) are

compared with the results obtained with the breadboard model des­

cribed here (table 4.2), it is noticed that in case of the sawtooth

-103-

Table 4.2: Losses ~n the horizontal deflection yoke

exited with a sinusoidal deflection current

ad 31,25 kHz

2 tot al eddy scanning p-p voltage total yoke yoke I R current & hys-frequency over Lh loss loss teresis loss

like deflection current the eddy current and hysteresis losses are

larger than in case of the sinusoidal deflection current. If, for

example, table !.la (at 31.5 kHz) is compared with table 4.2, the

total losses appear to be reduced from 15.2 1.Jatt, in the conven­

tional case, to 8.75 Watt, in case of the sinusoidal deflection

current. The power dissipated in the amplifier end-stage has nat

been considered yet, but it appears that this dissipation will

be far less than the dissipation ~n the horizontal output tran­

sistor and damper, especially at line frequencies of 31.25 kHz

and higher. There is also less power dissipated in the pin-cus­

hion correction circuitry of the breadboard model than in the eer­

responding circuitry of the conventional TV-set, because in the

farmer case the correction is realised on small signal level and

in the latter case on large signal level.

The high fly-back voltages in the conventional system limit the

self-inductance, Lh' of the horizontal deflection yoke. In the

system used by Babcock and Nedam [3J, see par. 1.2, the self­

inductance had to be lowered to limit the high fly-back voltages.

These voltages had to be limited because the components would

nat have withstood higher voltages. At the scanning rate of

31.5 kHz, they used a self-inductance of 0.26 mH to limit the

fly-back voltage to 880 volt peak-to-peak (p-p). In the bread­

board model described here, a self-inductance of I .35 mH was

used giving a voltage over Lh of only 700 volts p-p. Again the

deflection circuit using the sinusoidal current appears to be

superior.

~104-

The High Voltage (HV) for the picture tube in the conventional

system is being made by transforming the high fly-back voltages on

the isolation coil (see figure 1.6). In the breadboard model this

HV thus far has been made in the conventional way by using a dummy

coil in a waoden box. If syrnmetrical scanning actually is going to

be used a new HV souree has to be developped. This HV souree will

have to be developped independently from the horizontal deflection

circu • However, this will offer the possibility to optimize them

both separately (e.g. reduction of the internal resistance of the

HV source).

It is obvious that the radiation produced by the sinusoidal dri-

ven deflection circu is less than the radiation produced by the

conventional circuit, because the high frequency components due

to fly-back are eliminated.

-105-

CONCLUSIONS

It has been demonstrated, theoretically and experimentally that

a sinusoirlal current can be used for horizontal deflection ~n a

TV-tube without giving unacceptable geometrical distortion. The

alignrnent of the lines scanned in opposite direction was and will

be the major problem in a TV system, which uses a symrnetrical ho­

rizontal deflection current. Special attention will have to be

payed to the alignment, if the ampl ier-end-stage changed

such, that the Q-factor of the horizontal deflection resonance

circuit is increased, because this will enlarge the phase shifts

of the horizontal deflection current.

Owing to the use of the sinusoirlal current the power losses and

the high voltage have been reduced considerably. At a line fre­

quency of 31.25 kHz, as used in 100Hz TV-sets, the total losses

in the horizontal deflection yoke have been brought back from

15.2 watt, in the conventional case with the sawtooth like current

waveforrn, to 8.75 watt in the case of the sinusoirlal current at

the same line frequency. The peak-to-peak voltage over the hori­

zontal deflection yoke has been reduced from ca. 2500 volt to

700 volt, while maintaining the same self-inductance (1.35 mH) as

in the conventional system at 15.625 kHz. The radiation caused by

the horizontal deflection circuit has also been reduced.

In the breadboard model the large signal part of the horizontal

deflection system can be reduced to an amplifier end-stage, a ca­

pacitor a measurement resistor and the horizontal deflection coil.

If this compared with the large signal part of the horizontal

deflection system in the conventional TV (East-West correction in­

cluded), it appears that the power dissipated in the large signal

part will be far less ~n the breadboard model than in the conven­

tional TV; especially at frequencies of 31.25 kHz and higher.

The breadboard model, however, has extra small-signal circuitry

in the horizontal deflection system. Although the power dissipa-

ted this circuitry can be reduced by means of monolithic inte-

gration, it should be taken into account when the total power dis­

sipation of the breadboard model is determined.

Befere a full compar~son, concerning picture quality and power

dissipation can be made the following needs to be clone:

- The vertical sawtooth current has to be modified into a stair­

case current, see par. 2.5;

- a new way has to be found to generate the high voltage for the

picture tube independently from the horizontal deflection c

cuit;

- an amplifier end-stage, especially adapted to the horizontal

deflection circuit, used here, has to be designed.

The separation of the high voltage souree from the horizontal de­

flection circuit makes it possible to optimize them both separa­

tely {e.g. reduction of the internal resistance of the high vol­

tage source).

-107-

REFERENCES

[1] P.S. Carnt and G.B. Townsend,

Colour Television: NTSC-systern,

life Books, Ltd., 1961.

[2] P.S. Carnt and G.B. Townsend,

Colour Television: PAL, SECAM and other systerns,

life Books, Ltd., 1969.

[3] Uwe E. Kraus,

"Verrneiding des Grossflächenflirrrrnerns 1n Fernseh- Heirn­

empfängern",

Rundfunktechnische Mitteilungen, Jahrg. 25, 1981.

[4] W.E. Bab<".ock and W.F. Wedarn,

·~racticalconsiderations in the design of horizontal deflection

systerns for high definition television displays",

IEEE Transaction on Consurner Electronics, Vol. CE-29, No. 3,

August 1983.

[5] A. Bruce Carlson,

Corrrrnunication Systerns, second edition;

appendix C: "Television and Facsimile systerns",

Tokyo, McGraw-Hill, Kogakusha, 1975.

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[9] J.Davidse,

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