Design of converter from single-phase AC supply to 1kV pulsating DC supply having 1.2kW of power

Abhishek Dr. C. Lakshminarayana Student, M.Tech Power Electronics Professor and HOD Dept, of EEE, BMSCE Dept, of EEE, BMSCE Bengaluru, INDIA Bengaluru, INDIA

Abstract— This paper presents the design and simulation of a cost-effective, high-power quality and high-voltage DC-DC converter for pulsed power applications like Magnetron, Thyratron, Pulse transformer for Industrial and Defense applications. In this paper a system is proposed to convert single phase 230V, 50Hz AC supply to 1.2kW, 1kV pulsating DC power supply. The Electromagnetic Interference filter is used to minimize the interference induced by the converter. To achieve high power factor and low input current distortion while the output voltage is stable, the 1.5kW pre-regulator active power factor correction technology is used. For a high boost voltage, a soft-commutated full-bridge resonant DC-DC converter is used and the total pulse is obtained a pulsed direct voltage with a width of 1 millisecond and a turn-on time of 1μsec, using a switching module. Simulation of the proposed system is carried-out using MATLAB/Simulink for resistive load and results are tabulated by varying the supply and load, while maintaining the constant output voltage. Also, the obtained results are compared with half-bridge resonant, conventional full-bridge converters. Keywords— Electromagnetic Interference (EMI), Power Factor Correction (PFC), Total Harmonics Distortions (THD), Power Factor (PF), Full-Bridge Resonant DC-DC converter and Pulsed Power Supply.

I. INTRODUCTION

EMI filter used to reduce the noise induced by the power converter and it’s design are shown [1], and the problem is arising due to the sudden variation in voltage (dv/dt) and/or current (di/dt) in the level of waveforms and the design of EMI filter with PFC boost converter is shown [2], modelling and characteristics technique are used to find the suitable model of EMI design and the simulation are shown by using the parasitic inductance and parasitic capacitor of converter [3], the need of DC voltage from the grid uses the rectifier, which results poor power factor and to increases the power factor using passive or active power factor correction technology, [4] passive PFC provides tuning of LC circuit and gives DC output voltage, with an improved input power factor of 1 to reduce total harmonic distortion and also providing a regulated DC output voltage [5], with closed loop operation of the PFC boost converter in fractional order PI controller is shown [6], analysis and PI controller design of PFC boost rectifier are given [7,9]. A full-bridge resonant DC-DC converter is used for step-up application [13], from the switching bridge of square wave to sinusoidal current can be obtained and the design procedure for resonant tank operation shown [14], the Fundamental Harmonic Approximation (FHA) treats current and voltage waveforms as sine waves at the fundamental frequency and is a widely used technique in LLC analysis. Demonstrated the steady-state analysis and frequency response of LLC resonant

converter with loss calculation, and performed FHA-based LLC resonant converter design with ZVS and ZCS output diodes with wide output voltage range in [15,16], for generation of pulsating DC voltage IGBT switch with freewheeling diode is used and provided isolated gate drive [17], the solid-state device pulse generator using MOSFET / IGBTs and a pulse transformer avoids the limitations associated with PFN-based designs and is suitable for high-power applications [18], different intervals of "on" and "off" time can be used to produce different power levels, certain specifically built feedback systems employ "on-off" control of the power supplied to the magnetron [19], the need of high-voltage converters requires high-voltage insulation for safety, as well as precise adjust the current and voltage to maintain the constant current needed to drive the magnetron [20], hence high power quality is necessary for industrial application.

II. METHODLOGY FOR PROPOSED SYSTEM

Figure 1 shows a block diagram of the proposed pulsed power system.

Figure 1: Block diagram of proposed system for pulsed power supply

A. DESIGN OF ELECTROMAGNETIC INTERFERENCE (EMI) FILTER

The noises are produced by conduction and radiated emission in converter and conducted noise is produced via a wire which are divided into two category a differential mode noise and common-mode noise then connected to the converter as shown in figure 2 [1,2].

Figure 2: EMI filter structure

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The corner frequency for common mode (FR,CM) noise is 28 kHz, whereas the corner frequency for differential mode (FR,DM) noise is 20.5 kHz for any boost power factor correction converter [2], and the calculation of required components for EMI filter are as follows:

1) COMMON MODE COMPONENTS For the selection of components for reduction of common-mode noise first need to choose the Cy1 and Cy2 that is limited to 5400pF because of the time limit for receiving an electric shock from a stored charge if the power source is removed from the AC line. Choosing Cy=Cy1=Cy2 to the largest value of 5400pF for the smallest choke according to (1)

LCM=

∗(,)

∗…………………….………… (1)

=

∗

∗ = 3mH

LCM will be 3mH according to (1), and the leakage inductance will be 1.25 % of 3mH for differential mode, resulting in Leak being 41.25µH. The leak is usually between 0.5 and 2% of the common mode inductance [2].

2) DIFFERENTIAL MODE COMPONENTS

After knowing the value of FR,DM and leak we can easily find the value of Cx1 and Cx2 from (2)

Cx1 = Cx2=

∗,

………………………… (2)

=

∗

.µ = 1.46µF

B. FULL-BRIDGE DIODE RECTIFIER

From figure 3, full-bridge diode rectifier circuit diagram, which converter ac to dc but it’s has large ripple at output voltage hence it adds the high value of filter capacitor (Cout) that results reduced output voltage ripple but it makes input current non-sinusoidal periodic waveforms at the input side, hence it will create a lot of lower order harmonics as shown in figure 4 around 282.93%, which results in poor power factor because of increasing the harmonics distortion at a fundamental frequency as shown in figure 5, to improve the power factor need to add power factor correction circuit.

Figure 3: Diode full-bridge rectifier circuit diagram

Figure 4: Input and Output voltage with current waveforms of rectifier

Figure 5: Input current waveform THD measured by FFT Analysis

C. SIGNIFICANCE OF POWER FACTOR

Conventional rectifiers use diodes/thyristors to achieve a controlled/uncontrolled DC output voltage. They suffer from input current distortion and low power factor, as shown in figure 4. Another disadvantage is the need for large value filter capacitors resulting in large converter size. A reduction in harmonic content with increasing power factor (PF) can be achieved using passive or active power factor correction techniques. The passive technique uses an LC tuning filter. However, this method is not effective if the PF load changes, passive network elements such as inductors and capacitors are tuned and set to filter out high-order harmonics. But there are some limitations, in which the value of PF cannot be effectively improved. In addition, the size and weight of the device is increased. Power quality is a major concern and international requirements, such as IEC 100032, have been enacted to limit harmonic currents drawn by offline equipment. As a result, the PFC technique has become common practice in front-end converters to ensure that harmonics are kept low and that the input current waveform is close to a sinusoid [5]. In addition to THD regulations, standards for high frequency EMI limits, such as EN55022, are also requirements for power supplies [3]. Power factor is defined as the ratio between the real power and the apparent power and its value varies from 0 to 1.

=

……………………….. (3)

PFC is used in switch mode power supplies (SMPS) to make smoothen the pulsating AC input current, maintaining its voltage and current waveform are in-phase and improving the PF. Active PFC gives can be made with the combination of power electronics switch and passive elements that obtained PF nearer to 1. The active PFC function can be performed by the input average current mode command or the peak current mode command. Active PFC can be implemented using buck, boost, buck and boost topology [5]. Required step-up voltage hence go for boost converter and its advantages like gate drive circuit are easy due to lower side switch. An active PFC with a boost converter is addressed in

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this work. Depending on the current passing through the boost inductor during the switching cycle, the boost switch can operate in discontinuous conduction mode (DCM), continuous conduction mode (CCM) and critical conduction mode (CrCM). Compared to CCM and CrCM, DCM mode is more suitable because it has a higher peak current through the energy transfer inductor. In CrCM mode, the switch lights exactly where the current through the energy transfer inductor is zero, which necessitates a very accurate and quick sensing and control circuit. CrCM causes increased copper and core losses [5]. Because the CCM switch does not turn on at zero-current, these restrictions can be addressed by regulating it with a fast reverse recovery diode to decrease losses. A power factor correction method can be achieved by boost converter because of its advantages.

D. BOOST CONVERTER

Boost converters are most often used because they have several advantages over other APFC circuits. A boost converter mainly consists of a boost inductor (L), an IGBT or MOSFET as power switch and a boost diode (D), as shown in Figure 6. The switch (S) is operated by the controller. IGBTs or MOSFETs are controlled by Pulse Width Modulation Technology (PWM) operating at a fixed switching frequency, but the pulse width varies with duty cycle. A circuit representation of the switch and off states is shown in figure 7.

Figure 6: Conventional of boost converter

(a) (b) Figure 7: Equivalent circuit (a) when switch is ON and (b) when switch is

OFF

During the turn on, the boost inductor current is connected across the input voltage and is generated with a

slope of

which the inductor stores energy during this

period. At turnoff, the boost inductor current drops with a

slope of ()

. Inductors connect across inputs and

outputs through diodes to provide energy during this period. During one switching period, the average voltage across the inductor should be 0 based on the volt-second balancing principle [8]. As a result, the boost converter output voltage equation is

Vout = ()

..…………….………………………...….…… (4)

E. CONROLLER CIRCUIT DIAGRAM PFC BOOST CONVERTER

We can see, that from the proposed block diagram from figure 1, PFC converter consists of a single-phase full-bridge diode rectifier and boost converter. And for the control circuit, consists of the reference voltage (Voref), voltage error amplifier (VEA), multiplier (M), current error amplifier (CEA), and pulse width modulator (PWM) as shown in figure 8 [9].

Figure 8: Control strategy of rectifier with PFC boost Converter

Now we'll talk about the PFC principle, the output voltage is compared to the reference voltage, and the result is sent to the voltage error amplifier. The output of the voltage error amplifier and the rectified input voltage are fed into a multiplier, and the multiplier's output is used as the current feedback control reference. The result is sent into the current error amplifier, which controls the on and off of the switch S, after comparing the reference current with the input detected current. As a result, the input current and rectifier input voltage may be almost in phase, and there is less harmonic current, allowing us to enhance the power factor and stabilize the output voltage. Inductor current waveform for average current mode control with variation of duty cycle is shown in figure 9.

Figure 9: Average current mode control inductor current waveform

The primary issue with peak current mode regulates is that we are trying to control the peak of the inductor current while failing to ensure that the average input current is proportional to the input current. And in other circumstances, it will produce a big mistake, resulting in enormous distortion that we did not expect, with fluctuating duty cycles that are sometimes greater than 0.5 and sometimes lower than 0.5. The peak current, on the other hand, is extremely susceptible to noise. As a result, the active power factor correction technique is recommended. We take the result of multiplying the rectified input voltage by the amplified error signal of the output voltage and calling it the reference signal. We can regulate the average current and bring it in phase with the input voltage via current loop regulation. The input current is

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immediately detected and compared to the reference current, after which the current error amplifier averages the high-frequency components. The amplified average current error is then compared to the sawtooth wave ramp, and the switch driving signal is generated, which determines the duty circle. As a result, the existing mistake will be quickly eliminated [12]. The UCC28180 is used for controlling the active power factor correction for the boost converter and it is operating at constant switching frequency in continuous conduction mode. The UCC28180 requires some external components for the function of the active PFC pre-regulator. The UCC28180 uses two control loops. The internal error amplifier and 5V reference provide a slow outer loop to control the output voltage. External compensation for this outer loop is applied to use the VCOMP pin. The internal current loop forms a common input current to form a sinusoidal input voltage [10]. The internal current loop does not need to detect the input voltage through a means to improve the duty cycle by utilizing the connection to and from the input voltage. External compensation for the internal current loop is applied to take advantage of the ICMP pins. The switching frequency is programmed from 18kHz to 250kHz with the addition of an external resistor designed over ground. The UCC28180 consists of a number of safety features designed to be highly reliable and can provide safe operation in all conditions, including above or defect conditions [10,11]. The advantage of average current mode control is that the variable is the average of input current, so the THD and EMI is small; it is not sensitive to noise; it can work under both CCM and DCM mode, and the switching frequency is constant so it is good for high power applications. And this is the most widely used control mode in PFC.

F. FULL-BRIDGE RESONANT DC-DC CONVERTER

Figure 10: Full-Bridge resonant DC-DC converter

A full-bridge resonant DC-DC converter is used as a step-up after the PFC boost converter, using a high-frequency transformer and a (second) full-bridge rectifier to convert into DC. The load looks to be in series with the LC resonant tank, hence it is also, called as series-loaded resonant converter. In which the primary side leakage inductance of the transformer is used as the resonating inductance, hence no need for additional inductance. Figure 10, shows a typical full-bridge LLC resonant converter circuit diagram where Lr is the resonant inductor created by the leaking inductor, Lm is the magnetizing inductor and Cr is the resonant capacitor. S1, S4 and S2, S3 are main switches that are switched on and off with a 50% duty cycle and 180º out of phase, and the LLC

converter regulates them via frequency modulation to regulate the output voltage. Because the impedance of the resonant tank, produced by Lr and Cr, is zero at the resonant frequency, the input and output voltages are essentially linked together. As a result, converter voltage gain is one for all load situations. When an input AC line is present, the PFC boost converter stage generates DC-DC stage input voltage at the front-end, which is regulated at 380V(min) to 400V(max) and 390V(nom). In this scenario, by selecting a suitable transformer turns ratio, the converter may always run at the resonant frequency. Conduction and switching losses can be minimized as a consequence. During holdup time, the large holdup time capacitor distributes energy to the load. The converter's switching frequency lowers as the DC-DC input voltage falls, allowing it to operate in boost mode and control output voltage. Because of the complexity of the resonant tank, the LLC resonant converter's design must consider three key factors: Inductor ratio, resonant frequency, and characteristic factor.

Figure 11: Equivalent circuit of resonant converter

The converter gain is given by the product of switching bridge gain, resonant tank gain, and transformer turn ratio. And the switching bridge gain for the half-bridge is 0.5 and the full-bridge is 1, the resonant tank can be analyzed by drawing the equivalent circuit as shown in figure 11. The magnitude of the transformer function is obtained at zero initial condition is given by (5) [14]:

K(Q,m,Fx)= _()

_() =

²()

(.).().().²...................................... (5)

Where Q is quality factor, Rac is reflected load resistance, fx is normalized switching frequency, fr is resonant frequency and m is the ratio of total primary inductance to resonant inductance.

Q = (/)

………………..…………………………... (6)

Rac= ∗²

∏²∗² =

∗∗²

∏²∗∗……………………...………. (7)

Fx =

................................................................................. (8)

Fr =

∏(∗).…. ………………………………..……… (9)

m =

.……………………...………………………. (10)

As shown in figure 12, all gain curves include the peak value that defines the boundary between the inductive and capacitive impedance of the resonant tank. Therefore, we will use shading to designate the inductive and capacitive operating areas in figure 12 [14]. Because it is desired to maintain inductive operation over the entire range of input voltage and load current, and never slip into the operating capacitive area, these two areas must be defined. Because zero voltage switching (ZVS) can only be performed in the inductive region, there is such a demand. In addition,

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capacitive operation means that the current precedes the voltage, so the current in the MOSFET will reverse before the MOSFET is turned off, and the reverse current will flow through the diode of the MOSFET body after the MOSFET is turned off. Diode switch. Once the other MOSFET in the bridge is turned on, this will cause reverse recovery losses and noise, and may cause high current spikes and device failure [14].

Figure 12: Load operation analysis curve, gain vs normalised switching

frequency

G. DESIGN METHODOLOGY OF LLC RESONANT CONVERTER

Figure 13 shows the design flow chart of LLC resonant

converter [13].

Figure 13: Flow chart for design of LLC resonant converter

1) Step 1: Selecting the Qmax value, quality factor depends upon the load current, under heavy load high value of Q is required and under light load lower value of Q is required, figure 14, shows the gain vs normalized switching

frequency at different Q values like 0.2, 0.3, 0.4, 0.5, 0.7 and 1. Figure 15, plotted with different Q values will depend upon the gain Vs normalized switching frequency we can analyze at the peak of the whole curve with our required gain should be higher than the calculated gain, from figure 15, we can see that selected Q as 0.4 and Mg_max as 1.2.

Figure 14: Gain Vs Normalized switching frequency at different Q value

Figure 15: Gain Vs Normalized switching frequency at selected Q=0.4 value

2) Step 2: Selecting the m value, lower the value of m higher the boost gain, and more flexible operation. Higher the value of m, the higher the magnetizing inductance and the higher the efficiency. Hence the low value is chosen and plotted the graph as shown in above figure 14 and figure 15.

3) Step 3: After selecting the value of Qmax and the initial value of m to find the minimum normalized switching frequency, we want to find the minimum normalized switching frequency, which will ensure inductive operation under conditions of Qmax (maximum load), and this minimum frequency as well it will. Ensure inductive operation of all other loads.

4) Step 4: Voltage gain verification, this is an important step to find the gain value that should be higher than the calculated gain as shown in figure 15.

5) Step 5: Calculating resonant components values as shown above the equation from 6 to 10 based on that the required values of Lr and Cr are calculated. There are three different modes of operation which will happen at resonant, below resonant, and above resonant as shown in figure 16 [14].

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Figure 16: Different modes of operation

Modes of operation: • At resonant tank has unity gain and best-optimized operation and high efficiency, therefore, transformer turns ratio is meant such the converter operates at now at nominal input and output voltages. • Below the resonant frequency the converter operates during this mode at low input voltage, where a step-up gain or boost operation is required. • Above the resonant converter operates during this mode at higher input voltage, where a step-down gain or buck operation is required [14].

H. SWITCHING MODULE

The switching module is used to convert into pulsating dc voltage from regulated output of the full-bridge resonant converter with a fixed switching frequency of 5kHz. For switching IGBT switch is selected having a high voltage rating for safety.

Figure 13: Switching module

The IGBT switch is connected in series with the 1kV power supply that is the output of the resonant full-bridge isolated DC-DC converter then connected to resistive load. The operating switching frequency is 5kHz, for IGBT and the total time of pulse width is 1msec, 1µsec is turn ON time of the pulse with maintaining the 1000V pulsating DC voltage,

1.2Amp of current, and having a total power of 1200Watt.

III. SPECIFICATIONS & DESIGN METHODOLOGY:

A. DESIGN OF 1.5kW PRE-REGULATOR PFC BOOST CONVERTER

AC Input voltage:185-265Volts, Input frequency:47-63Hz, Power factor: ≥0.9, Switching frequency:45kHz, Output voltage:390V(min) to 400V(max), Allowed ripple in output voltage: ≤2%, Efficiency: ≥90% and Hold-up time:21.27msec. By considering the efficiency of 90%, the input power drawn by the converter can be calculated as,

Pinput = ()

=

.= 1666.67W…………………. (11)

The maximum average output current Iout(max) can be calculated as

Iout(max)= ()

=

= 3.845A…………………... (12)

The maximum input RMS line current IIN_RMS (max) is calculated by assuming the efficiency and power factor of the converter, and Vin(min) is185V.

IIN_RMS (max)= ()

∗()∗=

.∗∗.= 10.01A … (13)

Assuming that wave form is sinusoidal, based upon the calculated RMS value, the maximum input current IIN_(max) can be determined.

IIN_(max)= √2* IIN_RMS (max)= √2*10.01=14.15A………... (14)

Then, IIN_AVG (max)=∗

∏=

∗.

∏= 9.012A ……….. (15)

UCC28180 IC has user programmable switching frequency with a single resistor on the grounded FREQ pin. In this design, the switching frequency Fsw was selected at 45kHz. To set the switching frequency, you can choose the appropriate resistor or calculate the value using the constant scaling values of FTYP and RTYP. In all cases, FTYP is the same constant as 65kHz, RINT is the same constant as 1MΩ, and RTYP is the same constant as 32.7kΩ. Simply applying the calculation below will generate the appropriate resistors that need to be placed between FREQ and GND [10]. RFREQ= (FTYP*RTYP*RINT) / (FSW * RINT) + (RTYP * FSW) - (RTYP * FTYP)………………………………………….. (16)

RFREQ=∗.∗

(∗)(.∗)(.∗)=47.93kΩ.

………………………………………………. (17) Selecting 47kΩ frequency resistor and the switching frequency will be around 44kHz.

1) Input Capacitor Input capacitors are selected based on input current ripple and high frequency input voltage ripple. Inductor ripple current ΔIRIPPLE, 40%, and high frequency voltage ripple will be considered from 3% to 9% coefficient allowed, choosing the ∆VRPPPLE_IN of 7%. To calculate CIN need to determine the input current and input voltage ripple as follows. IIN_RIPPLE = ∆IRIPPLE x IIN_(max)=0.4 x 14.15=5.66A…….. (18) VIN_RIPPLE = ∆VRPPPLE_IN x VIN_RECTIFIED(MIN) = 0.07 x

(√2x185) =18.31V…………….……………………….. (19) Maximum input capacitor value

CIN=

∗∗=

.

∗∗.= 0.860µF ………… (20)

To regulate the output voltage at the minimum input voltage by varying the pulse width of the controller with a duty ratio of Dmax, and same for the maximum input voltage condition by varying with pulse width of the controller with minimum duty ratio Dmin. To find the expression for Dmax and Dmin can be derived from (4) and expressed as

Dmax =

=

=0.52………..………… (21)

and Dmin =

=

= 0.32 ………… (22)

2) Boost Inductor

LBOOST (min) = ∗∗()

∗=

∗.∗(.)

∗.=

340µH………………………………………………… (23)

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Considering the higher value of inductance because the load

is varying from no-load to full-load and converter is operating in continuous conduction mode (CCM).

3) Boost Diode Silicon carbide Schottky diode is chosen based on the elimination of reverse recovery losses and lower power dissipation hence efficiency of the converter will increase and the diode losses are estimated based upon the forward voltage drop, VF at maximum operating temperature and reverse recovery charge of the selected diode.

4) Output Capacitor The output capacitor is sized to meet the requirements of the holdup time during one cycle. Assuming the downstream converters require the output of the PFC stage to never fall below 300V, VOUT_HOLDUP (min), during one line cycle [10],

tHOLD-UP =

()=

= 21.28msec………………. (24)

now calculate the minimum value of output capacitor requirement is given by

COUT (min)=∗()∗

² _() =

∗∗.

²=

1028µF……………………………………………..….. (25) Practically selecting capacitor higher value by 10% the actual value, hence 1130.82µF is used. The voltage divider samples the output voltage using a sampling resistor. When the output reaches the level of 390V, the voltage of 5V is displayed on R13 as 13kΩ as shown in Fig. 11. Compares the detected output to the reference, amplifies the error, and changes the duty cycle compared to the voltage of the current sense resistor. Switch [5]. The value of the current detection resistor is calculated in [10]. The pre-charge boost diode is connected to the output of the boost converter with a rectified DC output to support pre-charging the output bulk capacitor up to the peak of the AC line voltage [5]. The overall design schematic of EMI with the PFC boost converter is shown in figure 18.

B. DESIGN OF FULL-BRIDGE RESONANT DC-DC

CONVERTER

From the flow chart figure 14, converter specification is given Vin_min=380V, Vin_nom=390V, Vin_max=400V, Vout=1000V, Pout=1200W, Fs=50kHz and Fr=100kHz.

1) Determining the transformer turns ratio(n)

Transformer turns ration(n) is the ratio between the number of turns on primary turns to the number of turns on secondary side in to gain of the converter. The nominal gain Mnom=1 for full-bridge converter then,

n=

=

_

∗ Mnom =

∗ 1 = 0.39…..……..... (26)

Maximum gain (Mmax)=_

_∗ Mnom =

∗ 1 = 1.026

…………………………………………………………. (27)

and Minimum gain (Mmin)=_

_∗ Mnom =

∗ 1 =

0.975…………………………………………………… (28) According to steps 1 and 2 selecting the value of Q is 0.4 and m is 6. Finding the minimum normalized switching frequency from the figure 16 is around 0.5 and calculating the minimum switching frequency is given by fsmin=Fxmin*Fr=50kHz.

Voltage gain verification by Qmax@Vmin=Qmax*_

=

0.4 ∗

= 0.38............................................................... (29)

And check for Kmax=1.5 > Mmax (hence no need to change m value).

2) Calculating the resonant components

Rac = ∗∗

∏ ∗ ∗=

∗∗

∏ ∗ ∗= 102.74Ω……….... (30)

Qmax= 0.4 =

. …………………………………….... (31)

and Fr = 100kHz =

∏√(∗)………………..………… (32)

Obtained the values of Cr= 38.73nF, Lr=65.41µH and Lm=327.03µH.

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C. DESIGN PARAMETER’S FOR SWITCHING MODULE

VIN= 1000V, VOUT=1000V pulsating DC, Iout=1.2A, switching frequency (Fsw)= 5kHz, Total time = 1ms, turn ON time=1µsec and load resistance is given by (33)

R=

=

.= 833.33 Ω ……………………………….. (33)

IV. SIMULATION RESULTS AND DISCUSSION:

Proposed system simulation is carried-out at MATLAB Simulink tool to create pulsating DC voltage from grid illustrated in figure 19. In the PFC boost converter PI controller receives an error when the voltage is compared to the reference voltage, which is minimized. The voltage differential, or steady-state error, is reduced. When the voltage at the source is measured, it equals one. Because the amplitude was calculated using the source's mod block voltage as a reference, the current reference. The constant value from the PI controller is multiplied by the unity amplitude voltage to create the current reference. The error is transmitted to the PI controller when the current reference is compared to the current in the reference is compared to the current in the inductor. Error pulses are generated, and switching is controlled as necessary. The inductor current will be half sinusoidal if the reference current is compelled to be half sinusoidal and next followed by the resonant converter for step-up voltage and pulsating switching device is connected with the resistive load as shown in above figure 19.

1) Simulation of EMI Filter

Figure 20: Simulation Diagram of EMI Filter

V. SIMULATION RESULTS

Output voltage and current waveform of the EMI filter as

shown in figure 21, in which both input voltage and current are sinusoidal at the designed values.

Figure 21: Output voltage and current waveform of EMI filter

Before power factor correction, the source voltage and current are out of phase, and the current is distorted, indicating a high THD around 282.93% for non-sinusoidal periodic before PFC as shown in figure 22. And figure 5 shows the result of before PFC THD measured by FFT analysis.

Figure 22: Before PFC AC Input voltage and current.

Figure 19: Simulation of proposed system using MATLAB Simulink

Figure 18: Design schematic of EMI filter and full-bridge diode rectifier with PFC boost converter

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Output voltage of the boost converter is around 390V with the input voltage 185V having a duty ratio of 0.52 shown in figure 23, with the reference of 390V.

Figure23: Output voltage of boost converter

Because the reference output voltage is multiplied with the phase of the input voltage and the results are compared with the inductor current error sent to the PI controller, the circuit output changes with the change in reference, as illustrated in figure 24 with reference 350V.

Figure 24: Output voltage of boost converter

After PFC the AC input voltage and input current are made sinusoidal and the power factor is made nearer to unity as shown in figure 25, hence THD also reduces from 282.93% to 4.53% at fundamental frequency as shown in figure 26 as per standard requirement.

Figure 25: After PFC the AC Input Voltage and Input Current

Figure 26: After PFC THD measured by FFT Analysis

The inductor current is continuous, and the source current is sinusoidal and of the same magnitude as the inductor current as shown in figure 27.

Figure 27: AC Input current and Inductor current

Input voltage of full-bridge converter that is the output of PFC boost converter as shown in figure 23. The PWM signal for switches is shown in figure 28, with 50% duty cycle and 180º phase shift. Before the resonant, the output voltage of the full-bridge converter is the same as square wave as shown in figure 29, after adding a resonant component in series with the primary of the transformer will make square wave to nearer sinusoidal wave as shown in figure 30, for primary voltage and by the step-up with high frequency transformer the voltage is stepping up from 390V to 1000V is shown in figure 31, for secondary voltage.

Figure 28: PWM signal for switches with 50% duty cycle

Figure 29: Before the resonant the output voltage of the full bridge

Figure 30: Primary voltage of transformer

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Figure 31: Secondary voltage of transformer

The resonant converter is operating at zero voltage switching (ZVS) operation of the converter when the switch

is conducting voltage across the switch is zero and current flowing it as shown in figure 32.

Figure 32: ZVS operation of the converter

The output voltage of the LLC full bridge resonant converter is around 1000V as shown in figure 33, and the required 1kV pulsating voltage and pulsating current from the proposed converter is shown in figure 34 and figure 35 respectively where the turn-on time is 1µsec and total time is 1msec.

Figure 33: Output voltage of the LLC full bridge converter

Figure 34: 1kV pulsed output voltage

Figure 35: 1.2Amp pulsed output current

Table 1: Components required for proposed system

SL.

NO

REQUIRED COMPONENTS VALUE

1. Common mode Inductor (LDM) 3mH

2. Differential mode Inductor (LCM) 41.25µH

3. Common mode Capacitor

(Cy=Cy1=Cy2)

5400pF

4. Differential mode Capacitor

(Cx=Cx1=Cx2)

1.46µF

5. CBridge 0.860µF

6. Boost Inductor (LBoost) 340µH

AC Main

Supply

voltage (V)

Rectifier

Vout

(V)

PFC Boost

Output

voltage (V)

Full-Bridge

output

voltage(V)

Full-Bridge

output

current (A)

Output

Power

(W)

THD

(%)

Output

voltage

ripple

(%)

185 261.62 387 998 1.18 1177.64 5.61 0.027

200 282.84 388 998 1.19 1188.81 4.88 0.048

220 311.11 389 1000 1.20 1199.94 4.56 0.0014

230 325.26 390 1000 1.20 1200 4.53 0.0032

240 339.41 390.5 1000.5 1.21 1215 4.49 0.00013

265 374.76 392 1000.9 1.22 1220 5.07 0.00019

Table 2: Supply variation all input and output voltages with THD and output voltage ripple are measured with constant load

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7. Boost Capacitor (CBoost) 1130µF

8. Resonant Inductor (Lr) 65.41µH

9. Resonant Capacitor (Cr) 38.73nF

10. Magnetising Inductor (Lm) 327.03µH

11. Output Filter Capacitor (Cout) 3.3µF

12. Load Resistance (RL) 833.33Ω

13. Transformer turns ratio (n)

experimentally

2.56

By load variation and supply variation all input, output voltages with THD and output voltage ripple are measured and tabulated in table 2 and table 3 respectively. From the figure 36 and figure 37 shows the graph for supply variation Vs THD and load variation Vs THD are compared with conventional, half-bridge and proposed converter and absorbed that THD is less than 6% for all converter as per as standards but has higher ripple output voltage as shown in figure 38 and figure 39, the graph for supply variation Vs output ripple voltage(v) and load variation vs output ripple voltage(v) is compared with conventional, half-bridge and

proposed converter and absorbed that proposed system has very less output ripple voltage than other converters is around 0.025% are shown. If we go for a half-bridge resonant converter then needs a large size of the transformer compare to proposed converter because the turns ratio of the half-bridge is around 5.2 and 2.56 for proposed converter, by knowing that turns ratio increases the size of the transformer is also increases hence it’s not implemented, and by the conventional full-converter

AC Main

supply

voltage

(V)

Rectifier

output

voltage

(V)

PFC Boost

output

voltage

(V)

Full-

Bridge

output

voltage (V)

Full-

Bridge

output

current (A)

Output

Power

(W)

THD

(%)

Load

Variat

ion

(%)

Output

voltage

ripple

(%)

230 325.26 389.2 995.8 1.21 1204.9 4.51 50 0.012

230 325.26 389 999.4 1.20 1199.2 4.48 70 0.0058

230 325.26 390 1000 1.20 1200 4.50 90 0.00019

230 325.26 390 1000 1.20 1200 4.53 100 0.00026

230 325.26 390 1000 1.19 1190 4.52 110 0.00085

230 325.26 390 999.9 1.18 1185 4.52 120 0.00025

Parameters Half-Bridge Resonant

converter

Conventional Full-Bridge

converter

Proposed System

Supply

variation (V)

THD

(%)

Output

voltage ripple (V)

THD

(%)

Output

voltage ripple (V)

THD

(%)

Output

voltage ripple (V)

185V 2.71 0.059 2.46 0.02 5.61 0.012

200V 2.87 0.060 2.68 0.06 4.88 0.0058

220V 3.15 0.061 2.98 0.05 4.56 0.00019

230V 3.35 0.059 3.19 0.04 4.53 0.00026

240V 3.52 0.055 3.31 0.07 4.49 0.00085

265 3.55 0.033 3.66 0.05 5.07 0.00025

Load

variation (%)

THD

(%)

Output

voltage ripple (V)

THD

(%)

Output

voltage ripple (V)

THD

(%)

Output

voltage ripple (V)

50 3.35 0.0516 3.15 0.09 4.51 0.027

70 3.36 0.021 3.14 0.05 4.48 0.0048

90 3.34 0.066 3.14 0.07 4.50 0.0014

100 3.35 0.059 3.15 0.04 4.53 0.0032

110 3.35 0.061 3.14 0.05 4.52 0.00013

120 3.34 0.069 3.14 0.07 4.52 0.00019

Table 3: Load variation all input and output voltages with THD and output voltage ripple are measured with constant AC input voltage

Table 4: Comparison of THD and output voltage ripple for half-bridge, conventional full-bridge and proposed converter

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Figure 36: Graph for supply variation Vs THD

hard switching is taking place hence switching loss will going to increase also, that needs extra passive components whereas the proposed system having a reduced turns ratio, less components, THD as per as standard and low output voltage ripple with constant output voltage is maintained by supply and low variation as shown in table 4.

Figure 37: Graph for Load variation Vs THD

Figure 38: Graph for supply variation Vs Output voltage ripple (V)

Figure 39: Graph for Load variation Vs Output voltage ripple (V)

VI. CONCLUSION

The power flow conversion of pulsed power applications is realized through the help of rectifier with the PFC boost converter, full-bridge resonant DC-DC converter and the switching module. In order to achieve high power quality and grid-stabilized DC voltage, a 1.5kW interface active PFC boost converter is used to make the power factor close to unity and THD is about 4.53%, with 390V regulated output DC voltage and full-bridge resonant DC-DC converter is used, in order to achieve zero voltage switching and stepping-up the voltage, thereby reducing switching loss and noise. According to the proposed design, a 1 kV DC voltage pulse output is obtained from the switching module, having a total pulse width of 1 millisecond and an on-time of 1 microsecond. The proposed system simulation was successfully carried out with resistive load in MATLAB/Simulink, and the results were tabulated by varying the power supply and load while maintains the output voltage constant. In addition, obtained results are compared with half-bridge resonant and conventional full-bridge converters which results the output voltage ripple is low, about 0.025% for the proposed system. The calculation and selection of the components for the proposed system is presented.

VII. REFERENCE

[1] Kotny, Jean-Luc, Thierry Duquesne, and Nadir Idir. "Design of EMI Filters for DC-DC converter." In 2010 IEEE Vehicle Power and Propulsion Conference, pp. 1-6. IEEE, 2010. [2] Mohan, P. Ram, M. Vijaya Kumar, and OV Raghava Reddy. "Simulation of a boost PFC converter with electro-magnetic interference filter." International Journal of Electrical and Computer Engineering 3 (2008): 957-961. [3] Yang, Liyu, Bing Lu, Wei Dong, Zhiguo Lu, Ming Xu, F. C. Lee, and W. G. Odendaal. "Modeling and characterization of a 1 KW CCM PFC converter for conducted EMI prediction." In Nineteenth Annual IEEE Applied Power Electronics Conference and Exposition, 2004. APEC'04., vol. 2, pp. 763-769. IEEE, 2004. [4] Umesh, Suma, L. Venkatesha, and A. Usha. "Active power factor correction technique for single phase full bridge rectifier." In 2014 International Conference on Advances in Energy Conversion Technologies (ICAECT), pp. 130-135. IEEE, 2014. [5] Babu, Kotari Sri Harsha, Ramesh Holde, Bhoopendra Kumar Singh, and Vinod S. Chippalkatti. "Power factor correction using Boost converter operating in CCM for front-end AC to DC conversion." In 2018 Technologies for Smart-City Energy Security and Power (ICSESP), pp. 1-6. IEEE, 2018. [6] Dwivedi, Prakash, and Sourav Bose. "Design and Analysis of Closed-Loop Control for Single-Phase Boost Rectifier by using Fractional Order PI Controller." In IECON 2020 The 46th Annual Conference of the IEEE Industrial Electronics Society, pp. 2951-2956. IEEE, 2020. [7] Kar, Abhishek, and Mainak Sengupta. "Design, analysis, fabrication and testing of a 3kW power factor correction boost rectifier." In 2016 IEEE International Conference on Power Electronics, Drives and Energy Systems (PEDES), pp. 1-6. IEEE, 2016. [8] Hart, Daniel W. Power electronics. Tata McGraw-Hill Education, 2011. [9] Bellec, Quentin, Jean-Claude Le Claire, Mohamed Fouad Benkhoris, and Peyofougou Coulibaly. "Power factor correction and DC voltage control limits for arc welding application using pulsed current." In IECON 2018- 44th Annual Conference of the IEEE Industrial Electronics Society, pp. 1406-1411. IEEE, 2018. [10] Data sheet UCC28180D- Texas Instruments Controller for power factor correction. [11] TIDA-00779 230-V,3.5-kW PFC with >98% Efficiency, Optimized for BOM and size reference Design. [12] On semiconductor power factor correction Handbook.

0

2

4

6

185 200 220 230 240 260

TH

D

Supply Variation(V)

Supply Variation Vs THDProposed Converter Conventional converterHalf-Bridge converter

0

5

50 70 90 100 110 120

TH

D

Load Variation(%)

Load Variation Vs THD

Proposed Converter Conventional converterHalf-Bridge converter

0

0.02

0.04

0.06

0.08

185 200 220 230 240 265Out

put

vo

ltag

e R

ipp

le (

V)

Supply Variation (V)

Supply Variation (V) Vs Output voltage Ripple (V)Proposed Converter Conventional converterHalf-Bridge converter

0

0.02

0.04

0.06

0.08

0.1

50 70 90 100 110 120

Out

put

Vol

tage

Rip

ple

(V

)

Load Variation(%)

Output Voltage Ripple(V) Vs Load VariationResonant Converter Conventional converterHalf-Bridge converter

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[13] Sharthi, M., and R. Seyezhai. "A simple design and simulation of full bridge LLC resonant DC-DC converter for PV applications." Middle-East J. Sci. Res 23 (2015): 285-292. [14] Abdel-Rahman, Sam. "Resonant LLC converter: Operation and design." Infineon Technologies North America (IFNA) Corp. (2012). [15] Fang, Zhijian, Shanxu Duan, Changsong Chen, Xi Chen, and Jianxing Zhang. "Optimal design method for LLC resonant converter with wide range output voltage." In 2013 Twenty-Eighth Annual IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 2106-2111. IEEE, 2013. [16] Xu, Minghui, Jing Ji, Zhan Li, Yuxi Wang, Jianhua Du, and Hao Ma. "Design methodology of LLC converters based on simplified mode analysis for wide output range." In 2016 IEEE 25th International Symposium on Industrial Electronics (ISIE), pp. 465-470. IEEE, 2016. [17] Ram, Subhash Kumar, Anand Abhishek, P. K. Pedapati, B. K. Verma, A. K. Dhakar, and Rahul Varma. "Development of high voltage pulse power

supply for microwave tube applications." In 2017 14th IEEE India Council International Conference (INDICON), pp. 1-5. IEEE, 2017. [18] Varma, Rahul, and K. S. Sangwan. "Development of a solid-state versatile pulsar for high voltage and high-power applications." In 2009 IEEE Pulsed Power Conference, pp. 1312-1316. IEEE, 2009. [19] Hasanien, B. M., and Khairy FA Sayed. "Current source ZCS PFM DC-DC converter for magnetron power supply." In 2008 12th International Middle-East Power System Conference, pp. 464-469. IEEE, 2008. [20] Song, Byeong-Mun, Moon-Ho Kye, and Rae-Young Kim. "Design of a cost-effective DC-DC converter with high power density for magnetron power supplies." In The 2010 International Power Electronics Conference-ECCE ASIA-, pp. 137-141. IEEE, 2010.

ISSN No: 0932-4747

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Zeichen Journal

Volume 7, Issue 8, 2021